1
LTC1474/LTC1475
Low Quiescent Current
High Efficiency Step-Down
Converters
LOAD CURRENT (mA)
EFFICIENCY (%)
100
90
80
70
60
50 0.03 3 300
1474/75 TA01
0.3 30
V
IN
= 5V
V
IN
= 10V
V
IN
= 15V
L = 100µH
V
OUT
= 3.3V
R
SENSE
= 0
LTC1474 Efficiency
Figure 1. High Efficiency Step-Down Converter
V
IN
GND
SENSE
LBI
RUN
V
FB
LBO
SW
LTC1474-3.3
+
6
3
8
1
2
5
7
4
100k
LOW BATTERY IN
RUN SHDN
0.1µF
10µF
25V
V
IN
4V TO 18V LOW BATTERY OUT
1474/75 F01
L1
100µH
D1
MBR0530
+
100µF
6.3V
V
OUT
3.3V AT 250mA
L1 = SUMIDA CDRH74-101
FEATURES
DESCRIPTION
U
High Efficiency: Over 92% Possible
Very Low Standby Current: 10
µ
A Typ
Available in Space Saving 8-Lead MSOP Package
Internal 1.4 Power Switch (V
IN
= 10V)
Wide V
IN
Range: 3V to 18V Operation
Very Low Dropout Operation: 100% Duty Cycle
Low-Battery Detector Functional During Shutdown
Programmable Current Limit with Optional
Current Sense Resistor (10mA to 400mA Typ)
Short-Circuit Protection
Few External Components Required
Active Low Micropower Shutdown: I
Q
= 6µA Typ
Pushbutton On/Off (LTC1475 Only)
3.3V, 5V and Adjustable Output Versions
The LTC
®
1474/LTC1475 series are high efficiency step-
down converters with internal P-channel MOSFET power
switches that draw only 10µA typical DC supply current at
no load while maintaining output voltage. The LTC1474
uses logic-controlled shutdown while the LTC1475 fea-
tures pushbutton on/off.
The low supply current coupled with Burst Mode
TM
opera-
tion enables the LTC1474/LTC1475 to maintain high effi-
ciency over a wide range of loads. These features, along
with their capability of 100% duty cycle for low dropout
and wide input supply range, make the LTC1474/LTC1475
ideal for moderate current (up to 300mA) battery-powered
applications.
The peak switch current is user-programmable with an
optional sense resistor (defaults to 325mA minimum if not
used) providing a simple means for optimizing the design
for lower current applications. The peak current control
also provides short-circuit protection and excellent start-
up behavior. A low-battery detector that remains functional
in shutdown is provided .
The LTC1474/LTC1475 series availability in 8-lead MSOP
and SO packages and need for few additional components
provide for a minimum area solution.
APPLICATIONS
U
Cellular Telephones and Wireless Modems
4mA to 20mA Current Loop Step-Down Converter
Portable Instruments
Battery-Operated Digital Devices
Battery Chargers
Inverting Converters
Intrinsic Safety Applications
TYPICAL APPLICATION
U
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
2
LTC1474/LTC1475
ABSOLUTE MAXIMUM RATINGS
W
WW
U
Operating Ambient Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ............................................ 40°C to 85°C
Junction Temperature (Note 1)............................ 125°C
Storage Temperature Range ................. 65°C to 150°C
Lead Temperature (Soldering, 10 sec)..................300°C
Input Supply Voltage (V
IN
).........................0.3V to 20V
Switch Current (SW, SENSE).............................. 750mA
Switch Voltage (SW) ............. (V
IN
20V) to (V
IN
+ 0.3V)
V
FB
(Adjustable Versions) ..........................0.3V to 12V
V
OUT
(Fixed Versions)................................ –0.3V to 20V
LBI, LBO ....................................................0.3V to 20V
RUN, SENSE ..................................0.3V to (V
IN
+ 0.3V)
PACKAGE/ORDER INFORMATION
W
UU
S8 PART MARKING
1475
1475I
147533
14755
Consult factory for Military grade parts.
1474
1474I
147433
14745
474I33
1474I5
LTBW
LTCR
LTCS
MS8 PART MARKING
LTBK
LTCP
LTCQ
MS8 PART MARKING S8 PART MARKING
TOP VIEW
S8 PACKAGE
8-LEAD PLASTIC SO
1
2
3
4
8
7
6
5
V
OUT
/V
FB
LBO
LBI
GND
RUN
V
IN
SENSE
SW
TOP VIEW
S8 PACKAGE
8-LEAD PLASTIC SO
1
2
3
4
8
7
6
5
V
OUT
/V
FB
LBO
LBI/OFF
GND
ON
V
IN
SENSE
SW
1
2
3
4
V
OUT
/V
FB
LBO
LBI
GND
8
7
6
5
RUN
V
IN
SENSE
SW
TOP VIEW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
1
2
3
4
V
OUT
/V
FB
LBO
LBI/OFF
GND
8
7
6
5
ON
V
IN
SENSE
SW
TOP VIEW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
T
JMAX
= 125°C, θ
JA
= 150°C/W T
JMAX
= 125°C, θ
JA
= 110°C/W T
JMAX
= 125°C, θ
JA
= 110°C/WT
JMAX
= 125°C, θ
JA
= 150°C/W
ORDER PART NUMBER ORDER PART NUMBER ORDER PART NUMBERORDER PART NUMBER
LTC1475CMS8
LTC1475CMS8-3.3
LTC1475CMS8-5
LTC1474CS8
LTC1474IS8
LTC1474CS8-3.3
LTC1474CS8-5
LTC1474IS8-3.3
LTC1474IS8-5
LTC1474CMS8
LTC1474CMS8-3.3
LTC1474CMS8-5
LTC1475CS8
LTC1475IS8
LTC1475CS8-3.3
LTC1475CS8-5
3
LTC1474/LTC1475
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
V
FB
Feedback Voltage I
LOAD
= 50mA 1.205 1.230 1.255 V
LTC1474/LTC1475
V
OUT
Regulated Output Voltage I
LOAD
= 50mA
LTC1474-3.3/LTC1475-3.3 3.234 3.300 3.366 V
LTC1474-5/LTC1475-5 4.900 5.000 5.100 V
I
FB
Feedback Current 030 nA
LTC1474/LTC1475 Only
I
SUPPLY
No Load Supply Current (Note 3) I
LOAD
= 0 (Figure 1 Circuit) 10 µA
V
OUT
Output Voltage Line Regulation V
IN
= 7V to 12V, I
LOAD
= 50mA 5 20 mV
Output Voltage Load Regulation I
LOAD
= 0mA to 50mA 2 15 mV
Output Ripple I
LOAD
= 10mA 50 mV
P-P
I
Q
Input DC Supply Current (Note 2) (Exclusive of Driver Gate Charge Current)
Active Mode (Switch On) V
IN
= 3V to 18V 100 175 µA
Sleep Mode (Note 3) V
IN
= 3V to 18V 9 15 µA
Shutdown V
IN
= 3V to 18V, V
RUN
= 0V 6 12 µA
R
ON
Switch Resistance I
SW
= 100mA 1.4 1.6
I
PEAK
Current Comp Max Current Trip Threshold R
SENSE
= 0325 400 mA
R
SENSE
= 1.170 76 85 mA
V
SENSE
Current Comp Sense Voltage Trip Threshold 90 100 110 mV
V
HYST
Voltage Comparator Hysteresis 5mV
t
OFF
Switch Off-Time V
OUT
at Regulated Value 3.5 4.75 6.0 µs
V
OUT
= 0V 65 µs
V
LBI, TRIP
Low Battery Comparator Threshold 1.16 1.23 1.27 V
V
RUN
Run/ON Pin Threshold 0.4 0.7 1.0 V
V
LBI, OFF
OFF Pin Threshold (LTC1475 Only) 0.4 0.7 1.0 V
I
LBO, SINK
Sink Current into Pin 2 V
LBI
= 0V, V
LBO
= 0.4V 0.45 0.70 mA
I
RUN, SOURCE
Source Current from Pin 8 V
RUN
= 0V 0.4 0.8 1.2 µA
I
SW, LEAK
Switch Leakage Current V
IN
= 18V, V
SW
= 0V, V
RUN
= 0V 0.015 1 µA
I
LBI, LEAK
Leakage Current into Pin 3 V
LBI
= 18V, V
IN
= 18V 0 0.1 µA
I
LBO, LEAK
Leakage Current into Pin 2 V
LBI
= 2V, V
LBO
= 5V 0 0.5 µA
ELECTRICAL CHARACTERISTICS
TA = 25°C, VIN = 10V, VRUN = open, RSENSE = 0, unless otherwise noted.
The denotes specifications which apply over the full operating
temperature range.
Note 1: T
J
is calculated from the ambient temperature T
A
and power
dissipation P
D
according to the following formulas:
LTC1474CS8/LTC1475CS8: T
J
= T
A
+ (P
D
• 110°C/W)
LTC1474CMS8/LTC1475CMS8: T
J
= T
A
+ (P
D
• 150°C/W)
Note 2: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 3: No load supply current consists of sleep mode DC current (9µA
typical) plus a small switching component (about 1µA for Figure 1 circuit)
necessary to overcome Schottky diode and feedback resistor leakage.
4
LTC1474/LTC1475
TYPICAL PERFORMANCE CHARACTERISTICS
UW
Current Trip Threshold vs
Temperature Switch Resistance vs
Input Voltage Supply Current in Shutdown
TEMPERATURE (°C)
0
CURRENT TRIP THRESHOLD (mA)
80
1474/75 G04
20 40 60
500
400
300
200
100
0
V
IN
= 10V
R
SENSE
= 0
R
SENSE
= 1.1
INPUT VOLTAGE (V)
0
R
DS(ON)
()
20
1474/75 G05
510 15
5
4
3
2
1
0
T = 70°C
T = 25°C
INPUT VOLTAGE (V)
0
SUPPLY CURRENT (µA)
20
1474/75 G06
510 15
10.0
7.5
5.0
2.5
0
Switch Leakage Current vs
Temperature VIN DC Supply Current Off-Time vs Output Voltage
% OF REGULATED OUTPUT VOLTAGE (%)
0
OFF-TIME (µs)
100
1474/75 G09
20 40 60 80
80
60
40
20
0
V
IN
= 10V
INPUT VOLTAGE (V)
0
SUPPLY CURRENT (µA)
4816 20
1474/75 G08
12
120
100
80
60
40
20
0
ACTIVE MODE
SLEEP MODE
TEMPERATURE (°C)
0
LEAKAGE CURRENT (µA)
1474/75 G07
40 60 80 100
1.0
0.8
0.6
0.4
0.2
020
VIN = 18V
Efficiency vs Input Voltage Line Regulation Load Regulation
INPUT VOLTAGE (V)
0
EFFICIENCY (%)
4816
1474/75 G01
12
100
95
90
85
80
75
70
I
LOAD
= 25mA
I
LOAD
= 200mA
FIGURE 1 CIRCUIT
L: CDRH73-101
I
LOAD
= 1mA
INPUT VOLTAGE (V)
0
V
OUT
(mV)
4816
1474/75 G02
12
40
30
20
10
0
–10
–20
FIGURE 1 CIRCUIT
I
LOAD
= 100mA
R
SENSE
= 0.33
R
SENSE
= 0
LOAD CURRENT (mA)
0
VOUT (mV)
40
30
20
10
0
–10
–20
–30
1474/75 G03
100 250
50 150 200 300
FIGURE 1 CIRCUIT
VIN = 15V
VIN = 10V
VIN = 5V
5
LTC1474/LTC1475
FUNCTIONAL DIAGRA
UU
W
LTC1474: LBI
LTC1475: LBI/OFF
+
+
+
×
1µA
LBI/OFF
LBO
4.75µs
1-SHOT
1.23V
REFERENCE
GND
1474/75 FD
4
2
8
1.23V
1M
3M
(5V VERSION)
1.68M
(3.3V VERSION)
READY
STRETCH
V
CC
V
IN
V
IN
V
OUT
V
OUT
/V
FB
1×
C
ON
V
LB
ON 5
SW
SENSE
R
SENSE
(OPTIONAL)
20×
100mV
WAKEUP
TRIGGER OUT
LTC1474: RUN
LTC1475: ON
1
5
6
7
+
+
×
CONNECTION NOT PRESENT IN LTC1474 SERIES
CONNECTION PRESENT IN LTC1474 SERIES ONLY
3OUTPUT DIVIDER IS
IMPLEMENTED EXTERNALLY IN
ADJUSTABLE VERSIONS
PIN FUNCTIONS
UUU
V
OUT
/V
FB
(Pin 1): Feedback of Output Voltage. In the fixed
versions, an internal resistive divider divides the output
voltage down for comparison to the internal 1.23V refer-
ence. In the adjustable versions, this divider must be
implemented externally.
LBO (Pin 2): Open Drain Output of the Low Battery
Comparator. This pin will sink current when Pin 3 is below
1.23V.
LBI/OFF (Pin 3): Input to Low Battery Comparator. This
input is compared to the internal 1.23V reference. For the
LTC1475, a momentary ground on this pin puts regulator
in shutdown mode.
GND (Pin 4): Ground Pin.
SW (Pin 5): Drain of Internal PMOS Power Switch. Cath-
ode of Schottky diode must be closely connected to this
pin.
SENSE (Pin 6): Current Sense Input for Monitoring Switch
Current and Source of Internal PMOS Power Switch.
Maximum switch current is programmed with a resistor
between SENSE and V
IN
pins.
V
IN
(Pin 7): Main Supply Pin.
RUN/ON (Pin 8): On LTC1474, voltage level on this pin
controls shutdown/run mode (ground = shutdown, open/
high = run). On LTC1475, a momentary ground on this pin
puts regulator in run mode. A 100k series resistor must be
used between Pin 8 and the switch or control voltage.
6
LTC1474/LTC1475
OPERATIO
U
(Refer to Functional Diagram)
The LTC1474/LTC1475 are step-down converters with
internal power switches that use Burst Mode operation to
keep the output capacitor charged to the proper output
voltage while minimizing the quiescent current. Burst
Mode operation functions by using short “burst” cycles to
ramp the inductor current through the internal power
switch and external Schottky diode, followed by a sleep
cycle where the power switch is off and the load current is
supplied by the output capacitor. During sleep mode, the
LTC1474/LTC1475 draw only 9µA typical supply current.
At light loads, the burst cycles are a small percentage of
the total cycle time; thus the average supply current is very
low, greatly enhancing efficiency.
Burst Mode Operation
At the beginning of the burst cycle, the switch is turned on
and the inductor current ramps up. At this time, the internal
current comparator is also turned on to monitor the switch
current by measuring the voltage across the internal and
optional external current sense resistors. When this volt-
age reaches 100mV, the current comparator trips and
pulses the 1-shot timer to start a 4.75µs off-time during
which the switch is turned off and the inductor current
ramps down. At the end of the off-time, if the output
voltage is less than the voltage comparator threshold, the
switch is turned back on and another cycle commences. To
minimize supply current, the current comparator is turned
on only during the switch-on period when it is needed to
monitor switch current. Likewise, the 1-shot timer will only
be on during the 4.75µs off-time period.
The average inductor current during a burst cycle will
normally be greater than the load current, and thus the
output voltage will slowly increase until the internal volt-
age comparator trips. At this time, the LTC1474/LTC1475
go into sleep mode, during which the power switch is off
and only the minimum required circuitry is left on: the
voltage comparator, reference and low battery compara-
tor. During sleep mode, with the output capacitor supply-
ing the load current, the V
FB
voltage will slowly decrease
until it reaches the lower threshold of the voltage com-
parator (about 5mV below the upper threshold). The
voltage comparator then trips again, signaling the LTC1474/
LTC1475 to turn on the circuitry necessary to begin a new
burst cycle.
Peak Inductor Current Programming
The current comparator provides a means for program-
ming the maximum inductor/switch current for each switch
cycle. The 1X sense MOSFET, a portion of the main power
MOSFET, is used to divert a sample (about 5%) of the
switch current through the internal 5 sense resistor. The
current comparator monitors the voltage drop across the
series combination of the internal and external sense
resistors and trips when the voltage exceeds 100mV. If the
external sense resistor is not used (Pins 6 and 7 shorted),
the current threshold defaults to the 400mA maximum due
to the internal sense resistor.
Off-Time
The off-time duration is 4.75µs when the feedback voltage
is close to the reference; however, as the feedback voltage
drops, the off-time lengthens and reaches a maximum
value of about 65µs when this voltage is zero. This ensures
that the inductor current has enough time to decay when
the reverse voltage across the inductor is low such as
during short circuit.
Shutdown Mode
Both LTC1474 and LTC1475 provide a shutdown mode
that turns off the power switch and all circuitry except for
the low battery comparator and 1.23V reference, further
reducing DC supply current to 6µA typical. The LTC1474’s
run/shutdown mode is controlled by a voltage level at the
RUN pin (ground = shutdown, open/high = run). The
LTC1475’s run/shutdown mode, on the other hand, is
controlled by an internal S/R flip-flop to provide pushbutton
on/off control. The flip-flop is set (run mode) by a momen-
tary ground at the ON pin and reset (shutdown mode) by
a momentary ground at the LBI/OFF pin.
Low Battery Comparator
The low battery comparator compares the voltage on the
LBI pin to the internal reference and has an open drain
N-channel MOSFET at its output. If LBI is above the
reference, the output FET is off and the LBO output is high
impedance. If LBI is below the reference, the output FET is
on and sinks current. The comparator is still active in
shutdown.
7
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
ments. Lower peak currents have the advantage of lower
output ripple (V
OUT
= I
PEAK
• ESR), lower noise, and less
stress on alkaline batteries and other circuit components.
Also, lower peak currents allow the use of inductors with
smaller physical size.
Peak currents as low as 10mA can be programmed with
the appropriate sense resistor. Increasing R
SENSE
above
10, however, gives no further reduction of I
PEAK
.
For R
SENSE
values less than 1, it is recommended that
the user parallel standard resistors (available in values
1) instead of using a special low valued shunt resistor.
Although a single resisor could be used with the desired
value, these low valued shunt resistor types are much
more expensive and are currently not available in case
sizes smaller than 1206. Three or four 0603 size standard
resistors require about the same area as one 1206 size
current shunt resistor at a fraction of the cost.
At higher supply voltages and lower inductances, the peak
currents may be slightly higher due to current comparator
overshoot and can be predicted from the second term in
the following equation:
IR
VV
L
PEAK SENSE
IN OUT
=++
()
()
01
025
25 10
7
.
.
.
(2)
Note that R
SENSE
only sets the maximum inductor current
peak. At lower dI/dt (lower input voltages and higher
inductances), the observed peak current at loads less than
I
MAX
may be less than this calculated peak value due to the
voltage comparator tripping before the current ramps up
high enough to trip the current comparator. This effect
improves efficiency at lower loads by keeping the I
2
R
losses down (see Efficiency Considerations section).
Inductor Value Selection
Once R
SENSE
and I
PEAK
are known, the inductor value can
be determined. The minimum inductance recommended
as a function of I
PEAK
and I
MAX
can be calculated from:
LVVt
II
MIN OUT D OFF
PEAK MAX
+
()
075.
(3)
where t
OFF
= 4.75µs.
MAXIMUM OUTPUT CURRENT (mA)
0
RSENSE ()
5
4
3
2
1
050 100 150 200
1474/75 F02
250 300
FOR LOWEST NOISE
FOR BEST EFFICIENCY
Figure 2. RSENSE Selection
The basic LTC1474/LTC1475 application circuit is shown
in Figure 1, a high efficiency step-down converter. External
component selection is driven by the load requirement and
begins with the selection of R
SENSE
. Once R
SENSE
is
known, L can be chosen. Finally D1, C
IN
and C
OUT
are
selected.
R
SENSE
Selection
The current sense resistor (R
SENSE
) allows the user to
program the maximum inductor/switch current to opti-
mize the inductor size for the maximum load. The LTC1474/
LTC1475 current comparator has a maximum threshold of
100mV/(R
SENSE
+ 0.25). The maximum average output
current I
MAX
is equal to this peak value less half the peak-
to-peak ripple current I
L
.
Allowing a margin for variations in the LTC1474/LTC1475
and external components, the required R
SENSE
can be
calculated from Figure 2 and the following equation:
R
SENSE
= (0.067/I
MAX
) – 0.25 (1)
for 10mA < I
MAX
< 200mA.
For I
MAX
above 200mA, R
SENSE
is set to zero by shorting
Pins 6 and 7 to provide the maximum peak current of
400mA (limited by the fixed internal sense resistor). This
400mA default peak current can be used for lower I
MAX
if
desired to eliminate the need for the sense resistor and
associated decoupling capacitor. However, for optimal
performance, the peak inductor current should be set to no
more than what is needed to meet the load current require-
8
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
If the L
MIN
calculated is not practical, a larger I
PEAK
should
be used. Although the above equation provides the mini-
mum, better performance (efficiency, line/load regulation,
noise) is usually gained with higher values. At higher
inductances, peak current and frequency decrease (im-
proving efficiency) and inductor ripple current decreases
(improving noise and line/load regulation). For a given
inductor type, however, as inductance is increased, DC
resistance (DCR) increases, increasing copper losses,
and current rating decreases, both effects placing an
upper limit on the inductance. The recommended range of
inductances for small surface mount inductors as a func-
tion of peak current is shown in Figure 3. The values in this
range are a good compromise between the trade-offs
discussed above. If space is not a premium, inductors with
larger cores can be used, which extends the recom-
mended range of Figure 3 to larger values.
section, increased inductance requires more turns of wire
and therefore copper losses will increase.
Ferrite and Kool Mµ
designs have very low core loss and
are preferred at high switching frequencies, so design
goals can concentrate on copper loss and preventing
saturation. Ferrite core material saturates “hard,” which
means that inductance collapses abruptly when the peak
design current is exceeded. This results in an abrupt
increase in inductor current above I
PEAK
and consequent
increase in voltage ripple.
Do not allow the core to satu-
rate!
Coiltronics, Coilcraft, Dale and Sumida make high
performance inductors in small surface mount packages
with low loss ferrite and Kool Mµ cores and work well in
LTC1474/LTC1475 regulators.
Catch Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As V
IN
approaches V
OUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition, the diode must
safely handle I
PEAK
at close to 100% duty cycle.
To maximize both low and high current efficiency, a fast
switching diode with low forward drop and low reverse
leakage should be used. Low reverse leakage current is
critical to maximize low current efficiency since the leak-
age can potentially approach the magnitude of the LTC1474/
LTC1475 supply current. Low forward drop is critical for
high current efficiency since loss is proportional to for-
ward drop. These are conflicting parameters (see Table 1),
but a good compromise is the MBR0530 0.5A Schottky
diode specified in the application circuits.
Table 1. Effect of Catch Diode on Performance
FORWARD NO LOAD
DIODE (D1) LEAKAGE DROP SUPPLY CURRENT EFFICIENCY*
BAS85 200nA 0.6V 9.7µA 77.9%
MBR0530 1µA 0.4V 10µA 83.3%
MBRS130 20µA 0.3V 16µA 84.6%
*Figure 1 circuit with V
IN
= 15V, I
OUT
= 0.1A
Kool Mµ is a registered trademark of Magnetics, Inc.
PEAK INDUCTOR CURRENT (mA)
10
50
500
INDUCTOR VALUE (µH)
100
1000
100 1000
1474/75 F03
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, as discussed in the previous
Figure 3. Recommended Inductor Values
9
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
C
IN
and C
OUT
Selection
At higher load currents, when the inductor current is
continuous, the source current of the P-channel MOSFET
is a square wave of duty cycle V
OUT
/V
IN
. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
capacitor current is given by:
CVVV
V
IN OUT IN OUT
IN
Required I = I
RMS MAX
()
[]
12/
This formula has a maximum at V
IN
= 2V
OUT
, where
I
RMS
= I
OUT
/2. This simple worst-case condition is com-
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Do not underspecify this component. An addi-
tional 0.1µF ceramic capacitor is also required on V
IN
for
high frequency decoupling.
The selection of C
OUT
is driven by the required effective
series resistance (ESR) to meet the output voltage ripple
and line regulation requirements. The output voltage ripple
during a burst cycle is dominated by the output capacitor
ESR and can be estimated from the following relation:
25mV < V
OUT, RIPPLE
= I
L
• ESR
where I
L
I
PEAK
and the lower limit of 25mV is due to the
voltage comparator hysteresis. Line regulation can also
vary with C
OUT
ESR in applications with a large input
voltage range and high peak currents.
ESR is a direct function of the volume of the capacitor.
Manufacturers such as Nichicon, AVX and Sprague should
be considered for high performance capacitors. The
OS-CON semiconductor dielectric capacitor available from
SANYO has the lowest ESR for its size at a somewhat
higher price. Typically, once the ESR requirement is satis-
fied, the capacitance is adequate for filtering. For lower
current applications with peak currents less than 50mA,
10µF ceramic capacitors provide adequate filtering and
are a good choice due to their small size and almost
negligible ESR. AVX and Marcon are good sources for
these capacitors.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations.
To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by:
I
RMS
= I
PEAK
/2
Once the ESR requirement for C
OUT
has been met, the
RMS current rating generally far exceeds the I
RIPPLE(P-P)
requirement.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting efficiency and which change would produce the
most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of the
losses in LTC1474/LTC1475 circuits: V
IN
current, I
2
R
losses and catch diode losses.
1. The V
IN
current is due to two components: the DC bias
current and the internal P-channel switch gate charge
current. The DC bias current is 9µA at no load and
increases proportionally with load up to a constant
100µA during continuous mode. This bias current is so
10
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
small that this loss is negligible at loads above a
milliamp but at no load accounts for nearly all of the
loss. The second component, the gate charge current,
results from switching the gate capacitance of the
internal P-channel switch. Each time the gate is switched
from high to low to high again, a packet of charge dQ
moves from V
IN
to ground. The resulting dQ/dt is the
current out of V
IN
which is typically much larger than the
DC bias current. In continuous mode, I
GATECHG
= fQ
P
where Q
P
is the gate charge of the internal switch. Both
the DC bias and gate charge losses are proportional to
V
IN
and thus their effects will be more pronounced at
higher supply voltages.
2. I
2
R losses are predicted from the internal switch,
inductor and current sense resistor. At low supply
voltages where the switch on-resistance is higher and
the switch is on for longer periods due to higher duty
cycle, the switch losses will dominate. Keeping the peak
currents low with the appropriate R
SENSE
and with
larger inductance helps minimize these switch losses.
At higher supply voltages, these losses are proportional
to load and result in the flat efficiency curves seen in
Figure 1.
3. The catch diode loss is due to the V
D
I
D
loss as the diode
conducts current during the off-time and is more pro-
nounced at high supply voltage where the on-time is
short. This loss is proportional to the forward drop.
However, as discussed in the Catch Diode section,
diodes with lower forward drops often have higher
leakage current, so although efficiency is improved, the
no load supply current will increase.
Adjustable Applications
For adjustable versions, the output voltage is programmed
with an external divider from V
OUT
to V
FB
(Pin 1) as shown
in Figure 4. The regulated voltage is determined by:
VOUT =1.23 1+ R2
R1
(4)
To minimize no-load supply current, resistor values in the
megohm range should be used. The increase in supply
current due to the feedback resistors can be calculated
from:
∆=+
IV
RR
V
V
VIN OUT OUT
IN
12
A 10pF feedforward capacitor across R2 is necessary due
to the high impedances to prevent stray pickup and
improve stability.
GND
V
FB
LTC1474
LTC1475
1
4
1474/75 F04
R2
R1
10pF
V
OUT
LBI
LBO
LTC1474/LTC1475
1474/75 F05
R4
R3
1.23V
REFERENCE
V
IN
+
Figure 4. LTC1474/LTC1475 Adjustable Configuration
Figure 5. Low Battery Comparator
Low Battery Comparator
The LTC1474/LTC1475 have an on-chip low battery com-
parator that can be used to sense a low battery condition
when implemented as shown in Figure 5. The resistive
divider R3/R4 sets the comparator trip point as follows:
VR
R
TRIP
=+
123 1 4
3
.
The divided down voltage at the LBI pin is compared to the
internal 1.23V reference. When V
LBI
< 1.23V, the LBO
output sinks current. The low battery comparator is active
all the time, even during shutdown mode.
11
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
LTC1475 Pushbutton On/Off and
Microprocessor Interface
The LTC1475 provides pushbutton control of power on/off
for use with handheld products. A momentary ground on
the ON pin sets an internal S/R latch to run mode while a
momentary ground on the LBI/OFF pin resets the latch to
shutdown mode. See Figure 6 for a comparsion of on/off
operation of the LTC1474 and LTC1475 and Figure 7 for a
comparison of the circuit implementations.
In the LTC1475, the LBI/OFF pin has a dual function as
both the shutdown control pin and the low battery com-
parator input. Since the “OFF” pushbutton is normally
open, it does not affect the normal operation of the low
battery comparator. In the unpressed state, the LBI/OFF
input is the divided down input voltage from the resistive
divider to the internal low battery comparator and will
normally be above or just below the trip threshold of
1.23V. When shutdown mode is desired, the LBI/OFF pin
is pulled below the 0.7V threshold to invoke shutdown.
the depressed switch state is detected by the microcon-
troller through its input. The microcontroller then pulls the
LBI/OFF pin low with the connection to one of its ouputs.
With the LBI/OFF pin low, the LTC1475 powers down
turning the microcontroller off. Note that since the I/O pins
of most microcontrollers have a reversed bias diode
between input and supply, a blocking diode with less than
1µA leakage is necessary to prevent the powered down
microcontroller from pulling down on the ON pin.
Figure 19 in the Typical Applications section shows how to
use the low battery comparator to provide a low battery
lockout on the “ON” switch. The LBO output disconnects
the pushbutton from the ON pin when the comparator has
tripped, preventing the LTC1475 from attempting to start
up again until V
IN
is increased.
RUN
RUN SHUTDOWN RUN
RUN
1474/75 F06
SHUTDOWN
RUN
MODE
ON OVERRIDES LBI/OFF
WHILE ON IS LOW
LTC1474
LTC1475
ON
LBI/OFF
MODE
Figure 6. Comparison of LTC1474 and LTC1475
Run/Shutdown Operation
The ON pin has precedence over the LBI/OFF pin. As seen
in Figure 6, if both pins are grounded simultaneously, run
mode wins.
Figure 18 in the Typical Applications section shows an
example for the use of the LTC1475 to control on/off of a
microcontroller with a single pushbutton. With both the
microcontroller and LTC1475 off, depressing the
pushbutton grounds the LTC1475 ON pin and starts up the
LTC1475 regulator which then powers up the microcon-
troller. When the pushbutton is depressed a second time,
1474/75 F07
RUN ON
OFF
RUN ON
LBI/OFF
100k
LTC1474 LTC1475
100k
V
IN
Figure 7. Simplified Implementation of
LTC1474 and LTC1475 On/Off
Absolute Maximum Ratings and Latchup Prevention
The absolute maximum ratings specify that SW (Pin 5) can
never exceed V
IN
(Pin 7) by more than 0.3V. Normally this
situation should never occur. It could, however, if the
output is held up while the supply is pulled down. A
condition where this could potentially occur is when a
battery is supplying power to an LTC1474 or LTC1475
regulator and also to one or more loads in parallel with the
the regulator’s V
IN
. If the battery is disconnected while the
LTC1474 or LTC1475 regulator is supplying a light load
and one of the parallel circuits is a heavy load, the input
capacitor of the LTC1474 or LTC1475 regulator could be
pulled down faster than the output capacitor, causing the
absolute maximum ratings to be exceeded. The result is
often a latchup which can be destructive if V
IN
is reapplied.
Battery disconnect is possible as a result of mechanical
stress, bad battery contacts or use of a lithium-ion battery
12
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
Figure 8. Preventing Absolute Maximum
Ratings from Being Exceeded
1474/75 F08
V
IN
V
OUT
LATCHUP
PROTECTION
SCHOTTKY
SW
LTC1474
LTC1475
+
with a built-in internal disconnect. The user needs to
assess his/her application to determine whether this situ-
ation could occur. If so, additional protection is necessary.
Prevention against latchup can be accomplished by sim-
ply connecting a Schottky diode across the SW and V
IN
pins as shown in Figure 8. The diode will normally be
reverse biased unless V
IN
is pulled below V
OUT
at which
time the diode will clamp the (V
OUT
– V
IN
) potential to less
than the 0.6V required for latchup. Note that a low leakage
Schottky should be used to minimize the effect on no-load
supply current. Schottky diodes such as MBR0530, BAS85
and BAT84 work well. Another more serious effect of the
protection diode leakage is that at no load with nothing to
provide a sink for this leakage current, the output voltage
can potentially float above the maximum allowable toler-
ance. To prevent this from occuring, a resistor must be
connected between V
OUT
and ground with a value low
enough to sink the maximum possible leakage current.
Thermal Considerations
In the majority of the applications, the LTC1474/LTC1475
do not dissipate much heat due to their high efficiency.
However, in applications where the switching regulator is
running at high ambient temperature with low supply
voltage and high duty cycles, such as dropout with the
switch on continuously, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated by the regulator
exceeds the maximum junction temperature of the part.
The temperature rise is given by:
T
R
= P • θ
JA
where P is the power dissipated by the regulator and θ
JA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature is given by:
T
J
= T
A
+ T
R
As an example consider the LTC1474/LTC1475 in dropout
at an input voltage of 3.5V, a load current of 300mA, and
an ambient temperature of 70°C. From the typical perfor-
mance graph of switch resistance, the on-resistance of the
P-channel switch at 70°C is 3.5. Therefore, power dissi-
pated by the part is:
P = I
2
• R
DS(ON)
= 0.315W
For the MSOP package, the θ
JA
is 150°C/W. Thus the
junction temperature of the regulator is:
T
J
= 70°C + (0.315)(150) = 117°C
which is near the maximum junction temperature of 125
o
C.
Note that at higher supply voltages, the junction tempera-
ture is lower due to reduced switch resistance.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1474/LTC1475. These items are also illustrated
graphically in the layout diagram of Figure 9. Check the
following in your layout:
1. Is the Schottky diode cathode
closely
connected to SW
(Pin 5)?
2. Is the 0.1µF input decoupling capacitor
closely
con-
nected between V
IN
(Pin 7) and ground (Pin 4)? This
capacitor carries the high frequency peak currents.
3. When using adjustable version, is the resistive divider
closely connected to the (+) and (–) plates of C
OUT
with
a 10pF capacitor connected across R2?
4. Is the 1000pF decoupling capacitor for the current
sense resistor connected as close as possible to Pins 6
and 7? If no current sense resistor is used, Pins 6 and
7 should be shorted.
13
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUUU
Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist)
+
+
V
OUT
V
IN
V
IN
R
SENSE
V
FB
C
OUT
C
IN
10pF
1000pF
R2
100k
LTC1474
R1
D1
LBO
LBI
GND
RUN
SENSE
SW
L
1474/75 F09
8
7
6
5
1
2
3
4
BOLD LINES INDICATE HIGH PATH CURRENTS
0.1µF
OUTPUT DIVIDER REQUIRED WITH
ADJUSTABLE VERSION ONLY
5. Are the signal and power grounds segregated? The
signal ground consists of the (–) plate of C
OUT
, Pin 4 of
the LTC1474/LTC1475 and the resistive divider. The
power ground consists of the Schottky diode anode,
the (–) plate of C
IN
and the 0.1µF decoupling capacitor.
6. Is a 100k resistor connected in series between RUN
(Pin 8) and the RUN control voltage? The resistor
should be as close as possible to Pin 8.
Design Example (Refer to R
SENSE
and Inductor
Selection)
As a design example, assume V
IN
= 10V, V
OUT
= 3V, and
a maximum average output current I
MAX
= 100mA. With
this information, we can easily calculate all the important
components:
From the equation (1),
R
SENSE
= (0.067/0.1) – 0.25 = 0.42
Using the standard resistors (1, 1 and 2) in parallel
provides 0.4 without having to use a more expensive
low value current shunt type resistor (see R
SENSE
Selec-
tion section).
With R
SENSE
= 0.4, the peak inductor current I
PEAK
is
calculated from (2), neglecting the second term, to be
150mA. The minimum inductance is, therefore, from the
equation (3) and assuming V
D
= 0.4V,
LsH
MIN
=+
()()
=
07533 04 475
015 01 264
....
..
µµ
From Figure 3, an inductance of 270µH is chosen from the
recommended region. The CDRH73-271 or CD54-271 is a
good choice for space limited applications.
For the feedback resistors, choose R1 = 1M to minimize
supply current. R2 can then be calculated from the equa-
tion (4) to be:
RVRM
OUT
2123 11143=−
•=
..
For the catch diode, the MBR0530 will work well in this
application.
For the input and output capacitors, AVX 4.7µF and 100µF,
respectively, low ESR TPS series work well and meet the
RMS current requirement of 100mA/2 = 50mA. They are
available in small “C” case sizes with 0.15 ESR. The
0.15 output capacitor ESR will result in 25mV of output
voltage ripple.
Figure 10 shows the complete circuit for this example.
14
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
1474/75 F10
V
IN
GND
SENSE
LBI
RUN
V
FB
LBO
SW
LTC1474
+
6
3
8
1
2
5
7
4
100k
1.43M
1%
RUN
1000pF
4.7µF
35V 0.1µF
V
IN
3.5V TO 18V
1M
1%
2**1** 1**
10pF
L*
270µH
D1
MBR0530
+
100µF
††
6.3V
V
OUT
3V
100mA
* SUMIDA CDRH73-271
** 3 PARALLEL STANDARD RESISTORS
PROVIDE LEAST EXPENSIVE SOLUTION
(SEE R
SENSE
SELECTION SECTION)
AVX TPSC475M035
††
AVX TPSC107M006
Figure 10. High Efficiency 3V/100mA Regulator (Design Example)
Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop
1474/75 F11
V
IN
GND
SENSE
LBI
RUN
V
OUT
LBO
SW
LTC1474-3.3
6
3
8
1
2
5
7
4
100k
RUN
TO A/D
MBR0530
1000pF
1µF
× 3
IN
+
4mA TO 20mA
IN
4mA TO 20mA
2
7.5M
1M
D2
††
12V
L*
330µH
D1
MBR0530
10µF**
V
OUT
3.3V
10mA
* COILCRAFT DO1608-334
** MARCON THCS50E1E106Z,
AVX 1206ZG106Z
OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER
††
MOTOROLA MMBZ5242BL
15
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
1474/75 F12
V
IN
GND
SENSE
LBI
RUN
V
FB
LBO
SW
LTC1474
6
3
8
1
2
5
7
4
100k
4.7M
1%
RUN
22µF**
16V
V
IN
3.5V TO 6V
536k
1%
10pF
L*
200µH
L*
200µHD1
MBR0530
+
+
+
22µF
††
25V
22µF
††
25V
10µF
25V
+
V
OUT
12V
70mA
V
OUT
12V
70mA
MBR0530
* COILTRONICS CTX200-4
** AVX TPSC226M016
AVX TPSC106M025
††
AVX TPSD226M025
0.1µF
V
IN
(V) I
LOAD(MAX)
3.5 30mA
4 50mA
5 70mA
6 90mA
Figure 12. 5V to ±12V Regulator
Figure 13. 5V Buck-Boost Converter
1474/75 F13
V
IN
GND
SENSE
LBI
RUN
V
OUT
LBO
SW
LTC1474-5
6
3
8
1
2
5
7
4
100k
RUN
10µF**
25V
0.1µF
V
IN
3.5V TO 12V
L*
100µH
L*
100µHD1
MBR0530
+
+
+
33µF
10V
10µF**
25V
V
OUT
5V
200mA AT V
IN
= 10V
* COILTRONICS CTX100-4
** AVX TPSC106MO25
AVX TPSC336M010
V
IN
(V) I
LOAD(MAX)
3.5 70mA
4 95mA
5 125mA
8 180mA
10 200mA
12 225mA
16
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
Figure 14. Positive-to-Negative (–5V) Converter
1474/75 F14
V
IN
GND
SENSE
LBI
RUN
V
OUT
LBO
SW
LTC1474-5
6
3
8
1
2
5
7
4
10µF**
25V
0.1µF
V
IN
3.5V TO 12V
L*
100µH
D1
MBR0530
+
+
33µF
10V
V
OUT
–5V
140mA AT V
IN
= 5V
* SUMIDA CDRH74-101
** AVX TPSC106M025
AVX TPSC336M010
††
RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V
IN
V
IN
(V) I
LOAD(MAX)
3.5 100mA
5 140mA
8 190mA
12 240mA
10M
ON/OFF
††
TP0610
1474/75 F15
V
IN
GND
SENSE
LBI
RUN
V
FB
LBO
SW
LTC1474
6
3
8
1
2
5
7
4
100k
4.69M
CHARGER
ON/OFF
4.7µF**
35V
0.1µF
V
IN
8V TO 18V
1M
10pF
L*
100µH
D1
MBR0530
MBR0530
+
+
47µF
16V
V
OUT
4-NiCd
200mA
* SUMIDA CDRH73-101
** AVX TPSC475M035
AVX TPSD476M016
Figure 15. 4-NiCd Battery Charger
17
LTC1474/LTC1475
TYPICAL APPLICATIONS
U
Figure 17. Pushbutton On/Off 5V/250mA Regulator
Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off
1474/75 F17
V
IN
GND
SENSE
LBI/OFF
ON
V
OUT
LBO
SW
LTC1475-5
6
3
8
1
2
5
7
4
100k
ON
OFF
4.7µF**
35V
0.1µF
V
IN
5.7V TO 18V
3.65M
L*
100µH
D1
MBR0530
+
+
33µF
10V
V
OUT
5V
250mA
1M
* SUMIDA CDRH73-101
** AVX TPSC475M035
AVX TPSC336M010
1474/75 F18
V
IN
GND
SENSE
LBI/OFF
ON V
OUT
LBO
SW
LTC1475-3.3
6
3
81
2
5
7
4
100k
ON/OFF
4.7µF**
35V 0.1µF
0.1µF
µC
V
IN
4V TO 18V
V
CC
L*
100µH
D1
MBR0530
+
100µF
6.3V
V
OUT
3.3V
250mA
MMBD914LT1
2.2M
1M
+
* SUMIDA CDRH73-101
** AVX TPSC475M035
AVX TPSC107M006
Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout
1474/75 F16
V
IN
GND
SENSE
LBI
RUN
V
OUT
LBO
SW
LTC1474-3.3
6
3
8
1
2
5
7
4
100k
RUN
4.7µF
35V
0.1µF
V
IN
4V TO 18V
2.2M
1M
L*
100µH
D1
MBR0530
+
+
100µF
††
6.3V
V
OUT
3.3V
250mA
* SUMIDA CDRH73-101
AVX TPSC475M035
††
AVX TPSC107M006
18
LTC1474/LTC1475
PACKAGE DESCRIPTION
U
Dimensions in inches (millimeters) unless otherwise noted.
MS8 Package
8-Lead Plastic MSOP
(LTC DWG # 05-08-1660)
MSOP (MS8) 1197
* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006"
(
0.152mm
)
PER SIDE
0.021 ± 0.006
(0.53 ± 0.015)
0° – 6° TYP
SEATING
PLANE
0.007
(0.18)
0.040 ± 0.006
(1.02 ± 0.15)
0.012
(0.30)
REF
0.006 ± 0.004
(0.15 ± 0.102)
0.034 ± 0.004
(0.86 ± 0.102)
0.0256
(0.65)
TYP
12
34
0.192 ± 0.004
(4.88 ± 0.10)
8765
0.118 ± 0.004*
(3.00 ± 0.102)
0.118 ± 0.004**
(3.00 ± 0.102)
19
LTC1474/LTC1475
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
PACKAGE DESCRIPTION
U
Dimensions in inches (millimeters) unless otherwise noted.
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
0.406 – 1.270
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 0996
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
20
LTC1474/LTC1475
TYPICAL APPLICATIO
N
U
sn14745 14745fas LT/TP 0398 4K REV A • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1997
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
FAX: (408) 434-0507
TELEX: 499-3977
www.linear-tech.com
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LTC1096/LTC1098 Micropower Sampling 8-Bit Serial I/O A/D Converter I
Q
= 80µA Max
LT1121/LT1121-3.3/LT1121-5 150mA Low Dropout Regulator Linear Regulator, I
Q
= 30µA
LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Selectable I
PEAK
= 300mA or 600mA
LTC1265 1.2A High Efficiency Step-Down DC/DC Converter Burst Mode
Operation, Internal MOSFET
LT1375/LT1376 1.5A 500kHz Step-Down Switching Regulators 500kHz, Small Inductor, High
Efficiency Switchers, 1.5A Switch
LTC1440/LTC1441/LTC1442 Ultralow Power Comparator with Reference I
Q
= 2.8µA Max
LT1495/LT1496 1.5µA Precision Rail-to-Rail Op Amps I
Q
= 1.5µA Max
LT1521/LT1521-3/LT1521-3.3/ 300mA Low Dropout Regulator Linear Regulator, I
Q
= 12µA
LT1521-5
LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode
LT1634-1.25 Micropower Precision Shunt Reference I
Q(MIN)
= 10µA
Figure 19. Pushbutton On/Off with Low Battery Lockout
1474/75 F19
V
IN
GND
SENSE
LBI/OFF
ON
V
FB
LBO
SW
LTC1475
6
3
8
1
2
5
7
4
100k
1.02M
1%
OFF
ON
4.7µF**
35V
0.1µF
V
IN
3.5V to 18V
1M
1%
1.8M
10pF
L*
100µH
D1
MBR0530
+
+
100µF
6.3V
V
OUT
2.5V
250mA
1M
MMBT2N2222LT1
1M
* SUMIDA CDRH73-101
** AVX TPSC475M035
AVX TPSC107M006