0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00
40
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
1.8Vout
3.3Vout
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00
40
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
1.8Vout
3.3Vout
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
LMR10520 SIMPLE SWITCHER
®
5.5Vin, 2.0A Step-Down Voltage Regulator in WSON
Check for Samples: LMR10520
1FEATURES DESCRIPTION
The LMR10520 regulator is a monolithic, high
• Input Voltage Range of 3V to 5.5V frequency, PWM step-down DC/DC converter in a 6
• Output Voltage Range of 0.6V to 4.5V Pin WSON package. It provides all the active
• Output Current up to 2A functions to provide local DC/DC conversion with fast
transient response and accurate regulation in the
• 1.6MHz (LMR10520X) and 3 MHz (LMR10520Y) smallest possible PCB area. With a minimum of
Switching Frequencies external components, the LMR10520 is easy to use.
• Low Shutdown Iq, 30 nA Typical The ability to drive 2.0A loads with an internal 150
• Internal Soft-Start mΩPMOS switch results in the best power density
available. The world-class control circuitry allows on-
• Internally Compensated times as low as 30ns, thus supporting exceptionally
• Current-Mode PWM Operation high frequency conversion over the entire 3V to 5.5V
• Thermal Shutdown input operating range down to the minimum output
voltage of 0.6V. The LMR10520 is internally
• WSON-6 (3 x 3 x 0.8 mm) Packaging compensated, so it is simple to use and requires few
• Fully Enabled for WEBENCH® Power Designer external components. Even though the operating
frequency is high, efficiencies up to 93% are easy to
PERFORMANCE BENEFITS achieve. External shutdown is included, featuring an
ultra-low stand-by current of 30 nA. The LMR10520
• Extremely Easy to Use utilizes current-mode control and internal
• Tiny Overall Solution Reduces System Cost compensation to provide high-performance regulation
over a wide range of operating conditions. Additional
APPLICATIONS features include internal soft-start circuitry to reduce
inrush current, pulse-by-pulse current limit, thermal
• Point-of-Load Conversions from 3.3V, and 5V shutdown, and output over-voltage protection.
Rails
• Space Constrained Applications
• Battery Powered Equipment
• Industrial Distributed Power Applications
• Power Meters
• Portable Hand-Held Instruments
System Performance
Efficiency vs Load Current - "X" VIN = 5V Efficiency vs Load Current - "Y" VIN = 5V
Figure 1. Figure 2.
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date. Copyright © 2011–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
1
2
34
6
5
EN
FB
SW
DAP VINA
VIND
GND
GND
FB
EN
VIN SW
VIN
C1
R3
D1
L1
R2
R1
C2 C3
VOUT
LMR10520
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Typical Application
Connection Diagram
Figure 3. 6-Pin WSON Package
See Package Number NGG0006A
Table 1. Pin Descriptions 6-Pin WSON
Pin Name Function
1 FB Feedback pin. Connect to external resistor divider to set output voltage.
2 GND Signal and power ground pin. Place the bottom resistor of the feedback network as close as
possible to this pin.
3 SW Switch Node. Connect to the inductor and catch diode.
4 VIND Power Input supply.
5 VINA Control circuitry supply voltage. Connect VINA to VIND on PC board.
6 EN Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than
VINA + 0.3V.
DAP Die Attach Pad Connect to system ground for low thermal impedance, but it cannot be used as a primary GND
connection.
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SNVS730B –OCTOBER 2011–REVISED APRIL 2013
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
VIN -0.5V to 7V
FB Voltage -0.5V to 3V
EN Voltage -0.5V to 7V
SW Voltage -0.5V to 7V
ESD Susceptibility 2kV
Junction Temperature(3) 150°C
Storage Temperature −65°C to +150°C
Soldering Information
For soldering specifications: http://www.ti.com/lit/SNOA549
(1) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(2) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics.
(3) Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device.
Operating Ratings
VIN 3V to 5.5V
Junction Temperature −40°C to +125°C
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Electrical Characteristics(1)(2)
VIN = 5V unless otherwise indicated under the Conditions column. Limits in standard type are for TJ= 25°C only; limits in
boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are ensured
through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ= 25°C, and are
provided for reference purposes only.
Parameter Test Conditions Min Typ Max Units
VFB Feedback Voltage 0.588 0.600 0.612 V
ΔVFB/VIN Feedback Voltage Line Regulation VIN = 3V to 5V 0.02 %/V
IBFeedback Input Bias Current 0.1 100 nA
VIN Rising 2.73 2.90 V
Undervoltage Lockout
UVLO VIN Falling 1.85 2.3
UVLO Hysteresis 0.43 V
LMR10520-X 1.2 1.6 1.95
FSW Switching Frequency MHz
LMR10520-Y 2.25 3.0 3.75
LMR10520-X 86 94
DMAX Maximum Duty Cycle %
LMR10520-Y 82 90
LMR10520-X 5
DMIN Minimum Duty Cycle %
LMR10520-Y 7
RDS(ON) Switch On Resistance 150 mΩ
ICL Switch Current Limit VIN = 3.3V 2.4 3.25 A
Shutdown Threshold Voltage 0.4
VEN_TH V
Enable Threshold Voltage 1.8
ISW Switch Leakage 100 nA
IEN Enable Pin Current Sink/Source 100 nA
LMR10520X VFB = 0.55 3.3 5
IQQuiescent Current (switching) mA
LMR10520Y VFB = 0.55 4.3 6.5
Quiescent Current (shutdown) All Options VEN = 0V 30 nA
Junction to Ambient 80
θJA °C/W
0 LFPM Air Flow(3)
θJC Junction to Case 18 °C/W
TSD Thermal Shutdown Temperature 165 °C
(1) Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlation
using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely parametric norm.
(3) Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air.
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0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00
40
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
LMR10520X
LMR10520Y
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00
40
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
1.8Vout
3.3Vout
0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00
40
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
1.8Vout
3.3Vout
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
Typical Performance Characteristics
Unless stated otherwise, all curves taken at VIN = 5.0V with configuration in typical application circuit shown in Figure 22.
TJ= 25°C, unless otherwise specified.
ηvs Load "X" Vin = 5V, Vo = 1.8V & 3.3V ηvs Load "Y" Vin = 5V, Vo = 3.3V & 1.8V
Figure 4. Figure 5.
Load Regulation
ηvs Load "X,and Y" Vin = 3.3V, Vo = 1.8V Vin = 3.3V, Vo = 1.8V (All Options)
Figure 6. Figure 7.
Load Regulation Load Regulation
Vin = 5V, Vo = 1.8V (All Options) Vin = 5V, Vo = 3.3V (All Options)
Figure 8. Figure 9.
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Product Folder Links: LMR10520
-45 -40 -10 20 50 80 110 125 130
TEMPERATURE (ºC)
4.0
4.1
4.2
4.3
4.4
4.5
4.6
IQ (mA)
-45 -40 -10 20 50 80 110 125 130
TEMPERATURE (ºC)
3.0
3.1
3.2
3.3
3.4
3.5
3.6
IQ (mA)
-45 -40 -10 20 50 80 110 125 130
TEMPERATURE (oC)
CURRENT LIMIT (mA)
2800
2900
3000
3100
3200
3300
3400
3500
3600
3700
3800
-45 -40 -10 20 50 80 110 125 130
TEMPERATURE (ºC)
OSCILLATOR FREQUENCY (MHz)
2.55
2.65
2.75
2.85
2.95
3.05
3.15
3.25
3.35
3.45
-45 -40 -10 20 50 80 110 125 130
TEMPERATURE (ºC)
OSCILLATOR FREQUENCY (MHz)
1.36
1.41
1.46
1.51
1.56
1.61
1.66
1.71
1.76
1.81
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless stated otherwise, all curves taken at VIN = 5.0V with configuration in typical application circuit shown in Figure 22.
TJ= 25°C, unless otherwise specified.
Oscillator Frequency vs Temperature - "X" Oscillator Frequency vs Temperature - "Y"
Figure 10. Figure 11.
Current Limit vs Temperature
Vin = 3.3V RDSON vs Temperature
Figure 12. Figure 13.
LMR10520X IQ(Quiescent Current) LMR10520Y IQ(Quiescent Current)
Figure 14. Figure 15.
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SNVS730B –OCTOBER 2011–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless stated otherwise, all curves taken at VIN = 5.0V with configuration in typical application circuit shown in Figure 22.
TJ= 25°C, unless otherwise specified.
Line Regulation
Vo = 1.8V, Io = 500mA VFB vs Temperature
Figure 16. Figure 17.
Gain vs Frequency Phase Plot vs Frequency
(Vin = 5V, Vo = 1.2V @ 1A) (Vin = 5V, Vo = 1.2V @ 1A)
Figure 18. Figure 19.
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Product Folder Links: LMR10520
cv
+
-
+
-
S
R
R
Q
+
-
GND
FB
SW
VIN
EN
+
-
+
-
DRIVER
Artificial
Ramp
SHDN
Thermal
SHDN
OVP
1.6 MHz
CompInternal-
SENSE
I
LIMIT
I
LDOInternal-
STARTSOFT-
PFET
SENSE
I
ENABLE and UVLO
15.1 x REF
VControl Logic
VREF = 0.6V
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Simplified Block Diagram
Figure 20.
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0
0
VIN
VD
TON
t
t
Inductor
Current
D = TON/TSW
VSW
TOFF
TSW
IL
IPK
SW
Voltage
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
APPLICATIONS INFORMATION
Theory of Operation
The following operating description of the LMR10520 will refer to the SIMPLIFIED BLOCK DIAGRAM (Figure 20)
and to the waveforms in Figure 21. The LMR10520 supplies a regulated output voltage by switching the internal
PMOS control switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge
of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on
the internal PMOS control switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN,
and the inductor current (IL) increases with a linear slope. ILis measured by the current sense amplifier, which
generates an output proportional to the switch current. The sense signal is summed with the regulator’s
corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the
feedback voltage and VREF. When the PWM comparator output goes high, the output switch turns off until the
next switching cycle begins. During the switch off-time, inductor current discharges through the Schottky catch
diode, which forces the SW pin to swing below ground by the forward voltage (VD) of the Schottky catch diode.
The regulator loop adjusts the duty cycle (D) to maintain a constant output voltage.
Figure 21. Typical Waveforms
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FB
GND
SW
EN VOUT
L1
D1
R1
R2
VINA/VIND
C1 C4
U1
C2
GND
EN
2
6
4, 5
1
3
R3
Chf
2.2 PF
VIN
C3
GND
22 PF
20k
10k 2.2 PF22 PF22 nF
(opt.)
1.0 PH
3.3 PH
(³;´YHUVLRQ)
1.8V
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Soft-Start
This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s
reference voltage ramps from 0V to its nominal value of 0.6V in approximately 600 µs. This forces the regulator
output to ramp up in a controlled fashion, which helps reduce inrush current.
Output Overvoltage Protection
The over-voltage comparator compares the FB pin voltage to a voltage that is 15% higher than the internal
reference VREF. Once the FB pin voltage goes 15% above the internal reference, the internal PMOS control
switch is turned off, which allows the output voltage to decrease toward regulation.
Undervoltage Lockout
Under-voltage lockout (UVLO) prevents the LMR10520 from operating until the input voltage exceeds 2.73V
(typ). The UVLO threshold has approximately 430 mV of hysteresis, so the part will operate until VIN drops below
2.3V (typ). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
Current Limit
The LMR10520 uses cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a
current limit comparator detects if the output switch current exceeds 3.25A (typ), and turns off the switch until the
next switching cycle begins.
Thermal Shutdown
Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature
exceeds 165°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature
drops to approximately 150°C.
Figure 22. Typical Application Schematic
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VIN - VOUT
L=2'iL
DTS
t
L
i'
OUT
I
S
T
S
DT
L
VOUT
L
- VOUT
VIN
D = VOUT + VD
VIN + VD - VSW
D =VOUT
VIN
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
Design Guide
Inductor Selection
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN):
(1)
The catch diode (D1) forward voltage drop and the voltage drop across the internal PMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
(2)
VSW can be approximated by:
VSW = IOUT x RDSON (3)
The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower the
VD, the higher the operating efficiency of the converter. The inductor value determines the output ripple current.
Lower inductor values decrease the size of the inductor, but increase the output ripple current. An increase in the
inductor value will decrease the output ripple current.
One must ensure that the minimum current limit (2.4A) is not exceeded, so the peak current in the inductor must
be calculated. The peak current (ILPK) in the inductor is calculated by:
ILPK = IOUT +ΔiL(4)
Figure 23. Inductor Current
(5)
In general,
ΔiL= 0.1 x (IOUT)→0.2 x (IOUT) (6)
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IRMS_IN = IOUT x D(1 - D)
IRMS_IN D IOUT2 (1-D) + 'i2
3
TS = 1
fS
x (VIN - VOUT)
L = 2'iL
DTS
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
If ΔiL= 20% of 2A, the peak current in the inductor will be 2.4A. The minimum ensured current limit over all
operating conditions is 2.4A. One can either reduce ΔiL, or make the engineering judgment that zero margin will
be safe enough. The typical current limit is 3.25A.
The LMR10520 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the Output Capacitor section for more details on calculating output voltage ripple. Now that the ripple current
is determined, the inductance is calculated by:
(7)
Where
(8)
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating
correctly. Because of the speed of the internal current limit, the peak current of the inductor need only be
specified for the required maximum output current. For example, if the designed maximum output current is 1.0A
and the peak current is 1.25A, then the inductor should be specified with a saturation current limit of > 1.25A.
There is no need to specify the saturation or peak current of the inductor at the 3.25A typical switch current limit.
The difference in inductor size is a factor of 5. Because of the operating frequency of the LMR10520, ferrite
based inductors are preferred to minimize core losses. This presents little restriction since the variety of ferrite-
based inductors is huge. Lastly, inductors with lower series resistance (RDCR) will provide better operating
efficiency. For recommended inductors, see Example Circuits.
Input Capacitor
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent
Series Inductance). The recommended input capacitance is 22 µF. The input voltage rating is specifically stated
by the capacitor manufacturer. Make sure to check any recommended deratings and also verify if there is any
significant change in capacitance at the operating input voltage and the operating temperature. The input
capacitor maximum RMS input current rating (IRMS-IN) must be greater than:
(9)
Neglecting inductor ripple simplifies the above equation to:
(10)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle D is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL
and a 0805 ceramic chip capacitor will have very low ESL. At the operating frequencies of the LMR10520,
leaded capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than that required
to provide stable operation. As a result, surface mount capacitors are strongly recommended.
Sanyo POSCAP, Tantalum or Niobium, Panasonic SP, and multilayer ceramic capacitors (MLCC) are all good
choices for both input and output capacitors and have very low ESL. For MLCCs it is recommended to use X7R
or X5R type capacitors due to their tolerance and temperature characteristics. Consult capacitor manufacturer
datasheets to see how rated capacitance varies over operating conditions.
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x R2
R1 = VREF
VOUT - 1
'VOUT = 'ILRESR + 8 x FSW x COUT
1
LMR10520
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SNVS730B –OCTOBER 2011–REVISED APRIL 2013
Output Capacitor
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is:
(11)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple will be approximately sinusoidal and 90° phase shifted from the switching action. Given the
availability and quality of MLCCs and the expected output voltage of designs using the LMR10520, there is really
no need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to
bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic
capacitances in the inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not.
Since the output capacitor is one of the two external components that control the stability of the regulator control
loop, most applications will require a minimum of 22 µF of output capacitance. Capacitance often, but not always,
can be increased significantly with little detriment to the regulator stability. Like the input capacitor, recommended
multilayer ceramic capacitors are X7R or X5R types.
Catch Diode
The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching
times and low forward voltage drop. The catch diode should be chosen so that its current rating is greater than:
ID1 = IOUT x (1-D) (12)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
Output Voltage
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VOand the FB pin. A good value for R2 is 10kΩ. When designing a unity gain
converter (Vo = 0.6V), R1 should be between 0Ωand 100Ω, and R2 should be equal or greater than 10kΩ.
(13)
VREF = 0.60V (14)
PCB Layout Considerations
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration is the close coupling of the GND connections of the input capacitor and the catch
diode D1. These ground ends should be close to one another and be connected to the GND plane with at least
two through-holes. Place these components as close to the IC as possible. Next in importance is the location of
the GND connection of the output capacitor, which should be near the GND connections of CIN and D1. There
should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node
island. The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise
pickup and inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with
the GND of R1 placed as close as possible to the GND of the IC. The VOUT trace to R2 should be routed away
from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW and VOUT
traces, so they should be as short and wide as possible. However, making the traces wide increases radiated
noise, so the designer must make this trade-off. Radiated noise can be decreased by choosing a shielded
inductor. The remaining components should also be placed as close as possible to the IC. Please see
Application Note AN-1229 SNVA054 for further considerations and the LMR10520 demo board as an example of
a good layout.
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PCOND= (IOUT2 x D) 1
3
1 + x'iL
IOUT
2RDSON
D = VOUT + VD + VDCR
VIN + VD + VDCR - VSW
D = VOUT + VD
VIN + VD - VSW
K = POUT
POUT + PLOSS
K =POUT
PIN
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Calculating Efficiency, and Junction Temperature
The complete LMR10520 DC/DC converter efficiency can be calculated in the following manner.
(15)
Or
(16)
Calculations for determining the most significant power losses are shown below. Other losses totaling less than
2% are not discussed.
Power loss (PLOSS) is the sum of two basic types of losses in the converter: switching and conduction.
Conduction losses usually dominate at higher output loads, whereas switching losses remain relatively fixed and
dominate at lower output loads. The first step in determining the losses is to calculate the duty cycle (D):
(17)
VSW is the voltage drop across the internal PFET when it is on, and is equal to:
VSW = IOUT x RDSON (18)
VDis the forward voltage drop across the Schottky catch diode. It can be obtained from the diode manufactures
Electrical Characteristics section. If the voltage drop across the inductor (VDCR) is accounted for, the equation
becomes:
(19)
The conduction losses in the free-wheeling Schottky diode are calculated as follows:
PDIODE = VDx IOUT x (1-D) (20)
Often this is the single most significant power loss in the circuit. Care should be taken to choose a Schottky
diode that has a low forward voltage drop.
Another significant external power loss is the conduction loss in the output inductor. The equation can be
simplified to:
PIND = IOUT2x RDCR (21)
The LMR10520 conduction loss is mainly associated with the internal PFET:
(22)
If the inductor ripple current is fairly small, the conduction losses can be simplified to:
PCOND = IOUT2x RDSON x D (23)
Switching losses are also associated with the internal PFET. They occur during the switch on and off transition
periods, where voltages and currents overlap resulting in power loss. The simplest means to determine this loss
is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node.
Switching Power Loss is calculated as follows:
PSWR = 1/2(VIN x IOUT x FSW x TRISE) (24)
PSWF = 1/2(VIN x IOUT x FSW x TFALL) (25)
PSW = PSWR + PSWF (26)
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SNVS730B –OCTOBER 2011–REVISED APRIL 2013
Another loss is the power required for operation of the internal circuitry:
PQ= IQx VIN (27)
IQis the quiescent operating current, and is typically around 3.3mA for the 1.6MHz frequency option.
Typical Application power losses are:
Table 2. Power Loss Tabulation
VIN 5.0V
VOUT 3.3V POUT 5.78W
IOUT 1.75A
VD0.45V PDIODE 262mW
FSW 1.6MHz
IQ3.3mA PQ16.5mW
TRISE 4nS PSWR 28mW
TFALL 4nS PSWF 28mW
RDS(ON) 150mΩPCOND 306mW
INDDCR 50mΩPIND 153mW
D 0.667 PLOSS 794mW
η88% PINTERNAL 379mW
ΣPCOND + PSW + PDIODE + PIND + PQ= PLOSS (28)
ΣPCOND + PSWF + PSWR + PQ= PINTERNAL (29)
PINTERNAL = 379mW (30)
Copyright © 2011–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LMR10520
RTJC=TJ - TC
Power
RTJA=TJ - TA
Power
RT='T
Power
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Thermal Definitions
TJ= Chip junction temperature
TA= Ambient temperature
RθJC = Thermal resistance from chip junction to device case
RθJA = Thermal resistance from chip junction to ambient air
Heat in the LMR10520 due to internal power dissipation is removed through conduction and/or convection.
Conduction: Heat transfer occurs through cross sectional areas of material. Depending on the material, the
transfer of heat can be considered to have poor to good thermal conductivity properties (insulator vs. conductor).
Heat Transfer goes as:
Silicon →package →lead frame →PCB
Convection: Heat transfer is by means of airflow. This could be from a fan or natural convection. Natural
convection occurs when air currents rise from the hot device to cooler air.
Thermal impedance is defined as:
(31)
Thermal impedance from the silicon junction to the ambient air is defined as:
(32)
The PCB size, weight of copper used to route traces and ground plane, and number of layers within the PCB can
greatly effect RθJA. The type and number of thermal vias can also make a large difference in the thermal
impedance. Thermal vias are necessary in most applications. They conduct heat from the surface of the PCB to
the ground plane. Four to six thermal vias should be placed under the exposed pad to the ground plane.
Thermal impedance also depends on the thermal properties of the application operating conditions (Vin, Vo, Io
etc), and the surrounding circuitry.
Silicon Junction Temperature Determination Method 1:
To accurately measure the silicon temperature for a given application, two methods can be used. The first
method requires the user to know the thermal impedance of the silicon junction to case temperature.
RθJC is approximately 18°C/Watt for the 6-pin WSON package with the exposed pad. Knowing the internal
dissipation from the efficiency calculation given previously, and the case temperature, which can be empirically
measured on the bench we have:
where
• TCis the temperature of the exposed pad and can be measured on the bottom side of the PCB. (33)
Therefore:
Tj= (RθJC x PLOSS) + TC(34)
From the previous example:
Tj= (RθJC x PINTERNAL) + TC(35)
Tj= 18°C/W x 0.213W + TC(36)
The second method can give a very accurate silicon junction temperature.
16 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LMR10520
RTJA=165°C - 120°C
379 mW = 119°C/W
RTJA=165° - Ta
PINTERNAL
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
The first step is to determine RθJA of the application. The LMR10520 has over-temperature protection circuitry.
When the silicon temperature reaches 165°C, the device stops switching. The protection circuitry has a
hysteresis of about 15°C. Once the silicon temperature has decreased to approximately 150°C, the device will
start to switch again. Knowing this, the RθJA for any application can be characterized during the early stages of
the design one may calculate the RθJA by placing the PCB circuit into a thermal chamber. Raise the ambient
temperature in the given working application until the circuit enters thermal shutdown. If the SW-pin is monitored,
it will be obvious when the internal PFET stops switching, indicating a junction temperature of 165°C. Knowing
the internal power dissipation from the above methods, the junction temperature, and the ambient temperature
RθJA can be determined.
(37)
Once this is determined, the maximum ambient temperature allowed for a desired junction temperature can be
found.
An example of calculating RθJA for an application using the LMR10520 is shown below.
A sample PCB is placed in an oven with no forced airflow. The ambient temperature was raised to 120°C, and at
that temperature, the device went into thermal shutdown.
From the previous example:
PINTERNAL = 379 mW (38)
(39)
Since the junction temperature must be kept below 125°C, then the maximum ambient temperature can be
calculated as:
Tj- (RθJA x PLOSS) = TA(40)
125°C - (119°C/W x 379 mW) = 80°C (41)
WSON Package
Figure 24. Internal WSON Connection
For certain high power applications, the PCB land may be modified to a "dog bone" shape (see Figure 25). By
increasing the size of ground plane, and adding thermal vias, the RθJA for the application can be reduced.
Copyright © 2011–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LMR10520
GND
FB
EN
VIN SW
VIN = 5V
C1
R3
D1
L1
LMR10520
R2
R1
C2
VO = 3.3V @ 2.0A
100k
22 PF
10V
2.2 PH
2.8A
2A
20V
45.3k
2 x 22 PF
6.3V
10k
GND
FB
EN
VIN SW
VIN = 5V
C1
R3
D1
L1
LMR10520
R2
R1
C2
VO = 1.2V @ 2.0A
100k
22 PF
10V
2.2 PH
3.5A
2A
20V
15k
15k 2 x 22 PF
6.3V
1
2
4
6
5
EN
FB
SW
VINA
VIND
GND GND
PLANE
3
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Figure 25. 6-Lead WSON PCB Dog Bone Layout
LMR10520X Design Example 1
Figure 26. LMR10520X (1.6MHz): Vin = 5V, Vo = 1.2V at 2.0A
LMR10520X Design Example 2
Figure 27. LMR10520X (1.6MHz): Vin = 5V, Vo = 3.3V at 2.0A
18 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LMR10520
GND
FB
EN
VIN SW
VIN = 5V
C1
R3
D1
L1
LMR10520
R2
R1
C2
VO = 1.2V @ 2.0A
100k
22 PF
10V
4.7 PH
2.7A
2A
20V
10k
2 x 22 PF
6.3V
10k
GND
FB
EN
VIN SW
VIN = 5V
C1
R3
D1
L1
LMR10520
R2
R1
C2
VO = 3.3V @ 2.0A
100k
22 PF
10V
3.3 PH
3.3A
2A
20V
45.3k
2 x 22 PF
6.3V
10k
LMR10520
www.ti.com
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
LMR10520Y Design Example 3
Figure 28. LMR10520Y (3MHz): Vin = 5V, Vo = 3.3V at 2.0A
LMR10520Y Design Example 4
Figure 29. LMR10520Y (3MHz): Vin = 5V, Vo = 1.2V at 2.0A
Copyright © 2011–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LMR10520
LMR10520
SNVS730B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
REVISION HISTORY
Changes from Revision A (April 2013) to Revision B Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 19
20 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LMR10520
PACKAGE OPTION ADDENDUM
www.ti.com 11-Apr-2013
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish MSL Peak Temp
(3)
Op Temp (°C) Top-Side Markings
(4)
Samples
LMR10520XSD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L266B
LMR10520XSDE/NOPB ACTIVE WSON NGG 6 250 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L266B
LMR10520XSDX/NOPB ACTIVE WSON NGG 6 4500 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L266B
LMR10520YSD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L267B
LMR10520YSDE/NOPB ACTIVE WSON NGG 6 250 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L267B
LMR10520YSDX/NOPB ACTIVE WSON NGG 6 4500 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L267B
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
PACKAGE OPTION ADDENDUM
www.ti.com 11-Apr-2013
Addendum-Page 2
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LMR10520XSD/NOPB WSON NGG 6 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
LMR10520XSDE/NOPB WSON NGG 6 250 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
LMR10520XSDX/NOPB WSON NGG 6 4500 330.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
LMR10520YSD/NOPB WSON NGG 6 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
LMR10520YSDE/NOPB WSON NGG 6 250 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
LMR10520YSDX/NOPB WSON NGG 6 4500 330.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Sep-2016
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LMR10520XSD/NOPB WSON NGG 6 1000 210.0 185.0 35.0
LMR10520XSDE/NOPB WSON NGG 6 250 210.0 185.0 35.0
LMR10520XSDX/NOPB WSON NGG 6 4500 367.0 367.0 35.0
LMR10520YSD/NOPB WSON NGG 6 1000 210.0 185.0 35.0
LMR10520YSDE/NOPB WSON NGG 6 250 210.0 185.0 35.0
LMR10520YSDX/NOPB WSON NGG 6 4500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Sep-2016
Pack Materials-Page 2
MECHANICAL DATA
NGG0006A
www.ti.com
SDE06A (Rev A)
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