0 20 40 60 80 100 120 140 160
LOAD (mA)
0
10
20
30
40
50
60
70
80
EFFICIENCY (%)
EFFICIENCY (%)
LOAD CURRENT (mA)
0100 200 300 400
70
80
90
100
500
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
LMR62014 SIMPLE SWITCHER
®
20Vout, 1.4A Step-Up Voltage Regulator in SOT-23
Check for Samples: LMR62014
1FEATURES DESCRIPTION
The LMR62014 switching regulator is a current-mode
23• Input Voltage Range of 2.7V to 14V boost converter operating at fixed frequency of
• Output Voltage up to 20V 1.6 MHz.
• Switch Current up to 1.4A The use of SOT-23 package, made possible by the
• 1.6 MHz Switching Frequency minimal power loss of the internal 1.4A switch, and
• Low Shutdown Iq, <1 µA use of small inductors and capacitors result in the
industry's highest power density. The LMR62014 is
• Cycle-by-Cycle Current Limiting capable of greater than 90% duty cycle, making it
• Internally Compensated ideal for boosting to voltages up to 20V.
• 5-Pin SOT-23 Packaging (2.92 x 2.84 x 1.08mm) These parts have a logic-level shutdown pin that can
• Fully Enabled for WEBENCH®Power Designer be used to reduce quiescent current and extend
battery life.
PERFORMANCE BENEFITS Protection is provided through cycle-by-cycle current
• Extremely Easy to Use limiting and thermal shutdown. Internal compensation
simplifies design and reduces component count.
• Tiny Overall Solution Reduces System Cost
APPLICATIONS
• Boost Conversions from 3.3V, 5V, and 12V
Rails
• Space Constrained Applications
• Embedded Systems
• LCD Displays
• LED Applications
System Performance
Efficiency vs Load Current Efficiency vs Load Current
VIN = 3.3V, VOUT = 12V VIN = 5V, VOUT = 12V
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2SIMPLE SWITCHER, WEBENCH are registered trademarks of Texas Instruments.
3All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2011–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
LMR62014
SW
FB
GND
VIN
SHDN
U1
R3
51k
SHDN
GND
5VIN
C1
2.2 PF
L1/10 PH
R2
13.3k CF
220 pF
D1
R1/117k
C2
4.7 PF
12V
OUT
500 mA
(TYP)
LMR62014
SNVS735B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Connection Diagram
Figure 1. 5-Lead SOT-23 (Top View)
See DBV Package
PIN DESCRIPTIONS
Pin Name Function
1 SW Drain of the internal FET switch.
2 GND Analog and power ground.
3 FB Feedback point that connects to external resistive divider.
4 SHDN Shutdown control input. Connect to Vin if the feature is not used.
5 VIN Analog and power input.
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formula: If power dissipation exceeds the maximum specified above, the internal thermal protection
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
Storage Temperature Range −65°C to +150°C
Operating Junction Temperature Range −40°C to +125°C
Lead Temp. (Soldering, 5 sec.) 300°C
Power Dissipation(3) Internally Limited
FB Pin Voltage −0.4V to +6V
SW Pin Voltage −0.4V to +22V
Input Supply Voltage −0.4V to +14.5V
SHDN Pin Voltage −0.4V to VIN + 0.3V
θJ-A (SOT-23) 265°C/W
ESD Rating Human Body Model(4) 2 kV
For soldering specifications see SNOA549
(1) Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply
when operating the device outside of the limits set forth under the operating ratings which specify the intended range of operating
conditions.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(3) The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature,
TJ(MAX) = 125°C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature,
TA. The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the
circuitry will protect the device by reducing the output voltage as required to maintain a safe junction temperature.
(4) The human body model is a 100 pF capacitor discharged through a 1.5 kΩresistor into each pin.
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Electrical Characteristics
Limits in standard typeface are for TJ= 25°C, and limits in boldface type apply over the full operating temperature range
(−40°C ≤TJ≤+125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL= 0A.
Symbol Parameter Conditions Min(1) Typical(2) Max(1) Units
VIN Input Voltage 2.7 14 V
VOUT (MIN) Minimum Output Voltage RL= 43Ω(3) VIN = 2.7V 5.4 7 V
Under Load VIN = 3.3V 810
VIN = 5V 13 17
RL= 15Ω(3) VIN = 2.7V 3.75 5
VIN = 3.3V 56.5
VIN = 5V 8.75 11
ISW Switch Current Limit See (4) 1.8 2 A
1.4
RDS(ON) Switch ON Resistance ISW = 100 mA, Vin = 5V 260 400 mΩ
500
ISW = 100 mA, Vin = 3.3V 300 450
550
SHDNTH Shutdown Threshold Device ON 1.5 V
Device OFF 0.50
ISHDN Shutdown Pin Bias Current VSHDN = 0 0 µA
VSHDN = 5V 0 2
VFB Feedback Pin Reference VIN = 3V 1.205 1.230 1.255 V
Voltage
IFB Feedback Pin Bias Current VFB = 1.23V 60 500 nA
IQQuiescent Current VSHDN = 5V, Switching 2 3.0 mA
VSHDN = 5V, Not Switching 400 500 µA
VSHDN = 0 0.024 1
ΔVFB FB Voltage Line Regulation 2.7V ≤VIN ≤14V 0.02 %/V
ΔVIN
FSW Switching Frequency(5) 11.6 1.85 MHz
DMAX Maximum Duty Cycle(5) 86 93 %
ILSwitch Leakage Not Switching VSW = 5V 1 µA
(1) Limits are ensured by testing, statistical correlation, or design.
(2) Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most
likely expected value of the parameter at room temperature.
(3) L = 10 µH, COUT = 4.7 µF, duty cycle = maximum
(4) Switch current limit is dependent on duty cycle (see Typical Performance Characteristics).
(5) Specified limits are the same for Vin = 3.3V input.
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FEEDBACK VOLTAGE (V)
1.222
1.223
1.224
1.225
1.226
1.227
1.228
1.229
1.23
1.231
-40 -25 025 50 75 100 125
TEMPERATURE (oC)
TEMPERATURE (oC)
FEEDBACK BIAS CURRENT (PA)
0
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
0.09
-50 -25 025 50 75 100 125 150
MAX DUTY CYCLE (%)
92.1
92.2
92.3
92.4
92.5
92.6
92.7
92.8
92.9
93
TEMPERATURE (oC)
VIN = 5V
VIN = 3.3V
-50 -25 025 50 75 100 125 150
IQ VIN (IDLE) (PA)
TEMPERATURE (oC)
340
345
350
355
360
365
370
375
380
-50 -25 025 50 75 100 125 150
-50 -25 0 25 50 75 100 125 150
TEMPERATURE (oC)
1.8
1.85
1.9
1.95
2
2.05
2.1
2.15
2.2
IQ VIN ACTIVE (mA)
OSCILLATOR FREQUENCY (MHz)
1.4
1.42
1.44
1.46
1.48
1.5
1.52
1.54
1.56
1.58
TEMPERATURE (oC)
VIN = 5V
VIN = 3.3V
-50 -25 025 50 75 100 125 150
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
Typical Performance Characteristics
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Iq Vin (Active) vs Temperature Oscillator Frequency vs Temperature
Figure 2. Figure 3.
Max. Duty Cycle vs Temperature Iq Vin (Idle) vs Temperature
Figure 4. Figure 5.
Feedback Bias Current vs Temperature Feedback Voltage vs Temperature
Figure 6. Figure 7.
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70
0
0100 200 300 400 500 600
LOAD (mA)
EFFICIENCY (%)
0
10
20
30
40
50
60
70
80
90
100
1400
0200 400 600 800 1000 1200
LOAD (mA)
EFFICIENCY (%)
0
10
20
30
40
50
60
70
80
90
100
2.5 3.5 4.5 5.5 6.5 7.5 8.5 9.5
VIN (V)
0
50
100
150
200
250
300
350
RDS_ON (m:)
050 100 150 200 250 300
LOAD (mA)
EFFICIENCY (%)
0
10
20
30
40
50
60
70
80
90
100
RDS(ON) (:)
-40 -25 025 50 75 100 125
TEMPERATURE (oC)
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
Vin = 5V
Vin = 3.3V
CURRENT LIMIT (A)
2
2.1
2.2
2.3
2.4
2.5
2.6
-40 -25 025 50 75 100 125
TEMPERATURE (oC)
LMR62014
SNVS735B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
RDS(ON) vs Temperature Current Limit vs Temperature
Figure 8. Figure 9.
Efficiency vs Load Current
RDS(ON) vs VIN VIN = 2.7V, VOUT = 5V
Figure 10. Figure 11.
Efficiency vs Load Current Efficiency vs Load Current
VIN = 3.3V, VOUT = 5V VIN = 4.2V, VOUT = 5V
Figure 12. Figure 13.
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350
050 100 150 200 250 300
LOAD (mA)
EFFICIENCY (%)
0
10
20
30
40
50
60
70
80
90
100
600
0100 200 300 400 500
LOAD (mA)
EFFICIENCY (%)
0
10
20
30
40
50
60
70
80
90
100
0 20 40 60 80 100 120 140 160
LOAD (mA)
0
10
20
30
40
50
60
70
80
EFFICIENCY (%)
0 10 20 30 40 50
0
10
20
30
40
50
60
70
80
EFFICIENCY (%)
LOAD (mA)
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Efficiency vs Load Current Efficiency vs Load Current
VIN = 2.7V, VOUT = 12V VIN = 3.3V, VOUT = 12V
Figure 14. Figure 15.
Efficiency vs Load Current Efficiency vs Load Current
VIN = 5V, VOUT = 12V VIN = 5V, VOUT = 18V
Figure 16. Figure 17.
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LMR62014
www.ti.com
SNVS735B –OCTOBER 2011–REVISED APRIL 2013
THEORY OF OPERATION
The LMR62014 is a switching converter IC that operates at a fixed frequency (1.6 MHz) for fast transient
response over a wide input voltage range and incorporates pulse-by-pulse current limiting protection. Because
this is current mode control, a 33 mΩsense resistor in series with the switch FET is used to provide a voltage
(which is proportional to the FET current) to both the input of the pulse width modulation (PWM) comparator and
the current limit amplifier.
At the beginning of each cycle, the S-R latch turns on the FET. As the current through the FET increases, a
voltage (proportional to this current) is summed with the ramp coming from the ramp generator and then fed into
the input of the PWM comparator. When this voltage exceeds the voltage on the other input (coming from the
Gm amplifier), the latch resets and turns the FET off. Since the signal coming from the Gm amplifier is derived
from the feedback (which samples the voltage at the output), the action of the PWM comparator constantly sets
the correct peak current through the FET to keep the output voltage in regulation.
Q1 and Q2 along with R3 - R6 form a bandgap voltage reference used by the IC to hold the output in regulation.
The currents flowing through Q1 and Q2 will be equal, and the feedback loop will adjust the regulated output to
maintain this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at
the FB node "multiplied up" by the ratio of the output resistive divider.
The current limit comparator feeds directly into the flip-flop that drives the switch FET. If the FET current reaches
the limit threshold, the FET is turned off and the cycle terminated until the next clock pulse. The current limit
input terminates the pulse regardless of the status of the output of the PWM comparator.
Application Hints
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LMR62014 are multi-layer ceramic capacitors. They have the lowest ESR
(equivalent series resistance) and highest resonance frequency which makes them optimum for use with high
frequency switching converters.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as
Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage,
they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor
manufacturer’s data curves before selecting a capacitor.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will provide sufficient output capacitance for most
applications. If larger amounts of capacitance are desired for improved line support and transient response,
tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used,
but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies
above 500 kHz due to significant ringing and temperature rise due to self-heating from ripple current. An output
capacitor with excessive ESR can also reduce phase margin and cause instability.
In general, if electrolytics are used, it is recommended that they be paralleled with ceramic capacitors to reduce
ringing, switching losses, and output voltage ripple.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each
time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We
recommend a nominal value of 2.2 µF, but larger values can be used. Since this capacitor reduces the amount of
voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other
circuitry.
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
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FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor Cf is required for stability (see Basic Application
Circuit). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for
the zero fz should be approximately 6 kHz. Cf can be calculated using the formula:
Cf = 1 / (2 X πX R1 X fz) (1)
SELECTING DIODES
The external diode used in the typical application should be a Schottky diode.The diode must be rated to handle
the maximum output voltage and load current. A 20V diode such as the MBR0520 is recommended.
The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications
exceeding 0.5A average, a Toshiba CRS08 can be used.
LAYOUT HINTS
High frequency switching regulators require very careful layout of components in order to get stable operation
and low noise. All components must be as close as possible to the LMR62014 device. It is recommended that a
4-layer PCB be used so that internal ground planes are available.
As an example, a recommended layout of components is shown:
Figure 18. Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short. Parasitic trace inductance in series with D1 and C2
will increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept close to the FB pin of U1 to prevent noise injection
on the FB pin trace.
3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1,
as well as the negative sides of capacitors C1 and C2.
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Duty Cycle = VOUT + VDIODE - VIN
VOUT + VDIODE - VSW
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and R2 (see Basic Application Circuit). A value of
approximately 13.3 kΩis recommended for R2 to establish a divider current of approximately 92 µA. R1 is
calculated using the formula:
R1 = R2 X (VOUT/1.23 −1) (2)
Figure 19. Basic Application Circuit
DUTY CYCLE
The maximum duty cycle of the switching regulator determines the maximum boost ratio of output-to-input
voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost
application is defined as:
(3)
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor?” (because they are the largest
sized component and usually the most costly). The answer is not simple and involves trade-offs in performance.
Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given
size of output capacitor). Larger inductors also mean more load power can be delivered because the energy
stored during each switching cycle is:
E = L/2 X (lp)2
where
• “lp” is the peak inductor current. (4)
An important point to observe is that the LMR62014 will limit its switch current based on peak current. This
means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.
Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current
range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as
not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift
over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous”
over a wider load current range.
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To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V (5)
Since the frequency is 1.6 MHz (nominal), the period is approximately 0.625 µs. The duty cycle will be 62.5%,
which means the ON time of the switch is 0.390 µs. It should be noted that when the switch is ON, the voltage
across the inductor is approximately 4.5V.
Using the equation:
V = L (di/dt) (6)
We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON time. Using
these facts, we can then show what the inductor current will look like during operation:
Figure 20. 10 µH Inductor Current,
5V–12V Boost (LMR62014X)
During the 0.390 µs ON time, the inductor current ramps up 0.176A and ramps down an equal amount during the
OFF time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to
about 33 mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode.
A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and
continuous operation will be maintained at the typical load current values.
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ILOAD(max) = (1 - DC) x (ISW(max) - DC (VIN - VSW))
2fL
20 30 40 50 60 70 80 90 100
DUTY CYCLE (%) = [1 - EFF*(VIN / VOUT)]
0
500
1000
1500
2000
2500
3000
SW CURRENT LIMIT (mA)
VIN = 5V
VIN = 3.3V
VIN = 2.7V
VIN = 3V
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the
application. This is illustrated in the graphs below which show typical values of switch current as a function of
effective (actual) duty cycle:
Figure 21. Switch Current Limit vs Duty Cycle
CALCULATING LOAD CURRENT
As shown in Figure 20 which depicts inductor current, the load current is related to the average inductor current
by the relation:
ILOAD = IIND(AVG) x (1 - DC)
where
• "DC" is the duty cycle of the application. (7)
The switch current can be found by:
ISW = IIND(AVG) + ½ (IRIPPLE) (8)
Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L) (9)
combining all terms, we can develop an expression which allows the maximum available load current to be
calculated:
(10)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode. For actual load current in typical applications, we took bench data for
various input and output voltages that displayed the maximum load current available for a typical device in graph
form:
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VIN (V)
MAX LOAD CURRENT (mA)
0
200
400
600
800
1000
1200
2 3 4 5 6 7 8 9 10 11
VOUT = 5V
VOUT = 8V
VOUT = 10V
VOUT = 12V
VOUT = 18V
LMR62014
SNVS735B –OCTOBER 2011–REVISED APRIL 2013
www.ti.com
Figure 22. Max. Load Current (typ) vs VIN
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the equations) is dependent on load current. A good
approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor
current.
FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input
voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is
internally clamped to 5V.
The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see
Typical Performance Characteristics curves.
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET means the maximum output current will be determined
by power dissipation within the LMR62014 FET switch. The switch power dissipation from ON-state conduction is
calculated by:
P(SW) = DC x IIND(AVE)2x RDS(ON) (11)
There will be some switching losses as well, so some derating needs to be applied when calculating IC power
dissipation.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for this product include, but are not limited to Sumida, Coilcraft, Panasonic,
TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to
avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and
wire power losses must be considered when selecting the current rating.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If this function is not going to be used, the pin should be
tied directly to VIN. If the SHDN function will be needed, a pull-up resistor must be used to VIN (approximately
50k-100kΩrecommended). The SHDN pin must not be left unterminated.
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Efficiency vs Load Current
EFFICIENCY (%)
LOAD (mA)
050 100 150 200 250
0
10
20
30
40
50
60
70
80
90
100
300
3.3 - 9V Boost
LMR62014
SW
FB
GND
VIN
SHDN
U1
R3
51k
SHDN
GND
3.3 VIN
C1
2.2 PFR2
13.3k CF
330 pF
D1
R1/84k
L1/10 PH
C2
4.7 PF
D2
D3
D4
D5
R4 R5
9V OUT
240 mA (typ)
LMR62014
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SNVS735B –OCTOBER 2011–REVISED APRIL 2013
Figure 23. Flash LED Application
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REVISION HISTORY
Changes from Revision A (April 2013) to Revision B Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 15
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PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead finish/
Ball material
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LMR62014XMF/NOPB ACTIVE SOT-23 DBV 5 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 SH1B
LMR62014XMFE/NOPB ACTIVE SOT-23 DBV 5 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 SH1B
LMR62014XMFX/NOPB ACTIVE SOT-23 DBV 5 3000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 SH1B
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 2
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LMR62014XMF/NOPB SOT-23 DBV 5 1000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMR62014XMFE/NOPB SOT-23 DBV 5 250 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMR62014XMFX/NOPB SOT-23 DBV 5 3000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Dec-2016
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LMR62014XMF/NOPB SOT-23 DBV 5 1000 210.0 185.0 35.0
LMR62014XMFE/NOPB SOT-23 DBV 5 250 210.0 185.0 35.0
LMR62014XMFX/NOPB SOT-23 DBV 5 3000 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Dec-2016
Pack Materials-Page 2
www.ti.com
PACKAGE OUTLINE
C
0.22
0.08 TYP
0.25
3.0
2.6
2X 0.95
1.9
1.45
0.90
0.15
0.00 TYP
5X 0.5
0.3
0.6
0.3 TYP
8
0 TYP
1.9
A
3.05
2.75
B
1.75
1.45
(1.1)
SOT-23 - 1.45 mm max heightDBV0005A
SMALL OUTLINE TRANSISTOR
4214839/E 09/2019
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. Refernce JEDEC MO-178.
4. Body dimensions do not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
0.2 C A B
1
34
5
2
INDEX AREA
PIN 1
GAGE PLANE
SEATING PLANE
0.1 C
SCALE 4.000
www.ti.com
EXAMPLE BOARD LAYOUT
0.07 MAX
ARROUND 0.07 MIN
ARROUND
5X (1.1)
5X (0.6)
(2.6)
(1.9)
2X (0.95)
(R0.05) TYP
4214839/E 09/2019
SOT-23 - 1.45 mm max heightDBV0005A
SMALL OUTLINE TRANSISTOR
NOTES: (continued)
5. Publication IPC-7351 may have alternate designs.
6. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
SYMM
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:15X
PKG
1
34
5
2
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
EXPOSED METAL
METAL
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
(PREFERRED)
SOLDER MASK DETAILS
EXPOSED METAL
www.ti.com
EXAMPLE STENCIL DESIGN
(2.6)
(1.9)
2X(0.95)
5X (1.1)
5X (0.6)
(R0.05) TYP
SOT-23 - 1.45 mm max heightDBV0005A
SMALL OUTLINE TRANSISTOR
4214839/E 09/2019
NOTES: (continued)
7. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
8. Board assembly site may have different recommendations for stencil design.
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
SCALE:15X
SYMM
PKG
1
34
5
2
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