MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
________________________________________________________________ Maxim Integrated Products 1
19-3281; Rev 1; 1/05
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
General Description
The MAX5936/MAX5937 are hot-swap controllers for
-10V to -80V rails. The MAX5936/MAX5937 allow circuit
line cards to be safely hot-plugged into a live back-
plane without causing a glitch on the power supply.
These devices integrate a circuit-breaker function
requiring no RSENSE.
The MAX5936/MAX5937 provide a controlled turn-on
for circuit cards, limiting inrush, preventing glitches on
the power-supply rail, and preventing damage to board
connectors and components. Before startup, the
devices perform a Load Probe™ test to detect the
presence of a short-circuit condition. If a short-circuit
condition does not exist, the device limits the inrush
current drawn by the load by gradually turning on the
external MOSFET. Once the external MOSFET is fully
enhanced, the MAX5936/MAX5937 provides overcur-
rent and short-circuit protection by monitoring the volt-
age drop across the RDS(ON) of the external power
MOSFET. The MAX5936/MAX5937 integrate a 400mA
fast GATE pulldown to guarantee that the power
MOSFET is rapidly turned off in the event of an overcur-
rent or short-circuit condition.
The MAX5936/MAX5937 protect the system against
input voltage (VIN) steps by providing VIN step immuni-
ty. The MAX5936/MAX5937 provide an accurate UVLO
voltage. The MAX5936 has an open-drain, active-low
PGOOD output and the MAX5937 has an open-drain,
active-high PGOOD output.
The MAX5936/MAX5937 are offered with 100mV,
200mV, and 400mV circuit-breaker thresholds, in addi-
tion to a non-circuit-breaker option. These devices are
offered in latched and autoretry fault management, are
available in 8-pin SO packages, and specified for the
extended (-40°C to +85°C) temperature range (see the
Selector Guide).
Applications
Servers
Telecom Line Cards
Network Switches
Solid-State Circuit Breaker
Network Routers
Features
♦-10V to -80V Operation
♦No RSENSE Required
♦Drives Large Power MOSFETS
♦Programmable Inrush Current Limit During Hot
Plug
♦100mV, 200mV, 400mV, and No-Circuit-Breaker
Threshold Options
♦Circuit-Breaker Fault with Transient Rejection
♦Shorted Load Detection (Load Probe) Before
Power MOSFET Turn-On
♦±2.4% Accurate Undervoltage Lockout (UVLO)
♦Autoretry and Latched Fault Management
Available
♦Low Quiescent Current
GATE
LPVEE
1
2
8
7
PGOOD
(PGOOD)
VOUT
UVLO
STEP_MON
GND
SO
TOP VIEW
3
4
6
5
MAX5936
MAX5937
( ) FOR THE MAX5937.
Pin Configuration
Load Probe is a trademark of Maxim Integrated Products, Inc.
Ordering Information
PART TEMP RANGE PIN-PACKAGE
MAX5936_ _ESA -40°C to +85°C 8 SO
MAX5937_ _ESA -40°C to +85°C 8 SO
Note: The first “_” represents A for the autoretry and L for the
latched fault management option.
The second “_” represents the circuit-breaker threshold.
See the Selector Guide for additional information.
Selector Guide and Typical Operating Circuit appear at end
of data sheet.
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
VEE, VOUT, PGOOD (PGOOD), LP,
STEP_MON to GND............................................+0.3V to -85V
PGOOD (PGOOD) to VOUT ....................................-0.3V to +85V
PGOOD (PGOOD), LP, STEP_MON to VEE............-0.3V to +85V
GATE to VEE ...........................................................-0.3V to +20V
UVLO to VEE .............................................................-0.3V to +6V
Input Current
LP (internally, duty-cycle limited).........................................1A
PGOOD (PGOOD) (continuous) .....................................80mA
GATE (during 15V clamp, continuous) ...........................30mA
GATE (during 2V clamp, continuous) .............................50mA
GATE (during gate pulldown, continuous)......................50mA
Continuous Power Dissipation (TA= +70°C)
8-Pin SO (derate 5.9mW/°C above +70°C)..................471mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature .....................................................+150°C
Storage Temperature Range ............................-65°C to +150°C
Lead Temperature (soldering, 10s) ................................+300°C
ELECTRICAL CHARACTERISTICS
(VEE = -10V to -80V, VIN = GND - VEE, VSTEP_MON =V
EE, RLP = 200Ω, UVLO open, TA= -40°C to +85°C, unless otherwise noted.
Typical values are at VEE = -48V, TA= +25°C.) (Notes 1, 2)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Operating Voltage Range VEE Referenced to GND -80 -10 V
Operating Supply Current ICC 0.95 1.4 mA
VUVLO,R lVEEl increasing -33.5 -31.0 -29.5
Default VEE Undervoltage Lockout VUVLO,F lVEEl decreasing -28 V
UVLO Reference Threshold,
VEE Rising VUVLO_REF
,
RVUVLO increasing 1.219 1.25 1.281 V
UVLO Reference Threshold,
VEE Falling VUVLO_REF
,
FVUVLO decreasing 1.069 1.125 1.181 V
UVLO Input Resistance 20 50 kΩ
UVLO Transient Rejection tOVREJ 0.8 1.5 2.25 ms
Power-Up Delay (Note 3) tONDLY 80 220 380 ms
VEE and UVLO Glitch Rejection
(Note 4) tREJ 0.8 1.5 2.25 ms
VOUT to VEE Leakage Current VEE = -80V, VOUT = GND 0.01 1 µA
LP to VEE Leakage Current VEE = -80V, VLP = GND 0.01 1 µA
VIN = 10V 6.5 6.8 7.2
External Gate-Drive Voltage VGS VGATE - VEE 14 ≤ VIN ≤ 80V 8.1 10 12.8 V
ICLAMP = 9mA 13.5 16
MOSFET fully
enhanced ICLAMP = 20mA 17 19.5
ICLAMP = 1mA 2.1 2.55
GATE to VEE Clamp Voltage
Power-off,
VEE = GND ICLAMP = 10mA 2.5 2.9
V
Open-Loop Gate-Charge Current VGATE = VEE, VOUT = GND -66 -52 -35 µA
VIN > 10V 9 14.1
GATE Pulldown Switch
On-Resistance RGATE VGATE - VEE =
500mV VIN > 14V 7.5 12.5 Ω
Output-Voltage Slew Rate SR l dVOUT/dt l 2.4 9 14.8 V/ms
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VEE = -10V to -80V, VIN = GND - VEE, VSTEP_MON =V
EE, RLP = 200Ω, UVLO open, TA= -40°C to +85°C, unless otherwise noted.
Typical values are at VEE = -48V, TA= +25°C.) (Notes 1, 2)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Circuit-Breaker Tempco -40°C < TA <+85°C 6000 ppm/°C
Circuit-Breaker Glitch Rejection tCB_DLY 1.0 1.2 1.6 ms
TA = +85°C 118 140 162
TA = +25°C 85 100 115
TA = -10°C 64 79 94
MAX5936LA/
MAX5936AA/
MAX5937LA/
MAX5937AA TA = -40°C 62
TA = +85°C 244 284 324
TA = +25°C 180 200 220
TA = -10°C 135 158 181
MAX5936LB/
MAX5936AB/
MAX5937LB/
MAX5937AB TA = -40°C 124
TA = +85°C 485 568 651
TA = +25°C 355 400 445
TA = -10°C 270 316 362
Circuit-Breaker Threshold VCB
MAX5936LC/
MAX5936AC/
MAX5937LC/
MAX5937AC TA = -40°C 248
mV
TA = +85°C 220 280 340
TA = +25°C 160 200 240
TA = -10°C 111 158 205
MAX5936LA/
MAX5936AA/
MAX5937LA/
MAX5937AA TA = -40°C 124
TA = +85°C 470 568 667
TA = +25°C 350 400 450
TA = -10°C 255 316 377
MAX5936LB/
MAX5936AB/
MAX5937LB/
MAX5937AB TA = -40°C 248
TA = +85°C 962 1136 1310
TA = +25°C 700 800 900
TA = -10°C 510 632 754
Short-Circuit Threshold VSC
MAX5936LC/
MAX5936AC/
MAX5937LC/
MAX5937AC TA = -40°C 496
mV
Short-Circuit Response Time
(Note 5)
150mV overdrive, CLOAD = 0,
to GATE below 1V 330 500 ns
INPUT-VOLTAGE-STEP PROTECTION
Input-Voltage-Step Detection
Threshold STEPTH 1.219 1.250 1.281 V
Input-Voltage-Step Threshold
Offset Current ISTEP_OS -10.8 -10.0 -9.2 µA
LOAD-PROBE CIRCUIT
Load-Probe Switch On-Resistance VLP - VEE = 1V 7.5 11 Ω
Load-Probe Timeout tLP 80 220 380 ms
Load-Probe Retry Time tLP_OFF 16 x
tLP s
Load-Probe Voltage Threshold VTHSC-DET Referenced to GND -220 -200 -180 mV
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
4 _______________________________________________________________________________________
Note 1: All currents into pins are positive and all currents out of pins are negative. All voltages referenced to VEE, unless otherwise
specified.
Note 2: All limits are 100% tested at +25°C and +85°C. Limits at -40°C and -10°C are guaranteed by characterization.
Note 3: Delay time from a valid on-condition until the load probe test begins.
Note 4: VEE or UVLO voltages below VUVLO,F or VUVLO_REF,F, respectively, are ignored during this time.
Note 5: The time (VOUT - VEE) > VSC + overdrive until (VGATE - VEE) drops to approximately 90% of its initial high value.
Note 6: The time when the PGOOD (PGOOD) condition is met until the PGOOD (PGOOD) signal is asserted.
ELECTRICAL CHARACTERISTICS (continued)
(VEE = -10V to -80V, VIN = GND - VEE, VSTEP_MON =V
EE, RLP = 200Ω, UVLO open, TA= -40°C to +85°C, unless otherwise noted.
Typical values are at VEE = -48V, TA= +25°C.) (Notes 1, 2)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
LOGIC AND FAULT MANAGEMENT
Autoretry Delay tRETRY 16 x
tLP s
|VOUT - VEE| falling 0.72 x
VCB
PGOOD (PGOOD) Assertion
Threshold Hysteresis 0.26 x
VCB
mV
PGOOD (PGOOD) Assertion Delay
Time (Note 6) 0.67 1.26 1.85 ms
PGOOD (PGOOD) Low Voltage VOL ISINK = 1mA, referenced to VOUT,
VOUT < GND - 5V for PGOOD (PGOOD)0.05 0.4 V
PGOOD (PGOOD) Open-Drain
Leakage ILVEE = -80V, VPGOOD(PGOOD),
VPGOOD(PGOOD) = GND 0.01 1 µA
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
_______________________________________________________________________________________ 5
SUPPLY CURRENT
vs. INPUT VOLTAGE
MAZ5936 toc01
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
706040 503020
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
0
10 80
SUPPLY CURRENT
vs. TEMPERATURE
MAX5936 toc02
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
603510-15
0.2
0.4
0.6
0.8
1.0
1.2
0
-40 85
VIN = 72V
VIN = 48V
VIN = 12V
GATE-DRIVE VOLTAGE
vs. INPUT VOLTAGE
MAX5936 toc03
INPUT VOLTAGE (V)
GATE-DRIVE VOLTAGE (V)
706040 503020
6.5
7.0
7.5
8.0
8.5
9.0
9.5
10.0
10.5
6.0
10 80
GATE PULLDOWN CURRENT
vs. GATE VOLTAGE
MAX5936 toc04
VGATE (V)
GATE PULLDOWN CURRENT (mA)
986 72 3 4 51
50
100
150
200
250
300
350
400
450
500
0
010
RETRY TIME
vs. TEMPERATURE
MAX5936 toc05
TEMPERATURE (°C)
RETRY TIME (s)
603510-15
3.1
3.2
3.3
3.4
3.5
3.6
3.7
3.8
3.9
4.0
3.0
-40 85
CGATE = 0, CLOAD = 100µF
STARTUP WAVEFORM
MAX5936 toc06
40ms/div
VIN
50V/div
VGATE
10V/div
VOUT
50V/div
IIN
2A/div
VPGOOD
50V/div
MAX5936_A CIRCUIT-BREAKER EVENT
MAX5936 toc07
1ms/div
VGATE
10V/div
VOUT
50V/div
IIN
2A/div
VPGOOD
50V/div
Typical Operating Characteristics
(VEE = -48V, GND = 0V, VIN = GND - VEE, all voltages are referenced to VEE, TA= +25°C, unless otherwise noted.)
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
6 _______________________________________________________________________________________
MAX5936_A SHORT-CIRCUIT EVENT
MAX5936 toc08
400ns/div
VGATE
10V/div
VOUT
50V/div
IIN
10A/div
VPGOOD
50V/div
NORMALIZED CIRCUIT-BREAKER
THRESHOLD vs. TEMPERATURE
MAX5936 toc09
TEMPERATURE (°C)
NORMALIZED CIRCUIT-BREAKER THRESHOLD (%)
603510-15
0.6
0.8
1.0
1.2
1.4
1.6
0.4
-40 85
VOUT SLEW RATE
vs. TEMPERATURE
MAX5936 toc10
TEMPERATURE (°C)
SLEW RATE (V/ms)
603510-15
5.5
6.0
6.5
7.0
7.5
8.0
8.5
9.0
9.5
10.0
5.0
-40 85
CGATE = 0, CLOAD = 100µF
VIN = 48V
VIN = 72V
VIN = 12V
MAX5936_A INPUT VOLTAGE
STEP EVENT (NO FAULT)
MAX5936 toc11
4ms/div
VGATE
10V/div
VOUT
50V/div
IIN
1A/div
VPGOOD
50V/div
VIN
50V/div
RLOAD = 75Ω
MAX5936_A INPUT VOLTAGE
STEP EVENT (FAULT)
MAX5936 toc12
4ms/div
VGATE
10V/div
VOUT
50V/div
IIN
2A/div
VPGOOD
50V/div
VIN
50V/div
RLOAD = 75Ω
GATE TO VEE CLAMP VOLTAGE
AT POWER OFF
MAX5936 toc13
ISINK (mA)
GATE CLAMPING VOLTAGE (V)
18161412108642
0.5
1.0
1.5
2.0
2.5
3.0
0
020
VEE = GND = 0V
GATE TO VEE CLAMP VOLTAGE
MOSFET FULLY ENHANCED
MAX5936 toc14
ISINK (mA)
GATE CLAMPING VOLTAGE (V)
181612 144 6 8 102
9
10
11
12
13
14
15
16
17
18
8
020
VEE = -48V, VUVLO = 2V
Typical Operating Characteristics (continued)
(VEE = -48V, GND = 0V, VIN = GND - VEE, all voltages are referenced to VEE, TA= +25°C, unless otherwise noted.)
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
_______________________________________________________________________________________ 7
Detailed Description
The MAX5936/MAX5937 hot-swap controllers incorpo-
rate overcurrent fault management and are intended for
negative-supply-rail applications. The MAX5936/
MAX5937 eliminate the need for an external RSENSE
and include VIN input-step protection and load probe,
which prevents powering up into a shorted load. They
are intended for negative 48V telecom power systems
where low cost, flexibility, multifault management, and
compact size are required. The MAX5936/MAX5937 are
ideal for the widest range of systems from those
requiring low current with small MOSFETs to high-
current systems requiring large power MOSFETs and
low on-resistance.
The MAX5936/MAX5937 control an external n-channel
power MOSFET placed in the negative supply path of
an external load. When no power is applied, the GATE
output of the MAX5936/MAX5937 clamps the VGS of the
MOSFET to 2V, keeping the MOSFET turned off. When
power is applied to the MAX5936/MAX5937, the 2V
clamp at the GATE output is replaced by a strong pull-
down device pulling GATE to VEE and the VGS of the
MOSFET to 0V. As shown in Figure 2, this transition
enables the MAX5936/MAX5937 to keep the power
MOSFET continually off during the board insertion
phase when the circuit board first makes contact with
the backplane. Without this clamp, the GATE output of
a powered-down controller would be floating and the
MOSFET reverse transfer capacitance (gate-to-drain)
would pull up and turn on the MOSFET gate when the
MOSFET drain is rapidly pulled up by the VIN step dur-
ing backplane contact. The MAX5936/MAX5937 GATE
clamp can overcome the gate-to-drain capacitance of
large power MOSFETs with added slew-rate control
(CSLEW) capacitors while eliminating the need for addi-
tional gate-to-source capacitance. The MAX5936/
MAX5937 will keep the MOSFET off indefinitely if the
supply voltage is below the user-set UVLO threshold or
if a short circuit is detected in the load connected to the
drain of the power MOSFET.
PIN
MAX5936 MAX5937
NAME FUNCTION
11GND Ground. The high-supply connection for a negative-rail hot-swap controller.
22UVLO
Undervoltage Lockout Input, On/Off Control. Referenced to VEE. Drive UVLO above the
1.25V rising threshold to turn on the device. To turn off the device, drive UVLO below the
1.125V falling threshold for the 1.5ms glitch rejection period. Leave UVLO disconnected for
the default 31V undervoltage lockout threshold. Cycle UVLO to unlatch the MAX5936L/
MAX5937L after a fault. Cycling UVLO low deasserts PGOOD.
33STEP_MON
Input Voltage Step Monitor. 1.25V voltage threshold referenced to VEE. Connect a resistor
between STEP_MON and VEE to set the step sensitivity. Connect a capacitor from GND to
STEP_MON to adjust the step response relative to a step increase at VEE to eliminate false
circuit-breaker and short-circuit faults. Connect to VEE to disable the step immunity function
(see the Selecting Resistor and Capacitor Values for Step Monitor section in the
Applications Information).
44VEE Negative Input Voltage
55LP Load-Probe Detect. Connect a resistor from LP to VOUT to set the load-probe test current.
Limit load-probe test current to 1A. Connect to VEE to disable the load-probe function.
66GATE Gate-Drive Output. Connect to the gate of the external n-channel MOSFET.
77VOUT Output Voltage Sense. VOUT is the negative rail of the load. Connect to the drain of the
external n-channel MOSFET.
8—PGOOD Power-Good, Active-Low, Open-Drain Output. Referenced to VOUT. PGOOD asserts low
when VOUT is within the limits and there is no fault.
—8PGOOD Power-Good, Active-High, Open-Drain Output. Referenced to VOUT. PGOOD asserts high
when VOUT is within limits and there is no fault.
Pin Description
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
8 _______________________________________________________________________________________
The MAX5936/MAX5937 conduct a load-probe test after
contact transients from the hot plug-in have settled. This
follows the MAX5936/MAX5937 power-up (when the
UVLO condition has been met for 220ms (tLP)) and prior
to the turn-on of the power MOSFET. This test pulls a
user-programmable current through the load (1A, max)
for up to 220ms and tests for a voltage of 200mV across
the load at VOUT. This current is set by an external resis-
tor, RLP, between VOUT and LP (Figure 14). When the
voltage across the load exceeds 200mV, the test is trun-
cated and the GATE turn-on sequence is started. If at the
end of the 220ms test period the voltage across the load
has not reached 200mV, the load is assumed to be short-
ed and the current to the load from the LP pin is shut off.
The MAX5936A_/MAX5937A_ will timeout for 16 x tLP
then retry the load-probe test. The MAX5936L_/
MAX5937L_ will latch the fault condition indefinitely until
the UVLO is brought below 1.125V for 1.5ms or the power
is recycled. See the Applications Information section for
recommendations on selecting RLP to set the current
level.
Upon successful completion of the load-probe test, the
MAX5936/MAX5937 enter the power-up GATE cycle and
begin ramping the GATE voltage with a 52µA current
source. This current source is restricted if VOUT begins
to ramp down faster than the default 9V/ms slew rate.
Charging up GATE enhances the power MOSFET in a
controlled manner and ramping VOUT at a user-settable
rate controls the inrush current from the backplane. The
MAX5936/MAX5937 continue to charge up the GATE
until one of two events occurs: a normal power-up GATE
cycle is completed or a power-up to fault management is
detected (see the GATE Cycles section in Appendix A).
Figure 1. Functional Block Diagram
10V REG
AND
5V REG
UVLO
LOGIC
CONTROL
BANDGAP
REF
VBG
10µA
FAULT
DETECTION
SEQUENCER
CONTROLLER
TIMER
GATE
CONTROL
LOAD PROBE
TEST
VSC, VCB, AND
75% OF VCB
COMPARATORS
STEP
TEMPERATURE-
COMPENSATED
CURRENT SOURCE
VBG (1.25V)
+5V
+10V
52µA
2V AND
15V
CLAMP
PGOOD
PGOOD
LOGIC
MAX5936
MAX5937
PGOOD
PGOOD
VOUT
GATE
LP
RLOAD CLOAD
UVLO
GND
STEP_MON
VEE
GND
VEE
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
_______________________________________________________________________________________ 9
In a normal power-up GATE cycle, the voltage at VOUT
(referenced to VEE) ramps to below 72% of the circuit-
breaker threshold voltage, VCB. At this time, the remaining
GATE voltage is rapidly pulled up to full enhancement.
PGOOD is asserted 1.26ms after GATE is fully enhanced
(see Figure 4). If the voltage at VOUT remains above 72%
of the VCB (when GATE reaches 90% of full enhance-
ment), then a power-up to fault management fault has
occurred (see Figure 5). GATE is rapidly pulled to VEE,
turning off the power MOSFET and disconnecting the
load. PGOOD remains deasserted and the MAX5936/
MAX5937 enter the fault management mode.
When the power MOSFET is fully enhanced, the
MAX5936/MAX5937 monitor the drain voltage (VOUT) for
circuit-breaker and short-circuit faults. The MAX5936/
MAX5937 make use of the power MOSFET’s RDS(ON) as
the current-sense resistance to detect excessive current
through the load. The short-circuit threshold voltage,
VSC, is twice VCB (VSC = 2 x VCB) and is available in
100mV, 200mV, and 400mV thresholds. VCB and VSC
are temperature-compensated (increasing with tempera-
ture) to track the normalized temperature coefficient of
RDS(ON) for typical power MOSFETs.
When the load current is increased during full enhance-
ment, this causes VOUT to exceed VCB but remains less
than VSC, and starts the 1.2ms circuit-breaker glitch
rejection timer. At the end of the glitch rejection period,
if VOUT still exceeds VCB, the GATE is immediately
pulled to VEE (330ns), PGOOD (PGOOD) is deasserted,
and the part enters fault management. Alternatively,
during full enhancement when VOUT exceeds VSC,
there is no glitch rejection timer. GATE is immediately
pulled to VEE, PGOOD is deasserted, and the part
enters fault management.
Figure 3. Load Probe Test During Initial Power-Up
40ms/div
VEE
20V/div
VLP
20V/div
VOUT
20V/div
ALL VOLTAGES
REFERENCED TO GND
Figure 2. GATE Voltage Clamp During Power-Up
VIN
20V/div
VGATE
1V/div
4ms/div
CIN = 100µF
Figure 4. MAX5936 Normal Condition
VIN
50V/div
VPGOOD
50V/div
40ms/div
VGATE
10V/div
VOUT
50V/div
IIN
2A/div
Figure 5. MAX5936 Startup in Fault Condition
VIN
50V/div
VPGOOD
50V/div
40ms/div
VGATE
10V/div
VOUT
50V/div
IIN
2A/div
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
10 ______________________________________________________________________________________
The VIN step immunity provides a means for transition-
ing through a large step increase in VIN with minimal
backplane inrush current and without shutting down
the load. Without VIN step immunity (when the power
MOSFET is fully enhanced), a step increase in VIN will
result in a high inrush current and a large step in VOUT,
which can trip the circuit breaker. With VIN step immu-
nity, the STEP_MON input detects the step before a
short circuit is detected at VOUT and alters the
MAX5936/MAX5937 response to VOUT exceeding
VSC due to the step. The 1.25V voltage threshold at
STEP_MON and a 10µA current source at STEP_MON
allow the user to set the sensitivity of the step detection
with an external resistor to VEE. A capacitor is placed
between GND and the STEP_MON input, which, in con-
junction with the resistor, sets the STEP_MON time con-
stant. When a step is detected by the STEP_MON input
to rise above its threshold (STEPTH), the overcurrent
fault management is blocked and remains blocked as
long as STEPTH is exceeded. When STEPTH is exceed-
ed, the MAX5936/MAX5937 take no action until VOUT
rises above VSC or above VCB for the 1.2ms circuit-
breaker glitch rejection period. When either of these
conditions occurs, a step GATE cycle begins and the
GATE is immediately brought to VEE, which turns off the
power MOSFET to minimize the resulting inrush current
surge from the backplane and PGOOD remains assert-
ed. GATE is held at VEE for 350µs, and after about 1ms,
begins to ramp up thereby enhancing the power
MOSFET in a controlled manner as in the power-up
GATE cycle. This provides a controlled inrush current
to charge the load capacitance to the new supply volt-
age (see the GATE Cycles section in Appendix A).
As in the case of the power-up GATE cycle, if VOUT
drops to less than 72% of the programmed VCB, inde-
pendent of the state of STEP_MON, the GATE voltage
is rapidly pulled to full enhancement. PGOOD remains
asserted throughout the step. Otherwise, if the
STEP_MON input has decayed below its threshold but
VOUT remains above 72% of the programmed VCB
(when GATE reaches 90% of full enhancement),
(a step-to-fault management fault has occurred). GATE
is rapidly pulled to VEE, turning off the power MOSFET
and disconnecting the load, PGOOD (PGOOD) is
deasserted, and the MAX5936/MAX5937 enter the fault
management mode.
Fault Management
Fault management can be triggered by the following
conditions:
•V
OUT exceeds 72% of VCB during GATE ramp at
90% of full enhancement,
•V
OUT exceeds the VCB for longer than 1.2ms during
full enhancement,
•V
OUT exceeds the VSC during full enhancement, and
• Load-probe test fails.
Once in the fault management mode, GATE will always
be pulled to VEE to turn off the external MOSFET and
PGOOD (PGOOD) will always be deasserted. The
MAX5936A_/MAX5937A_ have automatic retry following
a fault while the MAX5936L_/MAX5937L remain latched
in the fault condition.
Autoretry Fault Management
(MAX5936A_/MAX5937A_)
If the MAX5936A_/MAX5937A_entered fault management
due to circuit-breaker and short-circuit faults, the
autoretry timer starts immediately. The timer times out in
3.5s (typ) and at the end of the timeout, the sequencer
initiates a load-probe test. If this is successful, it starts a
normal power-up GATE cycle.
Figure 6. MAX5936 Response to a Step Input (VOUT < 0.74VCB)
VIN
5V/div
VPGOOD
20V/div
2ms/div
VGATE
10V/div
VOUT
20V/div
IIN
1A/div
CLOAD = 100µF
RLOAD = 100Ω
40V
Figure 7. MAX5936 Response to a Step Input (VOUT > 0.74VCB)
VIN
20V/div
VPGOOD
50V/div
4ms/div
VGATE
10V/div
VOUT
50V/div
IIN
5A/div
40V
20V
CLOAD = 100µF
RLOAD = 20Ω
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 11
Latched Fault Management
(MAX5936L_/MAX5937L_)
When the MAX5936L_/MAX5937L_ enter fault manage-
ment, they remain in this condition indefinitely until the
power is recycled or until UVLO is brought below
1.125V for 1.5ms (typ) (when the short-circuit or circuit-
breaker fault has cleared, the sequencer initiates a load-
probe test). If this is successful, it starts a normal
power-up GATE cycle. A manual reset circuit (Figure 8)
can be used to clear the latch.
Circuit-Breaker Thresholds
The MAX5936/MAX5937 are available with 100mV,
200mV, and 400mV circuit-breaker thresholds. The
short-circuit voltage threshold (VSC) is twice the circuit-
breaker threshold voltage (VCB). In the MAX5936/
MAX5937, VCB and VSC are temperature-compensated
(increasing with temperature) to track the normalized
temperature gradient of typical power MOSFETs.
The proper circuit-breaker threshold for an application
depends on the RDS(ON) of the external power MOSFET
and the maximum current the load is expected to draw.
To avoid false fault indication and dropping of the load,
the designer must take into account the load response
to voltage ripples and noise from the backplane power
supply, as well as switching currents in the downstream
DC-DC converter that is loading the circuit. While the
circuit-breaker threshold has glitch rejection that
ignores ripples and noise lasting less than 1.2ms, the
short-circuit detection is designed to respond very
quickly (less than 330ns) to a short circuit. VSC and
VCB must be selected from the three available ranges
with an adequate margin to cover all possible ripples,
noise, and system current transients.
The short-circuit and circuit-breaker voltages are sensed
at VOUT, which is the drain of the power MOSFET. The
RDS(ON) of the MOSFET is the current-sense resis-
tance, so the total current through the load and load
capacitance is the drain current of the power MOSFET.
Accordingly, the voltage at VOUT as a function of
MOSFET drain current is:
VOUT = ID,MOSFET x RDS(ON)
The temperature compensation of the MAX5936/
MAX5937 is designed to track the RDS(ON) of the typi-
cal power MOSFET. Figure 9 shows the typical normal-
ized tempco of the circuit-breaker threshold along with
the normalized tempco of RDS(ON) for two typical power
MOSFETS. When determining the circuit-breaker
threshold in an application, go to the data sheet of the
power MOSFET and locate the manufacturer’s maxi-
mum RDS(ON) at +25°C with a VGS of 10V. Next, find
the figure presenting the tempco of normalized RDS(ON)
or on-resistance vs. temperature. Because this curve is
in normalized units typically with a value of 1 at +25°C,
it is possible to multiply the curve by the drain voltage
at +25°C and convert the curve to drain voltage. Now
compare this curve to that of the MAX5936/MAX5937
normalized tempco of the circuit-breaker threshold
to make a determination of the tracking error in mV
between the power MOSFET [ID,MOSFET x RDS(ON)]
and the MAX5936/MAX5937 over the application’s
operating temperature range. If the tempco of the
power MOSFET is greater than that of the MAX5936/
MAX5937, then additional margin will be required in
selecting the circuit-breaker and short-circuit voltages
at higher temperatures as compared to +25°C. When
dissipation in the power MOSFET is expected to lead to
local temperature elevation relative to ambient condi-
tions, then it becomes imperative that the MAX5936/
MAX5937 be located as close as possible to the power
MOSFET. The marginal effect of temperature differ-
ences on circuit-breaker and short-circuit voltages can
be estimated from a comparative plot such as Figure 9.
MAX5936LN and MAX5937LN
The MAX5936LN and MAX5937LN do not have circuit-
breaker and short-circuit thresholds and these faults
are ignored. For these devices PGOOD (PGOOD)
asserts 1.26ms after GATE has ramped to 90% of full
enhancement. The step detection function of the
MAX5936LN and MAX5937LN responds to VIN and
VOUT steps with the same voltage thresholds as the
MAX5936_C and MAX5937_C.
Figure 8. Resetting MAX5936L/MAX5937L after a Fault
Condition Using a Push-Button Switch
MAX5936L
MAX5937L
GND
UVLO
VEE
VIN = (GND - VEE)
R2
R1
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
12 ______________________________________________________________________________________
PGOOD (
PGOOD
) Open-Drain Output
The power-good outputs, PGOOD (PGOOD), are open
drain and are referenced to VOUT. They assert and latch
if VOUT ramps below 72% of VCB, and with the built-in
delay this occurs 1.26ms after the external MOSFET
becomes fully enhanced. PGOOD (PGOOD) deasserts
any time the part enters fault management. PGOOD
(PGOOD) has a delayed response to UVLO. The GATE
goes to VEE when UVLO is brought below 1.125V for
1.5ms. This turns off the power MOSFET and allows
VOUT to rise depending on the RC time constant of the
load. PGOOD (PGOOD), in this situation, deasserts
when VOUT rises above VCB for more than 1.4ms or
above VSC, whichever occurs first (see Figure 12b).
Due to the open-drain driver, PGOOD (PGOOD)
requires an external pullup resistor to GND. Due to this
external pullup, PGOOD will not follow positive VIN
steps as well as if it were driven by an active pullup. As
a result, when PGOOD (PGOOD) is asserted high, an
apparent negative glitch appears at PGOOD (PGOOD)
during a positive VIN step. This negative glitch is a
result of the RC time constant of the external resistor
and the PGOOD pin capacitance lagging the VIN step.
It is not due to switching of the internal logic. To mini-
mize this negative transient, it may be necessary to
increase the pullup current and/or to add a small
amount of capacitance from PGOOD (PGOOD) to GND
to compensate for the pin capacitance.
WARNING: For the MAX5936_N/MAX5937_N, PGOOD
(PGOOD) asserts 1.26ms after the power MOSFET is fully
enhanced, independent of VOUT. Once the MOSFET is
fully enhanced and UVLO is pulled below its respective
threshold, GATE pulls to VEE to turn off the power
MOSFET and disconnect the load. When UVLO is
cycled low, PGOOD (PGOOD) is deasserted. In sum-
mary, once the MOSFET is fully enhanced, the
MAX5936_N/ MAX5937_N ignore VOUT and deassert
PGOOD (PGOOD) when UVLO goes low or when the
power to the MAX5936_N/ MAX5937_N is fully recy-
cled.
Undervoltage Lockout (UVLO)
UVLO provides an accurate means to set the turn-on volt-
age level for the MAX5936/MAX5937. Use a resistor-
divider network from GND to VEE to set the desired
turn-on voltage (Figure 11). UVLO has hysteresis with a
rising threshold of 1.25V and a falling threshold of 1.125V.
A startup delay of 220ms allows contacts and voltages to
settle prior to initiating the startup sequence (Figure 12a).
Figure 9. MAX5936/MAX5937 Normalized Circuit-Breaker
Threshold (VCB)
NORMALIZED MOSFET ON-RESISTANCE
vs. TEMPERATURE
TEMPERATURE (°C)
NORMALIZED MOSFET ON-RESISTANCE
603510-15
0.6
0.8
1.0
1.2
1.4
1.6
0.4
-40 85
IRF1310NS
NORMALIZED RON
IRFR3910
NORMALIZED RON
MAX5936/MAX5937
NORMALIZED VCB
Figure 10. Circuit-Breaker Voltage Margin for High and Low Tempco Power MOSFETS
CIRCUIT-BREAKER
TRIP REGION
CIRCUIT-BREAKER
TRIP REGION
TA = +25°CTA = +25°C
VCB
ID x RDS,ON
∆VCB,MIN
∆VCB,MIN
VCB
ID x RDS,ON
RDS(ON) HIGH TEMPCO RDS(ON) LOW TEMPCO
VOLTAGE
TEMPERATURETEMPERATURE
VOLTAGE
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 13
This startup delay is from a valid UVLO condition until the
start of the load-probe test. There is glitch rejection on
UVLO going low, which requires that VUVLO remains
below its falling threshold for 1.5ms to turn off the part
(Figure 12b). Use the following formula to calculate the
MAX5936/MAX59337 turn-on voltage:
Where VON is the desired turn-on voltage of the
MAX5936/MAX5937 and VUVLO_REF,R is the 1.25V
UVLO rising threshold.
Output Voltage (VOUT)
Slew-Rate Control
The VOUT slew rate controls the inrush current required
to charge the load capacitor. The MAX5936/MAX5937
have a default internal slew rate set for 9V/ms. The inter-
nal circuit establishing this slew rate accommodates up
to about 1000pF of reverse transfer capacitance (miller
capacitance) in the external power MOSFET without
effecting the default slew rate. Using the default slew
rate, the inrush current required to charge the load
capacitance is given by:
IINRUSH (mA) = CLOAD (µF) x SR (V/ms)
where SR = 9V/ms (default, typ).
Applications Information
Selecting Resistor and Capacitor
for Step Monitor
When a positive VIN step or ramp occurs, the VIN
increase results in a voltage rise at both STEP_MON
and VOUT relative to VEE. When the voltage at
STEP_MON is above STEPTH the MAX5936/MAX5937
block short-circuit and circuit-breaker faults. During this
STEP_MON high condition, if VOUT rises above VSC, the
MAX5936/MAX5937 immediately and very rapidly pull
GATE to VEE. This turns off the power MOSFET to avoid
inrush current spiking. GATE is held low for 350µs.
About 1ms after the start of GATE pulldown, the
MAX5936/MAX5937 begin to ramp GATE up to turn on
the MOSFET in a controlled manner, which results in
ramping VOUT down to the new supply level (see the
GATE Cycles section in Appendix A).
R2 V
V
R1
ON
UVLO_REF, R
=−
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟×1
Figure 11. Setting the MAX5936/MAX5937 Turn-On Voltage
MAX5936
MAX5937
GND
UVLO
VEE
VIN = (GND - VEE)
R2
R1
Figure 12. UVLO Timing Diagram
1.5ms
VUVLO_REF,F
UVLO
GATE
VOUT
PGOOD
UVLO
ISC_DET
160ms LOAD PROBE
DETECTION TEST
BEGINS
(a) (b)
VUVLO_REF,R
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
14 ______________________________________________________________________________________
This occurs with the least possible disturbance to VOUT,
although during the brief period that the MOSFET is off,
the voltage across the load droops slightly depending
on the load current and load storage capacitance.
PGOOD remains asserted throughout the VIN step
event.
The objective in selecting the resistor and capacitor for
the step monitor function is to ensure that the VIN steps
of all anticipated slopes and magnitudes will be proper-
ly detected and blocked, which otherwise would result
in a circuit-breaker or short-circuit fault. The following is
a brief analysis for finding the resistor and capacitor.
For a more complete analysis, see Appendix B.
Figure 13 is a functional diagram exhibiting the
elements of the MAX5936/MAX5937 involved in the
step immunity function. This block diagram shows the
parallel relationship between VOUT and VSTEP_MON.
Each has an I*R component establishing the DC level
prior to a step. While it is referred to as a VIN step, it is
the dynamic response to a finite voltage ramp that is
of interest.
Given a positive VIN ramp with a ramp rate of dV/dt, the
approximate response of VOUT to VIN is:
VOUT(t) = (dV/dt) x τCx (1-e(-t / τL,eqv) )
+ RDS(ON) x ILOAD
where τC= CLOAD x RDS(ON) and τL,eqv is the equiva-
lent time constant of the load that must be found empir-
ically (see Appendix B).
Similarly, the response of STEP_MON to a VIN ramp is:
VSTEP_MON(t) = (dV/dt) x τSTEP x (1-e(-t / τSTEP) ) + 10µA
x RSTEP
where τSTEP = RSTEP_MON x CSTEP_MON.
For proper step detection, VSTEP_MON must exceed
STEPTH prior to VOUT reaching VSC or within 1.4ms of
VOUT reaching VCB (overall VIN ramp rates anticipated in
the application). VSTEP_MON must be set below STEPTH
with adequate margin, ∆VSTEP_MON, to accommodate
the tolerance of both ISTEP_OS (±8%) and RSTEP_MON.
RSTEP_MON is typically set to 100kΩwhich gives a
∆VSTEP_MON for a worst-case high of 0.36V.
Figure 13. MAX5936/MAX5937 Step Immunity Functional Diagram
FAULT
MANAGEMENT
CYCLE
GATE
LOW
tCB_DLY
ESL
ESR
C
LOADCLOAD
CSTEP_MON
RSTEP_MON
VSTEP_MON
RDS,ON
GATEVEE VOUT
VIN STEP
STEP_MON
ISTEP
ISTEP_OS
GND
STEPTH
STEP_DET
VSC
VCB CB TRIP
MAX5936
MAX5937
SC TRIP
NOTE: VSC, VCB, VSTEPTH, VSTEP_MON, AND VOUT ARE REFERENCED TO VEE.
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 15
The margin of VOUT with respect to VSC and VCB was set
when VSC and VCB were selected from the three avail-
able ranges. This margin may be lower at one of the tem-
perature extremes and if so, that value should be used in
the following discussion. These margins will be called
∆VCB and ∆VSC and they represent the minimum VOUT
excursion required to trip the respective fault.
To set τSTEP to block all VCB and VSC faults for any
ramp rate, find the ratio of ∆VSTEP_MON to ∆VCB and
choose τSTEP so:
τSTEP = 1.2 x τCx ∆VSTEP_MON / ∆VCB
And since RSTEP_MON = 100kΩ. This results in
CSTEP_MON = τSTEP / 100kΩ.
After the first-pass component selection, if sufficient
timing margin exists (see Appendix B), it is possible but
not necessary to lower RSTEP_MON below 100kΩto
reduce the sensitivity of STEP_MON to VIN noise.
Appendix B gives a more complete analysis and dis-
cussion of the step monitor function. It provides meth-
ods for the characterization of the load response to a
VIN ramp and graphical verification of the step monitor
timing margins for a set of design parameters.
Selecting the PGOOD (
PGOOD
)
Pullup Resistor
Due to the open-drain driver, PGOOD (PGOOD) requires
an external pullup resistor to GND. This resistor should be
selected to minimize the current load while PGOOD
(PGOOD) is low. The PGOOD output specification for VOL
is 0.4V at 1mA. As described in the Detailed Description,
the external pullup interferes with the ability of PGOOD
(PGOOD) to follow positive VIN steps as well as if it were
driven by an active pullup. When PGOOD (PGOOD) is
asserted high, an apparent negative glitch appears at
PGOOD during a positive VIN step. To minimize this
negative transient it may be necessary to increase the
pullup current and/or to add a small amount of capaci-
tance from PGOOD (PGOOD) to GND to compensate for
the pin capacitance.
Setting the Test Current Level for
Load-Probe Test
The load-probe test is a current test of the load that
avoids turning on the power MOSFET. The MAX5936/
MAX5937 have an internal switch (Q1 in Figure 14) that
pulls current through the load and through an external
current-limiting resistor, RLP. During the test, this switch is
pulsed on for up to 220ms (typ). Current is pulled through
the load, which should charge up the load capacitance
unless there is a short. If the voltage across the load
exceeds 200mV, the test is truncated and normal power-
up is allowed to proceed. If the voltage across the load
does not reach 200mV in the 220ms period that the
current is on, the load is assumed to be shorted and the
current to the load from the LP pin is shut off. The
MAX5936A_/MAX5937A_ time out for 16 x tLP then retry
the load-probe test. The MAX5936L_/MAX5937L_ latch
the fault condition indefinitely until the UVLO is brought
below 1.125V for 1.5ms or the power is recycled.
In the application, the current-limiting resistor should be
selected to minimize the current pulled through the load
while guaranteeing that it charges the maximum expected
load capacitance to 220mV in 80ms. These parameters
are the maximum load-probe test voltage and the mini-
mum load-probe current pulse period, respectively. The
maximum current possible is 1A, which is adequate to test
a load capacitance as large as 170,000µF over the typical
telecom operating voltage range.
ITEST (A) = CLOAD,MAX (F) x 220mV / 80ms
Since the minimum intended VIN for the application
results in the lowest ITEST, during the load-probe test,
this VIN,MIN should be used to set the RLP. This voltage
will likely be near VON,FALLING or VOFF for the applica-
tion.
RTEST(Ω) = VIN,MIN / ITEST = VIN,MIN x 80ms /
(CLOAD(MAX) x 220mV)
Example: VIN operating range = 36V to 72V, CLOAD =
10,000µF. First, find the RTEST, which will guarantee a
successful test of the load.
RLP = 36V x 80ms / (10,000µF x 220mV) = 1,309Ω⇒
1.30kΩ±1%
Next, evaluate the RLP at the maximum operating volt-
age to verify that it will not exceed the 1A current limit
for the load-probe test:
ITEST,MAX = VIN,MAX / RLP = 72V / 1.30kΩ= 55.4mA
If the CLOAD(MAX) is increased to 170,000µF, the test
current will approach the limit. In this case, RTEST will
be a much lower value and must include the internal
switch resistance. To find the external series resistor
value that will guarantee a successful test at the lowest
supply voltage, the maximum value for the load-probe
switch on-resistance of 11Ωshould be used:
RLP,TOT = 36V x 80ms / (170,000µF x 220mV)
= 77Ω= 11Ω+ RLP
RLP = 77Ω- 11Ω= 66Ω⇒66.5Ω±1%
Again RLP must be evaluated at the maximum operat-
ing voltage to verify that it will not exceed the 1A cur-
rent limit for the load-probe test. In this case, the
minimum value for the load-probe switch on-resistance
of 6Ωshould be used:
ITEST,MAX = VIN,MAX / RLP,TOT = 72V / (66.5Ω+ 6Ω)
= 993mA
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
16 ______________________________________________________________________________________
Adjusting the VOUT Slew Rate
The default slew rate is set internally for 9V/ms. The
slew rate can be reduced by placing an external
capacitor from the drain of the power MOSFET to the
GATE output of the MAX5936/MAX5937. Figure 15
shows a graph of Slew Rate vs. CSLEW. This graph
shows that for CSLEW < 4700pF there is very little effect
to the addition of external slew-rate control capaci-
tance. This is intended so the GATE output can drive
large MOSFETs with significant gate capacitance and
still achieve the default slew rate. To select a slew-rate
control capacitor, go into the graph with the desired
slew rate and find the value of the miller capacitance.
When CSLEW > 4700pF, SR and CSLEW are inversely
related. Given the desired slew rate, the required
CSLEW is found as follows:
CSLEW(nF) = 23 / SR (V/ms)
From the data sheet of the power MOSFET find the
reverse transfer capacitance (gate-to-drain capacitance)
above 10V. If the reverse transfer capacitance of the
external power MOSFET is 5% or more of CSLEW, then it
should be subtracted from CSLEW in the equation above.
Figure 16 gives an example of the external circuit for
controlling slew rate. Depending on the parasitics asso-
ciated with the selected power MOSFET, the addition of
CSLEW may lead to oscillation while the MOSFET and
GATE control are in the linear range. If this is an issue, an
external resistor, RGATE, in series with the gate of the
MOSFET is recommended to prevent possible oscilla-
tion. It should be as small as possible, e.g., 5Ωto 10Ω, to
avoid impacting the MOSFET turn-off performance of the
MAX5936/MAX5937.
Layout Guidelines
To benefit from the temperature compensation designed
into the MAX5936/MAX5937, the part should be placed
as close as possible to the power MOSFET that it is con-
trolling. The VEE pin of the MAX5936/ MAX5937 should
be placed close to the source pin of the power MOSFET
and they should share a wide trace. A common top layer
plane would service both the thermal and electrical
requirements. The load-probe current must be taken into
account. If this current is high, the layout traces and cur-
rent-limiting resistor must be sized appropriately. Stray
inductance must be minimized in the traces of the over-
all layout of the hot-swap controller, the power MOSFET,
and the load capacitor. Starting from the board con-
tacts, all high-current traces should be short, wide, and
direct. The potentially high pulse current pins of the
MAX5936/MAX5937 are GATE (when pulling GATE low),
Figure 14. Load Probe Functional Diagram
TIMING
LOGIC
MAX5936
MAX5937
CLOAD LOAD
LOAD
OK
VIN
VEE
Q1
RON
GATE VOUT
RLP
ITEST
ILOAD
200mV
GND
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 17
load-probe, and VEE. Because of the nature of the hot-
swap requirement, no decoupling capacitor is recom-
mended for the MAX5936/MAX5937. Because there is
no decoupling capacitor, stray inductance can result in
excessive ringing at the GND pin during power-up or
during very rapid VIN steps. This should be examined
in every application design since ringing at the GND
pin may exceed the absolute maximum supply rating
for the part.
Input Transient Protection
During hot plug-in/unplug and fast VIN steps, stray
inductance in the power path can cause voltage ring-
ing above the normal input DC value, which may
exceed the absolute maximum supply rating. An input
transient such as that caused by lightning can also put
a severe transient peak voltage on the input rail. The
following techniques are recommended to reduce the
effect of transients:
1) Minimize stray inductance in the power path using
wide traces and minimize loop area including the
power traces and the return ground path.
2) Add a high-frequency (ceramic) bypass capacitor
on the backplane as close as possible to the plug-
in connector (Figure 17).
3) Add a 1kΩresistor in series with the MAX5936/
MAX5937’s GND pin and a 0.1µF capacitor from
GND to VEE to limit transient current going into this pin.
Appendix A
GATE Cycles
The power-up GATE cycle and the step GATE cycle are
quite similar but have distinct differences. Understanding
these differences may clarify application issues.
GATE Cycle During Power-Up
The power-up GATE cycle occurs during the initial
power-up of the MAX5936/MAX5937 and the associat-
ed power MOSFET and load. The power-up GATE
cycle can result in full enhancement or in a fault (all
voltages are relative to VEE).
Power-Up to Full Enhancement:
1) At the beginning of the power-up sequence to the
start of the power-up GATE cycle, the GATE is held
at VEE. Following a successful completion of the
load-probe test, GATE is held at VEE for an addi-
tional 350µs and then is allowed to float for 650µs.
At this point, the GATE begins to ramp with 52µA
charging the gate of the power MOSFET. [GATE
turn-on]
2) When GATE reaches the gate threshold voltage of
the power MOSFET, VOUT begins to ramp down
toward VEE. [VOUT ramp]
3) When VOUT ramps below 72% VCB, the GATE is
rapidly pulled to full enhancement and the power-
up GATE cycle is complete. 1.26ms after GATE is
pulled to full enhancement, PGOOD will assert. [Full
enhancement]
Figure 15. MAX5936/MAX5937 Slew Rate vs. CSLEW
SLEW RATE vs. CSLEW
CSLEW (nF)
SLEW RATE (V/ms)
100101
0.1
1
10
0.01
0.1 1000
Figure 16. Adjusting the MAX5936/MAX5937 Slew Rate
MAX5936
MAX5937
VOUT
CLOAD LOAD
GATEVEE
GND
CSLEW
RGATE
-48V
GND
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
18 ______________________________________________________________________________________
Power-Up to Fault Management:
1) Same as step 1 above. [GATE turn-on]
2) Same as step 2 above. [VOUT ramp]
3) GATE ramps to 90% of full enhancement while
VOUT remains above 72% VCB, at which point the
GATE is rapidly pulled to VEE and fault manage-
ment is initiated. [Fault management]
GATE Cycle During VIN Step
A step GATE cycle occurs only after a successful
power-up GATE cycle to full enhancement occurs and
as a result of a positive VIN step (all voltages are
relative to VEE).
Step to Full Enhancement:
1) A VIN step occurs resulting in STEP_MON rising
above STEPTH before VOUT rises above VSC. [Step
detection]
2) After a step is detected, VOUT rises above VSC in
response to the step. When VOUT rises above VSC,
GATE is immediately pulled to VEE, rapidly turning off
the power MOSFET. GATE is held at VEE for 350µs to
dampen any ringing. Once GATE is pulled to VEE,
the gate cycle has begun and STEP_MON can safely
drop below STEPTH and successfully complete a
step GATE cycle to full enhancement without initiat-
ing fault management. [GATE pulldown]
3) Following the 350µs of GATE pulldown, GATE is
allowed to float for 650µs. At this point, the GATE
begins to ramp with 52µA charging the gate of the
power MOSFET. [GATE turn-on]
4) When GATE reaches the gate threshold voltage of
the power MOSFET, VOUT begins to ramp down
toward the new lower VEE. In the interval where
GATE is below the MOSFET threshold, the MOSFET
is off and VOUT will droop depending on the RC
time constant of the load. [VOUT ramp]
5) When VOUT ramps below 72% VCB, the GATE pulls
rapidly to full enhancement and the step GATE
cycle is complete. If STEP_MON remains above
STEPTH when GATE has ramped to 90% of full
enhancement and VOUT remains above 72% of
VCB, GATE remains at 90% and will not be pulled to
full enhancement. In this condition, if VOUT drops
below 72% of VCB before STEP_MON drops below
STEPTH, GATE is rapidly pulled to full enhancement
and the step GATE cycle is complete. PGOOD
remains asserted throughout the step GATE cycle.
[Full enhancement]
Step to Fault Management:
1) Same as step 1 above. [Step detection]
2) Same as step 2 above. [GATE pulldown]
3) Same as step 3 above. [GATE turn-on]
4) Same as step 4 above. [VOUT ramp]
5) If STEP_MON is below STEPTH when GATE ramps
to 90% of full enhancement and VOUT remains
above 72% VCB, GATE is rapidly pulled to VEE.
Fault management is initiated and PGOOD is de-
asserted. If STEP_MON is above STEPTH when
GATE ramps to 90% of full enhancement and VOUT
remains above 72% of VCB, GATE remains at 90%.
It will not be pulled to full enhancement nor will it be
pulled to VEE. In this condition, if VOUT drops below
72% of VCB before STEP_MON drops below
STEPTH, GATE is rapidly pulled to full enhancement
and a fault is avoided. Conversely, if STEP_MON
drops below STEPTH first, the GATE is rapidly
pulled to VEE, fault management is initiated, and
PGOOD is deasserted. [Fault management]
It should be emphasized that while STEP_MON remains
above STEPTH the current fault management is
blocked. During this time it is possible for there to be
multiple events involving VOUT rising above VSC then
those falling below 75% VCB. In each of these events,
when VOUT rises above VSC, a full GATE cycle is initiat-
ed where GATE is first pulled low then allowed to ramp
up. Then finally, when VOUT conditions are met, it will
be fully enhanced.
Figure 17. Protecting the MAX5936/MAX5937 Input from High-
Voltage Transients
0.1µF1µF68V
TVS
1kΩ
100kΩ
VEE
PGOOD
GND
BACKPLANE
48V ±10%
PLUG-IN CARD
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 19
GATE Output
GATE is a complex output structure and its condition at
any moment is dependent on various timing sequences in
response to multiple inputs. A diode to VEE prevents neg-
ative excursions. For positive excursions, the states are:
1) Power-off with 2V clamp.
2) 10Ωpulldown to VEE.
a. Continuous during startup delay and during
fault conditions.
b. Pulsed following detected step or OV
condition.
3) Floating with 15V clamp. [Prior to GATE ramp]
4) 47µA current source with 15V clamp. [GATE ramp]
5) Pullup to internal 10V supply with 15V clamp. [Full
enhancement]
Appendix B
Step Monitor Component
Selection Analysis
As mentioned previously in the Selecting Resistor and
Capacitor for Step Monitor section, the AC response
from VIN to VOUT is dependent on the parasitics of the
load. This is especially true for the load capacitor in
conjunction with the power MOSFET’s RDS(ON). The
load capacitor (with parasitic ESR and LSR) and the
power MOSFET’s RDS(ON) can be modeled as a heavily
damped second-order system. As such, this system
functions as a bandpass filter from VIN to VOUT limiting
the ability of VOUT to follow the VIN ramp. STEP_MON
lags the VIN ramp with a first-order RC response, while
VOUT lags with an overdamped second-order
response.
Given a positive VIN ramp with ramp rate of dV/dt, the
approximate response of VOUT to VIN is:
VOUT(t) = (dV/dt) x τCx (1-e(-t / τL,eqv) )
+ RDS(ON) x ILOAD (Equation 1)
where τC= CLOAD x RDS(ON).
Equation 1 is a simplification for the overdamped sec-
ond-order response of the load to a ramp input, τC=
CLOAD x RDS(ON), and corresponds to the ability of the
load capacitor to transfer dV/dt current to the fully
enhanced power MOSFET’s RDS(ON). The equivalent
time constant of the load (τL,eqv) accounts for the para-
sitic series inductance and resistance of the capacitor
and board interconnect. Determine τL,eqv empirically
with a few tests to characterize the load dynamic
response to VIN ramps.
Similarly, the response of STEP_MON to a VIN ramp is:
VSTEP_MON(t) = (dV/dt) x τSTEP x (1-e(-t / τSTEP) )
+ 10µA x RSTEP_MON (Equation 2)
where τSTEP = RSTEP_MON x CSTEP_MON.
For proper step detection, VSTEP_MON must exceed
STEPTH prior to VOUT reaching VSC or within 1.4ms of
VOUT reaching VCB (or overall VIN ramp rates anticipat-
ed in the application). It is impossible to give a fixed set
of design guidelines that rigidly apply over the wide
array of applications that use the MAX5936/
MAX5937. There are, however, limiting conditions and
recommendations that should be observed.
One limiting condition that must be observed is to ensure
that the STEP_MON time constant, τSTEP, is not so low
that at the lowest ramp rate, the anticipated STEPTH can-
not be obtained. The product (dV/dt) x τSTEP =
τSTEP_MON,MAX, is the maximum differential voltage at
STEP_MON if the VIN ramp were to continue indefinitely.
A related condition is setting the STEP_MON voltage
below STEPTH with adequate margin, ∆VSTEP_MON, to
accommodate the tolerance of both ISTEP_OS (±8%) and
RSTEP_MON. In determining τSTEP_MON, use the 9.2µA
limit to ensure sufficient margin with worst-case ISTEP_OS.
The margin of VOUT (with respect to VSC and VCB) is
set when VSC and VCB were selected from the three
available ranges. This margin may be lower at one of
the temperature extremes and if so, that value should
be used in the following discussion. These margins will
be called ∆VCB and ∆VSC and they represent the mini-
mum VOUT excursion required to trip the respective
fault. RSTEP_MON is typically set to 100kΩ±1%. This
gives a ∆VSTEP_MON of 0.25V, a worst-case low of
0.16V, and a worst-case high of 0.37V. In finding τSTEP
in the equation below, use ∆VSTEP_MON = 0.37V to
ensure sufficient margin with worst-case ISTEP_OS.
To set τSTEP to block all VCB and VSC faults for any
ramp rate, find the ratio of ∆VSTEP_MON to ∆VCB and
choose τSTEP so:
τSTEP = 1.2 x τCx ∆VSTEP_MON / ∆VCB
and since RSTEP_MON = 100kΩ:
CSTEP_MON = τSTEP / RSTEP_MON = τSTEP / 100kΩ
After the first-pass component selection, if sufficient
timing margin exists, it is possible but not necessary to
lower RSTEP below 100kΩto reduce the sensitivity of
STEP_MON to VIN noise.
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
20 ___________________________________________________
Verification of the Step
Monitor Timing
It is prudent to verify conclusively that all circuit-breaker
and short-circuit faults will be blocked for all ramp
rates. To do this, some form of graphical analysis is
recommended but first, find the value of τL,eqv of the
load by a series of ramp tests as indicated earlier.
These tests include evaluating the load with a series of
VIN ramps of increasing ramp rates and monitoring the
rate of VOUT rise during the ramp. Each VIN ramp
should have a constant slope. The VOUT response data
must be taken only during the positive ramp. Data
taken after VIN has leveled off at the new higher value
must not be used.
Figure 18 shows the load in parallel with the load
capacitor, CLOAD, and the parallel connection in series
with the power MOSFET, which is fully enhanced with
VGS = 10V. The objective is to determine τL,eqv from
the VOUT response.
Figure 19 shows the general response of VOUT to a VIN
ramp over time t. Equation 1 gives the response of VOUT
to a ramp of dV/dt. The product (dV/dt) x τC=
∆VOUT(max) or the maximum VOUT voltage differential if
the VIN ramp were to continue indefinitely. The parame-
ter of interest is ∆VOUT due to the ramp dV/dt, thus it is
necessary to subtract the DC shift in VOUT due to the
load resistance. For some loads, which are relatively
independent of supply voltage, this may be insignificant.
VOUT(t) = VOUT(t) - RDS(ON) x ILOAD
where ILOAD is a function of the VOUT level that should
be determined separately with DC tests.
At any time (t) the ∆VOUT fraction of ∆VOUT(max) is:
∆VOUT(t) / [(dV/dt) x τC] = (1-e(-t / τL,eqv))
If VOUT(t) is measured at time t, then the equivalent
time constant of the load is found from:
τL,eqv = -t / ln(1 - ∆VOUT / [(dV/dt) x τC])
As mentioned earlier, several measurements of ∆VOUT
at times t1, t2, t3, and t4 should be made during the
ramp. Each of these may result in slightly different val-
ues of τL,eqv and all values should then be averaged.
In making the measurements, the VIN ramp duration
should be such that ∆VOUT reaches 2 or 3 times the
selected ∆VSC. The ramp tests should include three
ramp rates: ∆VSC / τC, 2 x ∆VSC / τCand 4 x ∆VSC / τC.
The values of τL,eqv may vary over the range of slew
rates due to measurement error, nonlinear dynamics in
the load, and due to the fact that Equation 1 is a simpli-
fication from a higher order dynamic system. The
resulting range of τL,eqv values should be used to vali-
date the performance of the final design.
Having τC, τL,eqv, RSTEP, and CSTEP in a graphical
analysis using Equation 1 and Equation 2 can verify the
step monitor function by displaying the relative timing
of tCB, tSTEP, and tSC, which are the times when VCB,
VSTEP_MON, and VSC voltage thresholds are exceeded.
A simple spreadsheet for this purpose can be supplied
by Maxim upon request. Figures 20, 21, and 22 graphi-
cally verify a particular solution over 3 decades of VIN
ramp rates. In addition, Figure 22 verifies that this solu-
tion will block all circuit-breaker and short-circuit faults
for even the lowest VIN ramp that will cause VOUT to
exceed VCB.
Figure 18. VIN Ramp Test of Load
LEQU
REQV
CLOAD
LOAD
LOAD CAPACITOR
WITH PARASITICS
VIN RAMP
10V
RDS,ON
Figure 19. General Response of VOUT to a VIN Ramp
dv
dt τC
dv
dt
VIN
VOUT.F
VOUTi
VIN RAMP
0
t1 t2 t3 t4
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
______________________________________________________________________________________ 21
Chip Information
TRANSISTOR COUNT: 2320
PROCESS: BiCMOS
Figure 20. VOUT Response to VIN Ramp of 300V/ms
VOUT RESPONSE TO VIN RAMP OF 300V/ms
TIME (µs)
∆VOLTAGES (V)
764 52 31
0.3
0.6
0.9
1.2
1.5
1.8
2.1
2.4
2.7
0
08
A
B
dVIN
dt
tCB tSC EF
C
tSTEP
A = VIN (GND - VEE)
B = ∆VSTEP_MON
C = ∆VOUT
D = ∆VSTEP,TH
E = ∆VCB
F = ∆VSC
Figure 21. VOUT Response to VIN Ramp of 30V/ms
VOUT RESPONSE TO VIN RAMP
TIME (µs)
∆VOLTAGES (V)
A
B
tCB E
F
C
D
A = VIN (GND - VEE)
B = ∆VSTEP_MON
C = ∆VOUT
D = ∆VSTEP,TH
E = ∆VCB
F = ∆VSC
3632282420161284
0.2
0.4
0.6
0.8
1.0
1.2
1.4
0
040
tSC
tSTEP
Figure 22. VOUT Response to VIN Ramp of 3V/ms
VOUT RESPONSE TO VIN RAMP OF 3V/ms
TIME (µs)
∆VOLTAGES (V)
A = VIN (GND - VEE)
B = ∆VSTEP_MON
C = ∆VOUT
D = ∆VSTEP,TH
E = ∆VCB
F = ∆VSC
400300100 200
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0
0500
A
B
D
E
F
C
tSTEP
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
22 ______________________________________________________________________________________
Timing Table
NAME SYMBOL TYPICAL
TIME
(
s
)
Power-Up Delay tONDLY 220m
Load Probe Test Timeout tLP 220m
Load Probe Retry Time tLP_OFF 3.5
PGOOD (PGOOD) Assertion
Delay Time tPGOOD 1.26m
Autoretry Delay tRETRY 3.5
Circuit-Breaker Glitch Rejection tCB_DLY 1.4m
UVLO Glitch Rejection tREJ 1.5m
GATE Pulldown Pulse Following
a VIN step — 350µ
GATE Low After a VIN Step,
Prior to Ramp —1m
Selector Guide
PART
CIRCUIT-
BREAKER
THRESHOLD
(mV)
FAULT
MANAGEMENT
PGOOD
ASSERTION
MAX5936LA 100 Latch Low
MAX5936LB 200 Latch Low
MAX5936LC 400 Latch Low
MAX5936LN No circuit
breaker Latch Low
MAX5936AA 100 Autoretry Low
MAX5936AB 200 Autoretry Low
MAX5936AC 400 Autoretry Low
MAX5937LA 100 Latch High
MAX5937LB 200 Latch High
MAX5937LC 400 Latch High
MAX5937LN No circuit
breaker Latch High
MAX5937AA 100 Autoretry High
MAX5937AB 200 Autoretry High
MAX5937AC 400 Autoretry High
MAX5937 PGOOD
VOUT
UVLO
CLOAD
DC-DC
CONVERTER
V+
ON
V-
BACKPLANE
GND
-48V
VIN
*
*
*
*
GATEVEE
STEP_MON LP
GND
*OPTIONAL COMPONENTS
Typical Operating Circuit
MAX5936/MAX5937
-48V Hot-Swap Controllers with VIN
Step Immunity and No RSENSE
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 23
© 2005 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
SOICN .EPS
PACKAGE OUTLINE, .150" SOIC
1
1
21-0041 B
REV.DOCUMENT CONTROL NO.APPROVAL
PROPRIETARY INFORMATION
TITLE:
TOP VIEW
FRONT VIEW
MAX
0.010
0.069
0.019
0.157
0.010
INCHES
0.150
0.007
E
C
DIM
0.014
0.004
B
A1
MIN
0.053A
0.19
3.80 4.00
0.25
MILLIMETERS
0.10
0.35
1.35
MIN
0.49
0.25
MAX
1.75
0.050
0.016L0.40 1.27
0.3940.386D
D
MINDIM
D
INCHES
MAX
9.80 10.00
MILLIMETERS
MIN MAX
16 AC
0.337 0.344 AB8.758.55 14
0.189 0.197 AA5.004.80 8
N MS012
N
SIDE VIEW
H 0.2440.228 5.80 6.20
e 0.050 BSC 1.27 BSC
C
HE
eBA1
A
D
0∞-8∞
L
1
VARIATIONS:
ENGLISH • ???? • ??? • ???
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MAX5936, MAX5937
-48V Hot-Swap Controllers with VIN Step Immunity and No RSENSE
Hot-Swap Controllers with VIN Step Immunity and No RSENSE for -10V to -80V Rails
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MAX5936LNESA+T
SOIC;8 pin;31 mm
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-40C to +85C
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Materials Analysis
MAX5936AAESA+
SOIC;8 pin;31 mm
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-40C to +85C
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SOIC;8 pin;31 mm
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SOIC;8 pin;31 mm
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SOIC;8 pin;31 mm
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SOIC;8 pin;31 mm
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SOIC;8 pin;31 mm
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SOIC;8 pin;31 mm
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Package:
TYPE PINS FOOTPRINT
DRAWING CODE/VAR *
Temp
RoHS/Lead-Free?
Materials Analysis
MAX5937AAESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LNESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LC ESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LC ESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LBESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LBESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937AAESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ABESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LAESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LAESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ANESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ANESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ACESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ACESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937ABESA+
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LNESA+T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8+5*
-40C to +85C
RoHS/Lead-Free: Lead Free
Materials Analysis
MAX5937LAESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937AAESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ABESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ABESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ACESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ACESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ANESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LAESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LBESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LBESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LC ESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LC ESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LNESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937LNESA-T
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937ANESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX5937AAESA
SOIC;8 pin;31 mm
Dwg: 21-0041B (PDF)
Use pkgcode/variation: S8-5*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
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Document Ref.: 1 9-3281; Rev 1; 2005-06-14
This page last modified: 20 0 7-09-06
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