LM2717MT-ADJ/NOPB - NATIONAL SEMICONDUCTOR

LM2717-ADJ

LM2717-ADJ Dual Step-Down DC/DC Converter

Literature Number: SNVS407B

March 4, 2008

LM2717-ADJ

Dual Step-Down DC/DC Converter

General Description

The LM2717-ADJ is composed of two PWM DC/DC buck

(step-down) converters. Both converters are used to generate

an adjustable output voltage as low as 1.267V. Both also fea-

ture low RDSON (0.16Ω) internal switches for maximum effi-

ciency. Operating frequency can be adjusted anywhere

between 300kHz and 600kHz allowing the use of small ex-

ternal components. External soft-start pins for each converter

enables the user to tailor the soft-start times to a specific ap-

plication. Each converter may also be shut down indepen-

dently with its own shutdown pin. The LM2717-ADJ is

available in a low profile 24-lead TSSOP package ensuring a

low profile overall solution.

Features

■Adjustable buck converter with a 2.2A, 0.16Ω, internal

switch (Buck 1)

■Adjustable buck converter with a 3.2A, 0.16Ω, internal

switch (Buck 2)

■Operating input voltage range of 4V to 20V

■Input undervoltage protection

■300kHz to 600kHz pin adjustable operating frequency

■Over temperature protection

■Small 24-Lead TSSOP package

Applications

■TFT-LCD Displays

■Handheld Devices

■Portable Applications

■Laptop Computers

■Automotive Applications

Typical Application Circuit

20167901

LM2717-ADJ Dual Step-Down DC/DC Converter

Connection Diagram

Top View

20167904

24-Lead TSSOP

Ordering Information

Order Number Spec Package Type NSC Package Drawing Supplied As

LM2717MT-ADJ TSSOP-24 MTC24 61 Units, Rail

LM2717MTX-ADJ TSSOP-24 MTC24 2500 Units, Tape and Reel

LM2717MT-ADJ NOPB TSSOP-24 MTC24 61 Units, Rail

LM2717MTX-ADJ NOPB TSSOP-24 MTC24 2500 Units, Tape and Reel

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LM2717-ADJ

Pin Descriptions

Pin Name Function

1 PGND Power ground. PGND and AGND pins must be connected together directly at the part.

2 PGND Power ground. PGND and AGND pins must be connected together directly at the part.

3 AGND Analog ground. PGND and AGND pins must be connected together directly at the part.

4 FB1 Buck 1 output voltage feedback input.

5 VC1 Buck 1 compensation network connection. Connected to the output of the voltage error amplifier.

6 VBG Bandgap connection.

7 VC2 Buck 2 compensation network connection. Connected to the output of the voltage error amplifier.

8 FB2 Buck 2 output voltage feedback input.

9 AGND Analog ground. PGND and AGND pins must be connected together directly at the part.

10 AGND Analog ground. PGND and AGND pins must be connected together directly at the part.

11 PGND Power ground. PGND and AGND pins must be connected together directly at the part.

12 PGND Power ground. PGND and AGND pins must be connected together directly at the part.

13 SW2 Buck 2 power switch input. Switch connected between VIN pins and SW2 pin.

14 VIN Analog power input. All VIN pins are internally connected and should be connected together directly

at the part.

15 VIN Analog power input. All VIN pins are internally connected and should be connected together directly

at the part.

16 CB2 Buck 2 converter bootstrap capacitor connection.

17 SHDN2 Shutdown pin for Buck 2 converter. Active low.

18 SS2 Buck 2 soft start pin.

19 FSLCT Switching frequency select input. Use a resistor to set the frequency anywhere between 300kHz

and 600kHz.

20 SS1 Buck 1 soft start pin.

21 SHDN1 Shutdown pin for Buck 1 converter. Active low.

22 CB1 Buck 1 converter bootstrap capacitor connection.

23 VIN Analog power input. All VIN pins are internally connected and should be connected together directly

at the part.

24 SW1 Buck 1 power switch input. Switch connected between VIN pins and SW1 pin.

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LM2717-ADJ

Block Diagram

20167903

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LM2717-ADJ

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN −0.3V to 22V

SW1 Voltage −0.3V to 22V

SW2 Voltage −0.3V to 22V

FB1, FB2 Voltages −0.3V to 7V

CB1, CB2 Voltages −0.3V to VIN+7V

(VIN=VSW)

VC1 Voltage 1.75V ≤ VC1 ≤ 2.25V

VC2 Voltage 0.965V ≤ VC2 ≤ 1.565V

SHDN1 Voltage −0.3V to 7.5V

SHDN2 Voltage −0.3V to 7.5V

SS1 Voltage −0.3V to 2.1V

SS2 Voltage −0.3V to 2.1V

FSLCT Voltage AGND to 5V

Maximum Junction Temperature 150°C

Power Dissipation(Note 2) Internally Limited

Lead Temperature 300°C

Vapor Phase (60 sec.) 215°C

Infrared (15 sec.) 220°C

ESD Susceptibility (Note 3)

Human Body Model 2kV

Operating Conditions

Operating Junction

Temperature Range

(Note 4) −40°C to +125°C

Storage Temperature −65°C to +150°C

Supply Voltage 4V to 20V

SW1 Voltage 20V

SW2 Voltage 20V

Switching Frequency 300kHz to 600kHz

Electrical Characteristics

Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature

Range (TJ = −40°C to +125°C). VIN = 5V, IL = 0A, and FSW = 300kHz unless otherwise specified.

Symbol Parameter Conditions Min

(Note 4)

Typ

(Note 5)

Max

(Note 4) Units

IQTotal Quiescent Current (both

switchers)

Not Switching 2.7 6mA

Switching, switch open 6 12 mA

VSHDN = 0V 9 27 µA

VBG Bandgap Voltage 1.248

1.230 1.267 1.294

1.299 V

%VBG/ΔVIN Bandgap Voltage Line

Regulation

-0.01 0.01

0.125 %/V

VFB1 Buck 1 Feedback Voltage 1.236

1.214 1.258 1.286

1.288 V

VFB2 Buck 2 Feedback Voltage 1.236

1.214 1.258 1.286

1.288 V

ICL1(Note 6) Buck 1 Switch Current Limit VIN = 8V (Note 7) 2.2 A

VIN = 12V, VOUT = 3.3V 1.4 1.65 2.0

ICL2(Note 6) Buck 2 Switch Current Limit VIN = 8V (Note 7) 3.2 A

VIN = 12V, VOUT = 5V 2.6 3.05 3.5

IB1 Buck 1 FB Pin Bias Current

(Note 8)

VIN = 20V 70 400 nA

IB2 Buck 2 FB Pin Bias Current

(Note 8)

VIN = 20V 65 400 nA

VIN Input Voltage Range 4 20 V

gm1 Buck 1 Error Amp

Transconductance

ΔI = 20µA 1340 µmho

gm2 Buck 2 Error Amp

Transconductance

ΔI = 20µA 1360 µmho

AV1 Buck 1 Error Amp Voltage

Gain

134 V/V

AV2 Buck 2 Error Amp Voltage

Gain

136 V/V

DMAX Maximum Duty Cycle 89 93 %

FSW Switching Frequency RF = 46.4k 240 300 360 kHz

RF = 22.6k 480 600 720 kHz

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LM2717-ADJ

Symbol Parameter Conditions Min

(Note 4)

Typ

(Note 5)

Max

(Note 4) Units

ISHDN1 Buck 1 Shutdown Pin Current 0V < VSHDN1 < 7.5V −5 5µA

ISHDN2 Buck 2 Shutdown Pin Current 0V < VSHDN2 < 7.5V −5 5µA

IL1 Buck 1 Switch Leakage

Current

VIN = 20V 0.01 5µA

IL2 Buck 2 Switch Leakage

Current

VIN = 20V 0.01 5µA

RDSON1 Buck 1 Switch RDSON (Note 9) ISW = 100mA 160 180

300 mΩ

RDSON2 Buck 2 Switch RDSON (Note 9) ISW = 100mA 160 180

300 mΩ

ThSHDN1 Buck 1 SHDN Threshold Output High 1.8 1.36 V

Output Low 1.33 0.7

ThSHDN2 Buck 2 SHDN Threshold Output High 1.8 1.36 V

Output Low 1.33 0.7

ISS1 Buck 1 Soft Start Pin Current 4915 µA

ISS2 Buck 2 Soft Start Pin Current 4915 µA

UVP On Threshold 43.8 V

Off Threshold 3.6 3.3

θJA Thermal Resistance

(Note 10)

TSSOP, package only 115 °C/W

Note 1: Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the device is intended

to be functional, but device parameter specifications may not be guaranteed. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance,

θJA, and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance. The maximum allowable power dissipation at any ambient

temperature is calculated using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and

the regulator will go into thermal shutdown.

Note 3: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 4: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% tested

or guaranteed through statistical analysis. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC)

methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).

Note 5: Typical numbers are at 25°C and represent the most likely norm.

Note 6: Duty cycle affects current limit due to ramp generator.

Note 7: Current limit at 0% duty cycle. See TYPICAL PERFORMANCE section for Switch Current Limit vs. Input Voltage.

Note 8: Bias current flows into FB pin.

Note 9: Includes the bond wires and package leads, RDSON from VIN pin(s) to SW pin.

Note 10: Refer to National's packaging website for more detailed thermal information and mounting techniques for the TSSOP package.

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LM2717-ADJ

Typical Performance Characteristics

Shutdown IQ vs. Input Voltage

20167960

Switching IQ vs. Input Voltage

(FSW = 300kHz)

20167961

Switching Frequency vs. Input Voltage

(FSW = 300kHz)

20167962

Buck 1 RDS(ON) vs. Input Voltage

20167963

Buck 2 RDS(ON) vs. Input Voltage

20167964

Buck 1 Efficiency vs. Load Current

(VOUT = 3.3V)

20167965

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LM2717-ADJ

Buck 2 Efficiency vs. Load Current

(VOUT = 15V)

20167966

Buck 2 Efficiency vs. Load Current

(VOUT = 5V)

20167967

Buck 1 Switch Current Limit vs. Input Voltage

20167968

Buck 2 Switch Current Limit vs. Input Voltage

20167969

Buck 1 Switch Current Limit vs. Temperature

(VIN = 12V)

20167912

Buck 2 Switch Current Limit vs. Temperature

(VIN = 12V)

20167913

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LM2717-ADJ

Buck 1 Switch ON Resistance vs. Temperature

20167910

Buck 2 Switch ON Resistance vs. Temperature

20167911

Switching Frequency vs. RF Resistance

20167914

Buck Operation

PROTECTION (BOTH REGULATORS)

The LM2717-ADJ has dedicated protection circuitry running

during normal operation to protect the IC. The Thermal Shut-

down circuitry turns off the power devices when the die tem-

perature reaches excessive levels. The UVP comparator

protects the power devices during supply power startup and

shutdown to prevent operation at voltages less than the min-

imum input voltage. The OVP comparator is used to prevent

the output voltage from rising at no loads allowing full PWM

operation over all load conditions. The LM2717-ADJ also fea-

tures a shutdown mode for each converter decreasing the

supply current to approximately 10µA (both in shutdown

mode).

CONTINUOUS CONDUCTION MODE

The LM2717-ADJ contains current-mode, PWM buck regula-

tors. A buck regulator steps the input voltage down to a lower

output voltage. In continuous conduction mode (when the in-

ductor current never reaches zero at steady state), the buck

regulator operates in two cycles. The power switch is con-

nected between VIN and SW1 and SW2.

In the first cycle of operation the transistor is closed and the

diode is reverse biased. Energy is collected in the inductor

and the load current is supplied by COUT and the rising current

through the inductor.

During the second cycle the transistor is open and the diode

is forward biased due to the fact that the inductor current can-

not instantaneously change direction. The energy stored in

the inductor is transferred to the load and output capacitor.

The ratio of these two cycles determines the output voltage.

The output voltage is defined approximately as:

where D is the duty cycle of the switch, D and D′ will be re-

quired for design calculations.

The LM2717-ADJ has a minimum switch ON time which cor-

responds to a minimum duty cycle of approximately 10% at

600kHz operation and approximately 5% at 300kHz opera-

tion. In the case of some high voltage differential applications

(low duty cycle operation) this minimum duty cycle may be

exceeded causing the feedback pin over-voltage protection

to trip as the output voltage rises. This will put the device into

a PFM type operation which can cause an unpredictable fre-

quency spectrum and may cause the average output voltage

to rise slightly. If this is a concern the switching frequency may

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LM2717-ADJ

be lowered and/or a pre-load added to the output to keep the

device full PWM operation. Note that the OVP function mon-

itors the FB pin so it will not function if the feedback resistor

is disconnected from the output. Due to slight differences be-

tween the two converters it is recommended that Buck 1 be

used for the lower of the two output voltages for best opera-

tion.

DESIGN PROCEDURE

This section presents guidelines for selecting external com-

ponents.

SETTING THE OUTPUT VOLTAGE

The output voltage is set using the feedback pin and a resistor

divider connected to the output as shown in Figure 4. The

feedback pin voltage (VFB) is 1.258V, so the ratio of the feed-

back resistors sets the output voltage according to the follow-

ing equation:

INPUT CAPACITOR

A low ESR aluminum, tantalum, or ceramic capacitor is need-

ed between the input pin and power ground. This capacitor

prevents large voltage transients from appearing at the input.

The capacitor is selected based on the RMS current and volt-

age requirements. The RMS current is given by:

The RMS current reaches its maximum (IOUT/2) when

VIN equals 2VOUT. This value should be calculated for both

regulators and added to give a total RMS current rating. For

an aluminum or ceramic capacitor, the voltage rating should

be at least 25% higher than the maximum input voltage. If a

tantalum capacitor is used, the voltage rating required is

about twice the maximum input voltage. The tantalum capac-

itor should be surge current tested by the manufacturer to

prevent being shorted by the inrush current. The minimum

capacitor value should be 47µF for lower output load current

applications and less dynamic (quickly changing) load condi-

tions. For higher output current applications or dynamic load

conditions a 68µF to 100µF low ESR capacitor is recom-

mended. It is also recommended to put a small ceramic

capacitor (0.1µF to 4.7µF) between the input pins and ground

to reduce high frequency spikes.

INDUCTOR SELECTION

The most critical parameter for the inductor in a current mode

switcher is the minimum value required for stable operation.

To prevent subharmonic oscillations and achieve good phase

margin a target minimum value for the inductor is:

Where VIN is the minimum input voltage and RDSON is the

maximum switch ON resistance. For best stability the inductor

should be in the range of 0.5LMIN (absolute minimum) and

2LMIN. Using an inductor with a value less than 0.5LMIN can

cause subharmonic oscillations. The inductor should meet

this minimum requirement at the peak inductor current ex-

pected for the application regardless of what the inductor

ripple current and output ripple voltage requirements are. A

value larger than 2LMIN is acceptable if the ripple require-

ments of the application require it but it may reduce the phase

margin and increase the difficulty in compensating the circuit.

The most important parameters for the inductor from an ap-

plications standpoint are the inductance, peak current and the

DC resistance. The inductance is related to the peak-to-peak

inductor ripple current, the input and the output voltages (for

300kHz operation):

A higher value of ripple current reduces inductance, but in-

creases the conductance loss, core loss, and current stress

for the inductor and switch devices. It also requires a bigger

output capacitor for the same output voltage ripple require-

ment. A reasonable value is setting the ripple current to be

30% of the DC output current. Since the ripple current in-

creases with the input voltage, the maximum input voltage is

always used to determine the inductance. The DC resistance

of the inductor is a key parameter for the efficiency. Lower DC

resistance is available with a bigger winding area. A good

tradeoff between the efficiency and the core size is letting the

inductor copper loss equal 2% of the output power.

OUTPUT CAPACITOR

The selection of COUT is driven by the maximum allowable

output voltage ripple. The output ripple in the constant fre-

quency, PWM mode is approximated by:

The ESR term usually plays the dominant role in determining

the voltage ripple. Low ESR ceramic, aluminum electrolytic,

or tantalum capacitors (such as MuRata MLCC, Taiyo Yuden

MLCC, Nichicon PL series, Sanyo OS-CON, Sprague 593D,

594D, AVX TPS, and CDE polymer aluminum) is recom-

mended. An aluminum electrolytic capacitor is not recom-

mended for temperatures below −25°C since its ESR rises

dramatically at cold temperatures. Ceramic or tantalum ca-

pacitors have much better ESR specifications at cold tem-

perature and is preferred for low temperature applications.

BOOTSTRAP CAPACITOR

A 4.7nF ceramic capacitor or larger is recommended for the

bootstrap capacitor. For applications where the input voltage

is less than twice the output voltage a larger capacitor is rec-

ommended, generally 0.1µF to 1µF to ensure plenty of gate

drive for the internal switches and a consistently low RDSON.

SOFT-START CAPACITOR (BOTH REGULATORS)

The LM2717-ADJ contains circuitry that can be used to limit

the inrush current on start-up of the DC/DC switching regula-

tors. This inrush current limiting circuitry serves as a soft-start.

The external SS pins are used to tailor the soft-start for a

specific application. A current (ISS) charges the external soft-

start capacitor, CSS. The soft-start time can be estimated as:

TSS = CSS*0.6V/ISS

When programming the soft-start time use the equation given

in the Soft-Start Capacitor section above. The soft-start func-

tion is used simply to limit inrush current to the device that

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LM2717-ADJ

could stress the input voltage supply. The soft-start time de-

scribed above is the time it takes for the current limit to ramp

to maximum value. When this function is used the current limit

starts at a low value and increases to nominal at the set soft-

start time. Under maximum load conditions the output voltage

may rise at the same rate as the soft-start, however at light or

no load conditions the output voltage will rise much faster as

the switch will not need to conduct much current to charge the

output capacitor.

SHUTDOWN OPERATION (BOTH REGULATORS)

The shutdown pins of the LM2717-ADJ are designed so that

they may be controlled using 1.8V or higher logic signals. If

the shutdown function is not to be used the pin may be left

open. The maximum voltage to the shutdown pin should not

exceed 7.5V. If the use of a higher voltage is desired due to

system or other constraints it may be used, however a 100k

or larger resistor is recommended between the applied volt-

age and the shutdown pin to protect the device.

SCHOTTKY DIODE

The breakdown voltage rating of D1 and D2 is preferred to be

25% higher than the maximum input voltage. The current rat-

ing for the diode should be equal to the maximum output

current for best reliability in most applications. In cases where

the input voltage is much greater than the output voltage the

average diode current is lower. In this case it is possible to

use a diode with a lower average current rating, approximate-

ly (1-D)*IOUT however the peak current rating should be higher

than the maximum load current.

LOOP COMPENSATION

The general purpose of loop compensation is to meet static

and dynamic performance requirements while maintaining

stability. Loop gain is what is usually checked to determine

small-signal performance. Loop gain is equal to the product

of control-output transfer function and the output-control

transfer function (the compensation network transfer func-

tion). The DC loop gain of the LM2717 is usually around 55dB

to 60dB when loaded. Generally speaking it is a good idea to

have a loop gain slope that is -20dB /decade from a very low

frequency to well beyond the crossover frequency. The

crossover frequency should not exceed one-fifth of the

switching frequency, i.e. 60kHz in the case of 300kHz switch-

ing frequency. The higher the bandwidth is, the faster the load

transient response speed will potentially be. However, if the

duty cycle saturates during a load transient, further increasing

the small signal bandwidth will not help. Since the control-

output transfer function usually has very limited low frequency

gain, it is a good idea to place a pole in the compensation at

zero frequency, so that the low frequency gain will be rela-

tively large. A large DC gain means high DC regulation ac-

curacy (i.e. DC voltage changes little with load or line

variations). The rest of the compensation scheme depends

highly on the shape of the control-output plot.

20167916

FIGURE 1. Control-Output Transfer Function

As shown in Figure 1, the example control-output transfer

function consists of one pole (fp), one zero (fz), and a double

pole at fn (half the switching frequency). The following can be

done to create a -20dB /decade roll-off of the loop gain: Place

the first pole at 0Hz, the first zero at fp, the second pole at fz,

and the second zero at fn. The resulting output-control trans-

fer function is shown in Figure 2.

20167917

FIGURE 2. Output-Control Transfer Function

The control-output corner frequencies, and thus the desired

compensation corner frequencies, can be determined ap-

proximately by the following equations:

Where Co is the output capacitance, Re is the output capaci-

tance ESR, Ro is the load resistance, L is the inductor value,

and f is the switching frequency used.

Since fp is determined by the output network, it will shift with

loading (Ro) and duty cycle. First determine the range of fre-

quencies (fpmin/max) of the pole across the expected load

range, then place the first compensation zero within that

range.

Example: Vo = 5V, Re = 20mΩ, Co = 100µF, Romax = 5V/100-

mA = 50Ω, Romin = 5V/1A = 5Ω, L = 10µH, f = 300kHz:

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LM2717-ADJ

Once the fp range is determined, Rc1 should be calculated

using:

Where B is the desired gain in V/V at fp (fz1), gm is the

transconductance of the error amplifier, and R1 and R2 are

the feedback resistors as shown in Figure 3. A gain value

around 10dB (3.3v/v) is generally a good starting point.

Example: B = 3.3 v/v, gm=1350µmho, R1 = 20 KΩ, R2 = 59

KΩ:

Bandwidth will vary proportional to the value of Rc1. Next, Cc1

can be determined with the following equation:

Example: fpmin = 297 Hz, Rc1 = 20 KΩ:

The value of Cc1 should be within the range determined by

fpmin/max. A higher value will generally provide a more stable

loop, but too high a value will slow the transient response time.

The compensation network (Figure 3) will also introduce a low

frequency pole which will be close to 0Hz.

A second pole should also be placed at fz. This pole can be

created with a single capacitor Cc2 and a shorted Rc2 (see

Figure 3). The minimum value for this capacitor can be cal-

culated by:

Cc2 may not be necessary, however it does create a more

stable control loop. This is especially important with high load

currents.

Example: fz = 80 kHz, Rc1 = 20 KΩ:

A second zero can also be added with a resistor in series with

Cc2. If used, this zero should be placed at fn, where the con-

trol to output gain rolls off at -40dB/dec. Generally, fn will be

well below the 0dB level and thus will have little effect on sta-

bility. Rc2 can be calculated with the following equation:

20167930

FIGURE 3. Compensation Network

Note that the values calculated here give a good baseline for

stability and will work well with most applications. The values

in some cases may need to be adjusted some for optimum

stability or the values may need to be adjusted depending on

a particular applications bandwidth requirements.

LAYOUT CONSIDERATIONS

The LM2717-ADJ uses two separate ground connections,

PGND for the drivers and boost NMOS power device and

AGND for the sensitive analog control circuitry. The AGND

and PGND pins should be tied directly together at the pack-

age. The feedback and compensation networks should be

connected directly to a dedicated analog ground plane and

this ground plane must connect to the AGND pin. If no analog

ground plane is available then the ground connections of the

feedback and compensation networks must tie directly to the

AGND pin. Connecting these networks to the PGND can in-

ject noise into the system and effect performance.

The input bypass capacitor CIN, as shown in Figure 4, must

be placed close to the IC. This will reduce copper trace re-

sistance which effects input voltage ripple of the IC. For

additional input voltage filtering, a 0.1µF to 4.7µF bypass ca-

pacitors can be placed in parallel with CIN, close to the VIN

pins to shunt any high frequency noise to ground. The output

capacitors, COUT1 and COUT2, should also be placed close to

the IC. Any copper trace connections for the COUTX capacitors

can increase the series resistance, which directly effects out-

put voltage ripple. The feedback network, resistors RFB1(3)

and RFB2(4), should be kept close to the FB pin, and away from

the inductor to minimize copper trace connections that can

inject noise into the system. Trace connections made to the

inductors and schottky diodes should be minimized to reduce

power dissipation and increase overall efficiency. For more

detail on switching power supply layout considerations see

Application Note AN-1149: Layout Guidelines for Switching

Power Supplies.

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LM2717-ADJ

Application Information

Some Recommended Inductors (Others May Be Used)

Manufacturer Inductor Contact Information

Coilcraft DO3316 and DT3316 series www.coilcraft.com

800-3222645

TDK SLF10145 series www.component.tdk.com

847-803-6100

Pulse P0751 and P0762 series www.pulseeng.com

Sumida CDRH8D28 and CDRH8D43 series www.sumida.com

Some Recommended Input And Output Capacitors (Others May Be Used)

Manufacturer Capacitor Contact Information

Vishay Sprague 293D, 592D, and 595D series tantalum www.vishay.com

Taiyo Yuden High capacitance MLCC ceramic www.t-yuden.com

Cornell Dubilier ESRD seriec Polymer Aluminum Electrolytic

SPV and AFK series V-chip series www.cde.com

MuRata High capacitance MLCC ceramic www.murata.com

20167958

FIGURE 4. 15V, 3.3V Output Application

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LM2717-ADJ

20167959

FIGURE 5. 5V, 3.3V Output Application

20167915

FIGURE 6. 3.3V, 1.8V Output Application

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LM2717-ADJ

Physical Dimensions inches (millimeters) unless otherwise noted

TSSOP-24 Pin Package (MTC)

For Ordering, Refer to Ordering Information Table

NS Package Number MTC24

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LM2717-ADJ

Notes

LM2717-ADJ Dual Step-Down DC/DC Converter

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