LM3249TLX/NOPB - NATIONAL SEMICONDUCTOR

SM72501

SM72501 SolarMagic Precision, CMOS Input, RRIO, Wide Supply Range Amplifier

Literature Number: SNIS157B

SM72501

May 10, 2011

SolarMagic Precision, CMOS Input, RRIO, Wide Supply

Range Amplifier

General Description

The SM72501 is a low offset voltage, rail-to-rail input and out-

put precision amplifier with a CMOS input stage and a wide

supply voltage range. The SM72501 is ideal for sensor inter-

face and other instrumentation applications.

The guaranteed low offset voltage of less than ±200 µV along

with the guaranteed low input bias current of less than ±1 pA

makes the SM72501 ideal for precision applications. The

SM72501 is built utilizing VIP50 technology, which allows the

combination of a CMOS input stage and a 12V common mode

and supply voltage range. This makes the SM72501 a great

choice in many applications where conventional CMOS parts

cannot operate under the desired voltage conditions.

The SM72501 has a rail-to-rail input stage that significantly

reduces the CMRR glitch commonly associated with rail-to-

rail input amplifiers. This is achieved by trimming both sides

of the complimentary input stage, thereby reducing the differ-

ence between the NMOS and PMOS offsets. The output of

the SM72501 swings within 40 mV of either rail to maximize

the signal dynamic range in applications requiring low supply

voltage.

The SM72501 is offered in the space saving 5-Pin SOT23.

This small package is an ideal solution for area constrained

PC boards and portable electronics.

Features

■Renewable Energy Grade

Unless otherwise noted, typical values at VS = 5V

■Input offset voltage ±200 µV (max)

■Input bias current ±200 fA

■Input voltage noise 9 nV/√Hz

■CMRR 130 dB

■Open loop gain 130 dB

■Temperature range −40°C to 125°C

■Unity gain bandwidth 2.5 MHz

■Supply current (SM72501) 715 µA

■Supply voltage range 2.7V to 12V

■Rail-to-rail input and output

Applications

■High impedance sensor interface

■Battery powered instrumentation

■High gain amplifiers

■DAC buffer

■Instrumentation amplifier

■Active filters

Typical Application

30142105

Precision Current Source

SM72501 SolarMagic Precision, CMOS Input, RRIO, Wide Supply Range Amplifier

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

ESD Tolerance (Note 2)

Human Body Model 2000V

Machine Model 200V

Charge-Device Model 1000V

VIN Differential ±300 mV

Supply Voltage (VS = V+ – V−)13.2V

Voltage at Input/Output Pins V++ 0.3V, V− − 0.3V

Input Current 10 mA

Storage Temperature Range −65°C to +150°C

Junction Temperature (Note 3) +150°C

Soldering Information

Infrared or Convection (20 sec) 235°C

Wave Soldering Lead Temp. (10

sec) 260°C

Operating Ratings (Note 1)

Temperature Range (Note 3) −40°C to +125°C

Supply Voltage (VS = V+ – V−)2.7V to 12V

Package Thermal Resistance (θJA (Note 3))

5-Pin SOT23 265°C/W

3V Electrical Characteristics (Note 4)

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 3V, V− = 0V, VCM = V+/2, and RL > 10 kΩ to V+/2.

Boldface limits apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)Units

VOS Input Offset Voltage ±37 ±200

±500 μV

TCVOS Input Offset Voltage Temperature

Drift

(Note 7) ±1 ±5 μV/°C

IBInput Bias Current (Note 7, Note 8)

−40°C ≤ TA ≤ 85°C

±0.2 ±1

±50

(Note 7, Note 8)

−40°C ≤ TA ≤ 125°C

±0.2 ±1

±400

IOS Input Offset Current 40 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 3V 86

130 dB

PSRR Power Supply Rejection Ratio 2.7V ≤ V+ ≤ 12V, Vo = V+/2 86

98 dB

CMVR Common Mode Voltage Range CMRR ≥ 80 dB

CMRR ≥ 77 dB

–0.2

3.2

3.2 V

AVOL Open Loop Voltage Gain RL = 2 kΩ

VO = 0.3V to 2.7V

100

114

RL = 10 kΩ

VO = 0.2V to 2.8V

100

124

VOUT Output Voltage Swing High RL = 2 kΩ to V+/2 40 80

120 mV

from V+

RL = 10 kΩ to V+/2 30 40

Output Voltage Swing Low RL = 2 kΩ to V+/2 40 60

80 mV

RL = 10 kΩ to V+/2 20 40

IOUT Output Current

(Note 3, Note 9)

Sourcing VO = V+/2

VIN = 100 mV

Sinking VO = V+/2

VIN = −100 mV

ISSupply Current 0.670 1.0

1.2 mA

SR Slew Rate (Note 10) AV = +1, VO = 2 VPP

10% to 90%

0.9 V/μs

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SM72501

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)Units

GBW Gain Bandwidth 2.5 MHz

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, R.L = 10 kΩ 0.02 %

enInput Referred Voltage Noise

Density

f = 1 kHz 9 nV/

inInput Referred Current Noise

Density

f = 100 kHz 1 fA/

5V Electrical Characteristics (Note 4)

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2, and RL > 10 kΩ to V+/2.

Boldface limits apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)Units

VOS Input Offset Voltage ±37 ±200

±500 μV

TCVOS Input Offset Voltage Temperature Drift (Note 7) ±1 ±5 μV/°C

IBInput Bias Current (Note 7, Note 8)

−40°C ≤ TA ≤ 85°C

±0.2 ±1

±50

(Note 7, Note 8)

−40°C ≤ TA ≤ 125°C

±0.2 ±1

±400

IOS Input Offset Current 40 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 5V 88

130 dB

PSRR Power Supply Rejection Ratio 2.7V ≤ V+ ≤ 12V, VO = V+/2 86

100 dB

CMVR Common Mode Voltage Range CMRR ≥ 80 dB

CMRR ≥ 78 dB

–0.2

5.2

5.2 V

AVOL Open Loop Voltage Gain RL = 2 kΩ

VO = 0.3V to 4.7V

100

119

RL = 10 kΩ

VO = 0.2V to 4.8V

100

130

VOUT Output Voltage Swing High RL = 2 kΩ to V+/2 60 110

130 mV

from V+

RL = 10 kΩ to V+/2 40 50

Output Voltage Swing Low RL = 2 kΩ to V+/2 50 80

90 mV

RL = 10 kΩ to V+/2 30 40

IOUT Output Current

(Note 3, Note 9)

Sourcing VO = V+/2

VIN = 100 mV

Sinking VO = V+/2

VIN = −100 mV

ISSupply Current 0.715 1.0

1.2 mA

SR Slew Rate (Note 10) AV = +1, VO = 4 VPP

10% to 90%

1.0 V/μs

GBW Gain Bandwidth 2.5 MHz

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 10 kΩ 0.02 %

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SM72501

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)Units

enInput Referred Voltage Noise Density f = 1 kHz 9 nV/

inInput Referred Current Noise Density f = 100 kHz 1 fA/

±5V Electrical Characteristics (Note 4)

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = −5V, VCM = 0V, and RL > 10 kΩ to 0V. Bold-

face limits apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)Units

VOS Input Offset Voltage ±37 ±200

±500 μV

TCVOS Input Offset Voltage Temperature Drift (Note 7) ±1 ±5 μV/°C

IBInput Bias Current (Note 7, Note 8)

−40°C ≤ TA ≤ 85°C

±0.2 1

±50

(Note 7, Note 8)

−40°C ≤ TA ≤ 125°C

±0.2 1

±400

IOS Input Offset Current 40 fA

CMRR Common Mode Rejection Ratio −5V ≤ VCM ≤ 5V 92

138 dB

PSRR Power Supply Rejection Ratio 2.7V ≤ V+ ≤ 12V, VO = 0V 86

98 dB

CMVR Common Mode Voltage Range CMRR ≥ 80 dB

CMRR ≥ 78 dB

−5.2

5.2

5.2 V

AVOL Open Loop Voltage Gain RL = 2 kΩ

VO = −4.7V to 4.7V

100

121

RL = 10 kΩ

VO = −4.8V to 4.8V

100

134

VOUT Output Voltage Swing High RL = 2 kΩ to 0V 90 150

170 mV

from V+

RL = 10 kΩ to 0V 40 80

100

Output Voltage Swing Low RL = 2 kΩ to 0V 90 130

150 mV

from V–

RL = 10 kΩ to 0V 40 50

IOUT Output Current

(Note 3, Note 9)

Sourcing VO = 0V

VIN = 100 mV

Sinking VO = 0V

VIN = −100 mV

ISSupply Current 0.790 1.1

1.3 mA

SR Slew Rate (Note 10) AV = +1, VO = 9 VPP

10% to 90%

1.1 V/μs

GBW Gain Bandwidth 2.5 MHz

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 10 kΩ 0.02 %

enInput Referred Voltage Noise Density f = 1 kHz 9 nV/

inInput Referred Current Noise Density f = 100 kHz 1 fA/

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics

Tables.

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SM72501

Note 2: Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC) Field-

Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).

Note 3: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is

PD = (TJ(MAX) – TA)/ θJA. All numbers apply for packages soldered directly onto a PC Board.

Note 4: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating

of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ >

TA.

Note 5: Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will

also depend on the application and configuration. The typical values are not tested and are not guaranteed on shipped production material.

Note 6: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the Statistical Quality

Control (SQC) method.

Note 7: This parameter is guaranteed by design and/or characterization and is not tested in production.

Note 8: Positive current corresponds to current flowing into the device.

Note 9: The short circuit test is a momentary test.

Note 10: The number specified is the slower of positive and negative slew rates.

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SM72501

Connection Diagram

5-Pin SOT23

30142102

Top View

Ordering Information

Package Part Number Package Marking Transport Media NSC Drawing

5-Pin SOT23 SM72501MFE S501 250 Units Tape and Reel MF05A

5-Pin SOT23 SM72501MF S501 1000 Units Tape and Reel MF05A

5-Pin SOT23 SM72501MFX S501 3000 Units Tape and Reel MF05A

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SM72501

Typical Performance Characteristics Unless otherwise noted: TA = 25°C, VCM = VS/2, RL > 10 kΩ.

Offset Voltage Distribution

30142136

TCVOS Distribution

30142141

Offset Voltage Distribution

30142137

TCVOS Distribution

30142142

Offset Voltage Distribution

30142138

TCVOS Distribution

30142143

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SM72501

Offset Voltage vs. Temperature

30142106

CMRR vs. Frequency

30142150

Offset Voltage vs. Supply Voltage

30142110

Offset Voltage vs. VCM

30142107

Offset Voltage vs. VCM

30142108

Offset Voltage vs. VCM

30142109

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SM72501

Input Bias Current vs. VCM

30142146

Input Bias Current vs. VCM

30142130

Input Bias Current vs. VCM

30142147

Input Bias Current vs. VCM

30142131

Input Bias Current vs. VCM

30142148

Input Bias Current vs. VCM

30142149

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SM72501

PSRR vs. Frequency

30142145

Supply Current vs. Supply Voltage (Per Channel)

30142111

Sinking Current vs. Supply Voltage

30142113

Sourcing Current vs. Supply Voltage

30142112

Output Voltage vs. Output Current

30142116

Slew Rate vs. Supply Voltage

30142117

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SM72501

Open Loop Frequency Response

30142115

Open Loop Frequency Response

30142114

Large Signal Step Response

30142118

Small Signal Step Response

30142120

Large Signal Step Response

30142119

Small Signal Step Response

30142126

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SM72501

Input Voltage Noise vs. Frequency

30142127

Open Loop Gain vs. Output Voltage Swing

30142152

Output Swing High vs. Supply Voltage

30142133

Output Swing Low vs. Supply Voltage

30142135

Output Swing High vs. Supply Voltage

30142132

Output Swing Low vs. Supply Voltage

30142134

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SM72501

THD+N vs. Frequency

30142128

THD+N vs. Output Voltage

30142129

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SM72501

Application Information

SM72501

The SM72501 is a low offset voltage, rail-to-rail input and out-

put precision amplifier with a CMOS input stage and wide

supply voltage range of 2.7V to 12V. The SM72501 has a very

low input bias current of only ±200 fA at room temperature.

The wide supply voltage range of 2.7V to 12V over the ex-

tensive temperature range of −40°C to 125°C makes the

SM72501 an excellent choice for low voltage precision appli-

cations with extensive temperature requirements.

The SM72501 has only ±37 μV of typical input referred offset

voltage and this offset is guaranteed to be less than ±500 μV

over temperature. This minimal offset voltage allows more

accurate signal detection and amplification in precision appli-

cations.

The low input bias current of only ±200 fA along with the low

input referred voltage noise of 9 nV/ gives the SM72501

superiority for use in sensor applications. Lower levels of

noise from the SM72501 means better signal fidelity and a

higher signal-to-noise ratio.

National Semiconductor is heavily committed to precision

amplifiers and the market segment they serve. Technical sup-

port and extensive characterization data is available for sen-

sitive applications or applications with a constrained error

budget.

The SM72501 is offered in the space saving 5-Pin SOT23.

This small package is an ideal solution for area constrained

PC boards and portable electronics.

CAPACITIVE LOAD

The SM72501 can be connected as a non-inverting unity gain

follower. This configuration is the most sensitive to capacitive

The combination of a capacitive load placed on the output of

an amplifier along with the amplifier's output impedance cre-

ates a phase lag which in turn reduces the phase margin of

the amplifier. If the phase margin is significantly reduced, the

response will be either underdamped or it will oscillate.

In order to drive heavier capacitive loads, an isolation resistor,

RISO, in Figure 1 should be used. By using this isolation re-

sistor, the capacitive load is isolated from the amplifier's

output, and hence, the pole caused by CL is no longer in the

feedback loop. The larger the value of RISO, the more stable

the output voltage will be. If values of RISO are sufficiently

large, the feedback loop will be stable, independent of the

value of CL. However, larger values of RISO result in reduced

output swing and reduced output current drive.

30142121

FIGURE 1. Isolating Capacitive Load

INPUT CAPACITANCE

CMOS input stages inherently have low input bias current and

higher input referred voltage noise. The SM72501 enhances

this performance by having the low input bias current of only

±200 fA, as well as, a very low input referred voltage noise of

9 nV/ . In order to achieve this a larger input stage has

been used. This larger input stage increases the input capac-

itance of the SM72501. The typical value of this input capac-

itance, CIN, for the SM72501 is 25 pF. The input capacitance

will interact with other impedances such as gain and feedback

resistors, which are seen on the inputs of the amplifier, to form

a pole. This pole will have little or no effect on the output of

the amplifier at low frequencies and DC conditions, but will

play a bigger role as the frequency increases. At higher fre-

quencies, the presence of this pole will decrease phase mar-

gin and will also cause gain peaking. In order to compensate

for the input capacitance, care must be taken in choosing the

feedback resistors. In addition to being selective in picking

values for the feedback resistor, a capacitor can be added to

the feedback path to increase stability.

The DC gain of the circuit shown in Figure 2 is simply –R2/

R1.

30142144

FIGURE 2. Compensating for Input Capacitance

For the time being, ignore CF. The AC gain of the circuit in

Figure 2 can be calculated as follows:

This equation is rearranged to find the location of the two

poles:

(1)

As shown in Equation 1, as values of R1 and R2 are increased,

the magnitude of the poles is reduced, which in turn decreas-

es the bandwidth of the amplifier. Whenever possible, it is

best to choose smaller feedback resistors. Figure 3 shows the

effect of the feedback resistor on the bandwidth of the

SM72501.

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SM72501

30142154

FIGURE 3. Closed Loop Gain vs. Frequency

Equation 1 has two poles. In most cases, it is the presence of

pairs of poles that causes gain peaking. In order to eliminate

this effect, the poles should be placed in Butterworth position,

since poles in Butterworth position do not cause gain peaking.

To achieve a Butterworth pair, the quantity under the square

root in Equation 1 should be set to equal −1. Using this fact

and the relation between R1 and R2, R2 = −AV R1, the optimum

value for R1 can be found. This is shown in Equation 2. If R1

is chosen to be larger than this optimum value, gain peaking

will occur.

(2)

In Figure 2, CF is added to compensate for input capacitance

and to increase stability. Additionally, CF reduces or elimi-

nates the gain peaking that can be caused by having a larger

feedback resistor. Figure 4 shows how CF reduces gain peak-

ing.

30142155

FIGURE 4. Closed Loop Gain vs. Frequency with

Compensation

DIODES BETWEEN THE INPUTS

The SM72501 has a set of anti-parallel diodes between the

input pins, as shown in Figure 5. These diodes are present to

protect the input stage of the amplifier. At the same time, they

limit the amount of differential input voltage that is allowed on

the input pins. A differential signal larger than one diode volt-

age drop might damage the diodes. The differential signal

between the inputs needs to be limited to ±300 mV or the input

current needs to be limited to ±10 mA.

30142125

FIGURE 5. Input of SM72501

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SM72501

PRECISION CURRENT SOURCE

The SM72501 can be used as a precision current source in

many different applications. Figure 6 shows a typical preci-

sion current source. This circuit implements a precision volt-

age controlled current source. Amplifier A1 is a differential

amplifier that uses the voltage drop across RS as the feedback

signal. Amplifier A2 is a buffer that eliminates the error current

from the load side of the RS resistor that would flow in the

feedback resistor if it were connected to the load side of the

RS resistor. In general, the circuit is stable as long as the

closed loop bandwidth of amplifier A2 is greater then the

closed loop bandwidth of amplifier A1. Note that if A1 and A2

are the same type of amplifiers, then the feedback around A1

will reduce its bandwidth compared to A2.

30142105

FIGURE 6. Precision Current Source

The equation for output current can be derived as follows:

Solving for the current I results in the following equation:

LOW INPUT VOLTAGE NOISE

The SM72501 has a very low input voltage noise of 9 nV/

. This input voltage noise can be further reduced by plac-

ing N amplifiers in parallel as shown in Figure 7. The total

voltage noise on the output of this circuit is divided by the

square root of the number of amplifiers used in this parallel

combination. This is because each individual amplifier acts as

an independent noise source, and the average noise of inde-

pendent sources is the quadrature sum of the independent

sources divided by the number of sources. For N identical

amplifiers, this means:

Figure 7 shows a schematic of this input voltage noise reduc-

tion circuit. Typical resistor values are:

RG = 10Ω, RF = 1 kΩ, and RO = 1 kΩ.

30142156

FIGURE 7. Noise Reduction Circuit

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SM72501

TOTAL NOISE CONTRIBUTION

The SM72501 has very low input bias current, very low input

current noise, and very low input voltage noise. As a result,

these amplifiers are ideal choices for circuits with high

impedance sensor applications.

Figure 8 shows the typical input noise of the SM72501 as a

function of source resistance where:

en denotes the input referred voltage noise

ei is the voltage drop across source resistance due to input

referred current noise or ei = RS * in

et shows the thermal noise of the source resistance

eni shows the total noise on the input.

Where:

The input current noise of the SM72501 is so low that it will

not become the dominant factor in the total noise unless

source resistance exceeds 300 MΩ, which is an unrealisti-

cally high value.

As is evident in Figure 8, at lower RS values, total noise is

dominated by the amplifier's input voltage noise. Once RS is

larger than a few kilo-Ohms, then the dominant noise factor

becomes the thermal noise of RS. As mentioned before, the

current noise will not be the dominant noise factor for any

practical application.

30142158

FIGURE 8. Total Input Noise

HIGH IMPEDANCE SENSOR INTERFACE

Many sensors have high source impedances that may range

up to 10 MΩ. The output signal of sensors often needs to be

amplified or otherwise conditioned by means of an amplifier.

The input bias current of this amplifier can load the sensor's

output and cause a voltage drop across the source resistance

as shown in Figure 9, where VIN+ = VS – IBIAS*RS

The last term, IBIAS*RS, shows the voltage drop across RS. To

prevent errors introduced to the system due to this voltage,

an op amp with very low input bias current must be used with

high impedance sensors. This is to keep the error contribution

by IBIAS*RS less than the input voltage noise of the amplifier,

so that it will not become the dominant noise factor.

30142159

FIGURE 9. Noise Due to IBIAS

pH electrodes are very high impedance sensors. As their

name indicates, they are used to measure the pH of a solu-

tion. They usually do this by generating an output voltage

which is proportional to the pH of the solution. pH electrodes

are calibrated so that they have zero output for a neutral so-

lution, pH = 7, and positive and negative voltages for acidic

or alkaline solutions. This means that the output of a pH elec-

trode is bipolar and has to be level shifted to be used in a

single supply system. The rate of change of this voltage is

usually shown in mV/pH and is different for different pH sen-

sors. Temperature is also an important factor in a pH elec-

trode reading. The output voltage of the senor will change with

temperature.

Figure 10 shows a typical output voltage spectrum of a pH

electrode. Note that the exact values of output voltage will be

different for different sensors. In this example, the pH elec-

trode has an output voltage of 59.15 mV/pH at 25°C.

30142160

FIGURE 10. Output Voltage of a pH Electrode

The temperature dependence of a typical pH electrode is

shown in Figure 11. As is evident, the output voltage changes

with changes in temperature.

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SM72501

30142161

FIGURE 11. Temperature Dependence of a pH Electrode

The schematic shown in Figure 12 is a typical circuit which

can be used for pH measurement. The LM35 is a precision

integrated circuit temperature sensor. This sensor is differen-

tiated from similar products because it has an output voltage

linearly proportional to Celcius measurement, without the

need to convert the temperature to Kelvin. The LM35 is used

to measure the temperature of the solution and feeds this

reading to the Analog to Digital Converter, ADC. This infor-

mation is used by the ADC to calculate the temperature

effects on the pH readings. The LM35 needs to have a resis-

tor, RT in Figure 12, to –V+ in order to be able to read

temperatures below 0°C. RT is not needed if temperatures are

not expected to go below zero.

The output of pH electrodes is usually large enough that it

does not require much amplification; however, due to the very

high impedance, the output of a pH electrode needs to be

buffered before it can go to an ADC. Since most ADCs are

operated on single supply, the output of the pH electrode also

needs to be level shifted. Amplifier A1 buffers the output of

the pH electrode with a moderate gain of +2, while A2 pro-

vides the level shifting. VOUT at the output of A2 is given by:

VOUT = −2VpH + 1.024V.

The LM4140A is a precision, low noise, voltage reference

used to provide the level shift needed. The ADC used in this

application is the ADC12032 which is a 12-bit, 2 channel con-

verter with multiplexers on the inputs and a serial output. The

12-bit ADC enables users to measure pH with an accuracy of

0.003 of a pH unit. Adequate power supply bypassing and

grounding is extremely important for ADCs. Recommended

bypass capacitors are shown in Figure 12. It is common to

share power supplies between different components in a cir-

cuit. To minimize the effects of power supply ripples caused

by other components, the op amps need to have bypass ca-

pacitors on the supply pins. Using the same value capacitors

as those used with the ADC are ideal. The combination of

these three values of capacitors ensures that AC noise

present on the power supply line is grounded and does not

interfere with the amplifiers' signal.

30142162

FIGURE 12. pH Measurement Circuit

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SM72501

Physical Dimensions inches (millimeters) unless otherwise noted

5-Pin SOT23

NS Package Number MF05A

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SM72501

Notes

SM72501 SolarMagic Precision, CMOS Input, RRIO, Wide Supply Range Amplifier

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