1
®
FN3094.4
ADC0803, ADC0804
8-Bit, Microprocessor-Compatible, A/D
Converters
The ADC080X family are CMOS 8-Bit, successive-
approximation A/D converters which use a modified
potentiometric ladder and are designed to operate with the
8080A control bus via three-state outputs. These converters
appear to the processor as memory locations or I/O ports,
and hence no interfacing logic is required.
The differential analog voltage input has good common-
mode-rejection and permits offsetting the analog zero-input-
voltage value. In addition, the voltage reference input can be
adjusted to allow encoding any smaller analog voltage span
to the full 8 bits of resolution.
Typical Application Schematic
Features
• 80C48 and 80C80/85 Bus Compatible - No Interfacing
Logic Required
• Conversion Time . . . . . . . . . . . . . . . . . . . . . . . . . . <100µs
• Easy Interface to Most Microprocessors
• Will Operate in a “Stand Alone” Mode
• Differential Analog Voltage Inputs
• Works with Bandgap Voltage References
• TTL Compatible Inputs and Outputs
• On-Chip Clock Generator
• Analog Voltage Input Range
(Single + 5V Supply) . . . . . . . . . . . . . . . . . . . . . . 0V to 5V
• No Zero-Adjust Required
• 80C48 and 80C80/85 Bus Compatible - No Interfacing
Logic Required
Pinout
ADC0803, ADC0804
(PDIP)
TOP VIEW
3
2
1
12
11
5
15
14
13
18
17
16
7
6
10
9
8
4
19
20
WR
RD
CS
DB6
DB7
INTR
DB3
DB4
DB5
DB0
DB1
DB2
CLK IN
CLK R
V+
VIN (-)
VIN (+)
DGND
VREF/2
AGND
ANY
µPROCESSOR
8-BIT RESOLUTION
OVER ANY
DESIRED
ANALOG INPUT
VOLTAGE RANGE
DIFF
INPUTS
10K
150pF
VREF/2
µP BUS
+5V
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
WR
RD
CS
CLK IN
INTR
VIN (-)
VIN (+)
DGND
VREF/2
AGND
V+ OR VREF
CLK R
DB0 (LSB)
DB1
DB2
DB3
DB4
DB5
DB6
DB7 (MSB)
Ordering Information
PART NUMBER ERROR EXTERNAL CONDITIONS TEMP. RANGE (oC) PACKAGE PKG. NO
ADC0803LCN ±1/2 LSB VREF/2 Adjusted for Correct Full Scale
Reading
0 to 70 20 Ld PDIP E20.3
ADC0804LCN ±1 LSB VREF/2 = 2.500VDC (No Adjustments) 0 to 70 20 Ld PDIP E20.3
Data Sheet August 2002
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 |Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2002. All Rights Reserved
2
Functional Diagram
1211 151413 181716
WR
RD
CS
INTR
CLK OSC
CLK R
V+
VIN (-)
VIN (+)
DGND
VREF/2
AGND
(VREF)
DAC
VOUT
COMP
CLK
GEN CLKS
CLK A
RESET
START F/F
LADDER
AND
DECODER
SUCCESSIVE
APPROX.
REGISTER
AND LATCH
8-BIT
SHIFT
REGISTER
D
RESET
SET
CONV. COMPL.
THREE-STATE
OUTPUT LATCHES
DIGITAL OUTPUTS
THREE-STATE CONTROL
“1” = OUTPUT ENABLE
DFF2
CLK A
XFER G2
Q
8 X 1/f
R
Q
INTR F/F
IF RESET = “0”
D
DFF1
Q
D
Q
CLK B START
CONVERSION
MSB
LSB
Q
“1” = RESET SHIFT REGISTER
“0” = BUSY AND RESET STATE RESET
READ
SET
3
2
1
5
7
6
10
9
8
4
19
20
CLK IN
MSB
G1
CLK
-
+
LSB
INPUT PROTECTION
FOR ALL LOGIC INPUTS
INPUT
TO INTERNAL
BV = 30V
CIRCUITS
∑
V+
+
-
ADC0803, ADC0804
3
Absolute Maximum Ratings Thermal Information
Supply Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.5V
Voltage at Any Input. . . . . . . . . . . . . . . . . . . . . . -0.3V to (V+ +0.3V)
Operating Conditions
Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Thermal Resistance (Typical, Note 1) θJA (oC/W)
PDIP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80
Maximum Junction Temperature
Plastic Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .150oC
Maximum Storage Temperature Range. . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering, 10s). . . . . . . . . . . . .300oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications (Notes 2, 8)
PARAMETER TEST CONDITIONS MIN TYP MAX UNITS
CONVERTER SPECIFICATIONS V+ = 5V, TA = 25oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0803 VREF/2 Adjusted for Correct Full Scale Reading - - ±1/2LSB
ADC0804 VREF/2 = 2.500V - - ±1LSB
VREF/2 Input Resistance Input Resistance at Pin 9 1.0 1.3 - kΩ
Analog Input Voltage Range (Note 3) GND-0.05 - (V+) + 0.05 V
DC Common-Mode Rejection Over Analog Input Voltage Range - ±1/16 ±1/8LSB
Power Supply Sensitivity V+ = 5V ±10% Over Allowed Input Voltage
Range
-±1/16 ±1/8LSB
CONVERTER SPECIFICATIONS V+ = 5V, 0oC to 70oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0803 VREF/2 Adjusted for Correct Full Scale Reading - - ±1/2LSB
ADC0804 VREF/2 = 2.500V - - ±1LSB
VREF/2 Input Resistance Input Resistance at Pin 9 1.0 1.3 - kΩ
Analog Input Voltage Range (Note 3) GND-0.05 - (V+) + 0.05 V
DC Common-Mode Rejection Over Analog Input Voltage Range - ±1/8±1/4LSB
Power Supply Sensitivity V+ = 5V ±10% Over Allowed Input Voltage
Range
-±1/16 ±1/8LSB
AC TIMING SPECIFICATIONS V+ = 5V, and TA = 25oC, Unless Otherwise Specified
Clock Frequency, fCLK V+ = 6V (Note 4) 100 640 1280 kHz
V+ = 5V 100 640 800 kHz
Clock Periods per Conversion (Note 5),
tCONV
62 - 73 Clocks/Conv
Conversion Rate In Free-Running Mode, CR INTR tied to WR with CS = 0V, fCLK = 640kHz - - 8888 Conv/s
Width of WR Input (Start Pulse Width),
tW(WR)I
CS = 0V (Note 6) 100 - - ns
Access Time (Delay from Falling Edge of
RD to Output Data Valid), tACC
CL = 100pF (Use Bus Driver IC for Larger CL) - 135 200 ns
Three-State Control (Delay from Rising
Edge of RD to Hl-Z State), t1H, t0H
CL = 10pF, RL= 10K
(See Three-State Test Circuits)
- 125 250 ns
Delay from Falling Edge of WR to Reset of
INTR, tWI, tRI
- 300 450 ns
Input Capacitance of Logic Control Inputs,
CIN
-5-pF
Three-State Output Capacitance (Data
Buffers), COUT
-5-pF
ADC0803, ADC0804
4
DC DIGITAL LEVELS AND DC SPECIFICATIONS V+ = 5V, and TMIN to TMAX, Unless Otherwise Specified
CONTROL INPUTS (Note 7)
Logic “1“ Input Voltage (Except Pin 4 CLK
IN), VINH
V+ = 5.25V 2.0 - V+ V
Logic “0“ Input Voltage (Except Pin 4 CLK
IN), VINL
V+ = 4.75V - - 0.8 V
CLK IN (Pin 4) Positive Going Threshold
Voltage, V+CLK
2.7 3.1 3.5 V
CLK IN (Pin 4) Negative Going Threshold
Voltage, V-CLK
1.5 1.8 2.1 V
CLK IN (Pin 4) Hysteresis, VH0.6 1.3 2.0 V
Logic “1” Input Current (All Inputs), IINHI VlN = 5V - 0.005 1 µΑ
Logic “0” Input Current (All Inputs), IINLO VlN = 0V -1 -0.005 - µA
Supply Current (Includes Ladder Current), I+ fCLK = 640kHz, TA = 25oC and CS = Hl - 1.3 2.5 mA
DATA OUTPUTS AND INTR
Logic “0” Output Voltage, VOL lO = 1.6mA, V+ = 4.75V - - 0.4 V
Logic “1” Output Voltage, VOH lO = -360µA, V+ = 4.75V 2.4 - - V
Three-State Disabled Output Leakage (All
Data Buffers), ILO
VOUT = 0V -3 - - µA
VOUT = 5V - - 3 µA
Output Short Circuit Current, ISOURCE VOUT Short to GND, TA = 25oC4.56-mA
Output Short Circuit Current, ISINK VOUT Short to V+, TA = 25oC9.016-mA
NOTES:
2. All voltages are measured with respect to GND, unless otherwise specified. The separate AGND point should always be wired to the DGND,
being careful to avoid ground loops.
3. For VIN(-) ≥ VIN(+) the digital output code will be 0000 0000. Two on-chip diodes are tied to each analog input (see Block Diagram) which will
forward conduct for analog input voltages one diode drop below ground or one diode drop greater than the V+ supply. Be careful, during testing
at low V+ levels (4.5V), as high level analog inputs (5V) can cause this input diode to conduct - especially at elevated temperatures, and cause
errors for analog inputs near full scale. As long as the analog VIN does not exceed the supply voltage by more than 50mV, the output code will
be correct. To achieve an absolute 0V to 5V input voltage range will therefore require a minimum supply voltage of 4.950V over temperature
variations, initial tolerance and loading.
4. With V+ = 6V, the digital logic interfaces are no longer TTL compatible.
5. With an asynchronous start pulse, up to 8 clock periods may be required before the internal clock phases are proper to start the conversion process.
6. The CS input is assumed to bracket the WR strobe input so that timing is dependent on the WR pulse width. An arbitrarily wide pulse width will
hold the converter in a reset mode and the start of conversion is initiated by the low to high transition of the WR pulse (see Timing Diagrams).
7. CLK IN (pin 4) is the input of a Schmitt trigger circuit and is therefore specified separately.
8. None of these A/Ds requires a zero-adjust. However, if an all zero code is desired for an analog input other than 0V, or if a narrow full scale span exists
(for example: 0.5V to 4V full scale) the VIN(-) input can be adjusted to achieve this. See the Zero Error description in this data sheet.
Electrical Specifications (Notes 2, 8) (Continued)
PARAMETER TEST CONDITIONS MIN TYP MAX UNITS
Timing Waveforms
FIGURE 1A. t1H FIGURE 1B. t1H, CL = 10pF
10K
V+
RD
CS
CL
DATA
OUTPUT
RD
2.4V
tr
90%
50%
10%
t1H
0.8V
DATA
OUTPUTS
GND
tr = 20ns
VOH 90%
ADC0803, ADC0804
5
FIGURE 1C. t0H FIGURE 1D. t0H, CL = 10pF
FIGURE 1. THREE-STATE CIRCUITS AND WAVEFORMS
Timing Waveforms (Continued)
10K
V+
RD
CS CL
DATA
OUTPUT
V+
RD
2.4V
tr
90%
50%
10%
t0H
0.8V
DATA
OUTPUTS
VOI
tr = 20ns
V+
10%
Typical Performance Curves
FIGURE 2. LOGIC INPUT THRESHOLD VOLTAGE vs SUPPLY
VOLTAGE
FIGURE 3. DELAY FROM FALLING EDGE OF RD TO OUTPUT
DATA VALID vs LOAD CAPACITANCE
FIGURE 4. CLK IN SCHMITT TRIP LEVELS vs SUPPLY
VOLTAGE
FIGURE 5. fCLK vs CLOCK CAPACITOR
-55oC TO 125oC
1.8
1.7
1.6
1.5
1.4
1.3
4.754.50 5.00 5.25 5.50
V+ SUPPLY VOLTAGE (V)
LOGIC INPUT THRESHOLD VOLTAGE (V)
DELAY (ns)
500
400
300
200
100
0
LOAD CAPACITANCE (pF)
200 400 600 800 1000
CLK IN THRESHOLD VOLTAGE (V)
3.5
3.1
2.7
2.3
1.9
1.5
4.50
V+ SUPPLY VOLTAGE (V)
-55oC TO 125oC
VT(-)
VT(+)
4.75 5.00 5.25 5.50
1000
CLOCK CAPACITOR (pF)
fCLK (kHz)
100
10010 1000
R = 10K
R = 50K
R = 20K
ADC0803, ADC0804
6
FIGURE 6. FULL SCALE ERROR vs fCLK FIGURE 7. EFFECT OF UNADJUSTED OFFSET ERROR
FIGURE 8. OUTPUT CURRENT vs TEMPERATURE FIGURE 9. POWER SUPPLY CURRENT vs TEMPERATURE
Typical Performance Curves (Continued)
FULL SCALE ERROR (LSBs)
7
6
5
4
3
2
1
0
fCLK (kHz)
0 400 800 1200 1600 2000
V+ = 4.5V
V+ = 5V
V+ = 6V
VIN(+) = VIN(-) = 0V
ASSUMES VOS = 2mV
THIS SHOWS THE NEED
FOR A ZERO ADJUSTMENT
IF THE SPAN IS REDUCED
OFFSET ERROR (LSBs)
16
14
12
10
8
6
4
2
VREF/2 (V)
0
0.01 0.1 1.0 5
OUTPUT CURRENT (mA)
8
7
6
5
4
3
2
-50
TA AMBIENT TEMPERATURE (oC)
-ISINK
VOUT = 0.4V
ISOURCE
VOUT = 2.4V
DATA OUTPUT
BUFFERS
V+ = 5V
-25 0 25 50 75 100 125
POWER SUPPLY CURRENT (mA)
TA AMBIENT TEMPERATURE (oC)
-50 -25 0 25 50 75 100 125
1.6
1.5
1.4
1.3
1.2
1.1
1.0
fCLK = 640kHz
V+ = 5.5V
V+ = 5.0V
V+ = 4.5V
Timing Diagrams
FIGURE 10A. START CONVERSION
tWI
tW(WR)I
1 TO 8 x 1/fCLK INTERNAL TC
CS
WR
ACTUAL INTERNAL
STATUS OF THE
CONVERTER
INTR
(LAST DATA READ)
(LAST DATA NOT READ)
“NOT BUSY”
“BUSY”
DATA IS VALID IN
OUTPUT LATCHES
INTR
ASSERTED
tVI 1/2 fCLK
ADC0803, ADC0804
7
FIGURE 10B. OUTPUT ENABLE AND RESET INTR
Timing Diagrams (Continued)
VALID
DATA
VALID
DATA
INTR RESET
INTR
CS
RD
DATA
OUTPUTS
THREE-STATE
(HI-Z)
tRI
tACC t1H , t0H
TRANSFER FUNCTION ERROR PLOT
FIGURE 11A. ACCURACY = ±0 LSB; PERFECT A/D
TRANSFER FUNCTION ERROR PLOT
FIGURE 11B. ACCURACY = ±1/2 LSB
FIGURE 11. CLARIFYING THE ERROR SPECS OF AN A/D CONVERTER
ANALOG INPUT (VIN)
DIGITAL OUTPUT CODE
D + 1
D
D - 1
A + 1
A
A - 1
3
21
56
4
3
2
15
64
ERROR
0
+1 LSB
-1 LSB
-1/2 LSB
+1/2 LSB
*QUANTIZATION ERROR
A
ANALOG INPUT (VIN)
A + 1A - 1
ANALOG INPUT (VIN)
DIGITAL OUTPUT CODE
D + 1
D
D - 1
A + 1
A
A - 1
3
2
1
5
6
4*
0
+1 LSB
-1 LSB
QUANTIZATION
ERROR
3
2
1
6
4
ANALOG INPUT (VIN)
A + 1AA - 1
ERROR
ADC0803, ADC0804
8
Understanding A/D Error Specs
A perfect A/D transfer characteristic (staircase wave-form) is
shown in Figure 11A. The horizontal scale is analog input
voltage and the particular points labeled are in steps of 1
LSB (19.53mV with 2.5V tied to the VREF/2 pin). The digital
output codes which correspond to these inputs are shown as
D-1, D, and D+1. For the perfect A/D, not only will center-
value (A - 1, A, A + 1, . . .) analog inputs produce the correct
output digital codes, but also each riser (the transitions
between adjacent output codes) will be located ±1/2 LSB
away from each center-value. As shown, the risers are ideal
and have no width. Correct digital output codes will be
provided for a range of analog input voltages which extend
±1/2 LSB from the ideal center-values. Each tread (the range
of analog input voltage which provides the same digital
output code) is therefore 1 LSB wide.
The error curve of Figure 11B shows the worst case transfer
function for the ADC080X. Here the specification guarantees
that if we apply an analog input equal to the LSB analog
voltage center-value, the A/D will produce the correct digital
code.
Next to each transfer function is shown the corresponding
error plot. Notice that the error includes the quantization
uncertainty of the A/D. For example, the error at point 1 of
Figure 11A is +1/2LSB because the digital code appeared
1/2 LSB in advance of the center-value of the tread. The
error plots always have a constant negative slope and the
abrupt upside steps are always 1 LSB in magnitude, unless
the device has missing codes.
Detailed Description
The functional diagram of the ADC080X series of A/D
converters operates on the successive approximation
principle (see Application Notes AN016 and AN020 for a
more detailed description of this principle). Analog switches
are closed sequentially by successive-approximation logic
until the analog differential input voltage [VlN(+) - VlN(-)]
matches a voltage derived from a tapped resistor string
across the reference voltage. The most significant bit is
tested first and after 8 comparisons (64 clock cycles), an 8-
bit binary code (1111 1111 = full scale) is transferred to an
output latch.
The normal operation proceeds as follows. On the high-to-low
transition of the WR input, the internal SAR latches and the
shift-register stages are reset, and the INTR output will be set
high. As long as the CS input and WR input remain low, the
A/D will remain in a reset state. Conversion will start from 1 to
8 clock periods after at least one of these inputs makes a low-
to-high transition. After the requisite number of clock pulses to
complete the conversion, the INTR pin will make a high-to-low
transition. This can be used to interrupt a processor, or
otherwise signal the availability of a new conversion. A RD
operation (with CS low) will clear the INTR line high again.
The device may be operated in the free-running mode by
connecting INTR to the WR input with CS = 0. To ensure start-
up under all possible conditions, an external WR pulse is
required during the first power-up cycle. A conversion-in-
process can be interrupted by issuing a second start
command.
Digital Operation
The converter is started by having CS and WR simultaneously
low. This sets the start flip-flop (F/F) and the resulting “1” level
resets the 8-bit shift register, resets the Interrupt (INTR) F/F
and inputs a “1” to the D flip-flop, DFF1, which is at the input
end of the 8-bit shift register. Internal clock signals then
transfer this “1” to the Q output of DFF1. The AND gate, G1,
combines this “1” output with a clock signal to provide a reset
signal to the start F/F. If the set signal is no longer present
(either WR or CS is a “1”), the start F/F is reset and the 8-bit
shift register then can have the “1” clocked in, which starts the
conversion process. If the set signal were to still be present,
this reset pulse would have no effect (both outputs of the start
F/F would be at a “1” level) and the 8-bit shift register would
continue to be held in the reset mode. This allows for
asynchronous or wide CS and WR signals.
After the “1” is clocked through the 8-bit shift register (which
completes the SAR operation) it appears as the input to
DFF2. As soon as this “1” is output from the shift register, the
AND gate, G2, causes the new digital word to transfer to the
Three-State output latches. When DFF2 is subsequently
clocked, the Q output makes a high-to-low transition which
causes the INTR F/F to set. An inverting buffer then supplies
the INTR output signal.
When data is to be read, the combination of both CS and RD
being low will cause the INTR F/F to be reset and the three-
state output latches will be enabled to provide the 8-bit
digital outputs.
Digital Control Inputs
The digital control inputs (CS, RD, and WR) meet standard
TTL logic voltage levels. These signals are essentially
equivalent to the standard A/D Start and Output Enable
control signals, and are active low to allow an easy interface
to microprocessor control busses. For non-microprocessor
based applications, the CS input (pin 1) can be grounded and
the standard A/D Start function obtained by an active low
pulse at the WR input (pin 3). The Output Enable function is
achieved by an active low pulse at the RD input (pin 2).
Analog Operation
The analog comparisons are performed by a capacitive
charge summing circuit. Three capacitors (with precise ratioed
values) share a common node with the input to an auto-
zeroed comparator. The input capacitor is switched between
VlN(+) and VlN(-), while two ratioed reference capacitors are
switched between taps on the reference voltage divider string.
The net charge corresponds to the weighted difference
between the input and the current total value set by the
ADC0803, ADC0804
9
successive approximation register. A correction is made to
offset the comparison by 1/2 LSB (see Figure 11A).
Analog Differential Voltage Inputs and Common-
Mode Rejection
This A/D gains considerable applications flexibility from the
analog differential voltage input. The VlN(-) input (pin 7) can
be used to automatically subtract a fixed voltage value from
the input reading (tare correction). This is also useful in 4mA
- 20mA current loop conversion. In addition, common-mode
noise can be reduced by use of the differential input.
The time interval between sampling VIN(+) and VlN(-) is 41/2
clock periods. The maximum error voltage due to this slight
time difference between the input voltage samples is given by:
where:
∆VE is the error voltage due to sampling delay,
VPEAK is the peak value of the common-mode voltage,
fCM is the common-mode frequency.
For example, with a 60Hz common-mode frequency, fCM, and
a 640kHz A/D clock, fCLK, keeping this error to 1/4 LSB (~5mV)
would allow a common-mode voltage, VPEAK, given by:
,
or
.
The allowed range of analog input voltage usually places
more severe restrictions on input common-mode voltage
levels than this.
An analog input voltage with a reduced span and a relatively
large zero offset can be easily handled by making use of the
differential input (see Reference Voltage Span Adjust).
Analog Input Current
The internal switching action causes displacement currents to
flow at the analog inputs. The voltage on the on-chip
capacitance to ground is switched through the analog
differential input voltage, resulting in proportional currents
entering the VIN(+) input and leaving the VIN(-) input. These
current transients occur at the leading edge of the internal
clocks. They rapidly decay and do not inherently cause errors
as the on-chip comparator is strobed at the end of the clock
perIod.
Input Bypass Capacitors
Bypass capacitors at the inputs will average these charges
and cause a DC current to flow through the output resistances
of the analog signal sources. This charge pumping action is
worse for continuous conversions with the VIN(+) input voltage
at full scale. For a 640kHz clock frequency with the VIN(+)
input at 5V, this DC current is at a maximum of approximately
5µA. Therefore, bypass capacitors should not be used at
the analog inputs or the VREF/2 pin for high resistance
sources (>1kΩ). If input bypass capacitors are necessary for
noise filtering and high source resistance is desirable to
minimize capacitor size, the effects of the voltage drop across
this input resistance, due to the average value of the input
current, can be compensated by a full scale adjustment while
the given source resistor and input bypass capacitor are both
in place. This is possible because the average value of the
input current is a precise linear function of the differential input
voltage at a constant conversion rate.
Input Source Resistance
Large values of source resistance where an input bypass
capacitor is not used will not cause errors since the input
currents settle out prior to the comparison time. If a low-
pass filter is required in the system, use a low-value series
resistor (≤1kΩ) for a passive RC section or add an op amp
RC active low-pass filter. For low-source-resistance
applications (≤1kΩ), a 0.1µF bypass capacitor at the inputs
will minimize EMI due to the series lead inductance of a long
wire. A 100Ω series resistor can be used to isolate this
capacitor (both the R and C are placed outside the feedback
loop) from the output of an op amp, if used.
Stray Pickup
The leads to the analog inputs (pins 6 and 7) should be kept
as short as possible to minimize stray signal pickup (EMI).
Both EMI and undesired digital-clock coupling to these inputs
can cause system errors. The source resistance for these
inputs should, in general, be kept below 5kΩ. Larger values of
source resistance can cause undesired signal pickup. Input
bypass capacitors, placed from the analog inputs to ground,
will eliminate this pickup but can create analog scale errors as
these capacitors will average the transient input switching
currents of the A/D (see Analog Input Current). This scale
error depends on both a large source resistance and the use
of an input bypass capacitor. This error can be compensated
by a full scale adjustment of the A/D (see Full Scale
Adjustment) with the source resistance and input bypass
capacitor in place, and the desired conversion rate.
Reference Voltage Span Adjust
For maximum application flexibility, these A/Ds have been
designed to accommodate a 5V, 2.5V or an adjusted voltage
reference. This has been achieved in the design of the IC as
shown in Figure 12.
Notice that the reference voltage for the IC is either 1/2 of the
voltage which is applied to the V+ supply pin, or is equal to
the voltage which is externally forced at the VREF/2 pin. This
allows for a pseudo-ratiometric voltage reference using, for
the V+ supply, a 5V reference voltage. Alternatively, a
voltage less than 2.5V can be applied to the VREF/2 input.
The internal gain to the VREF/2 input is 2 to allow this factor
of 2 reduction in the reference voltage.
VEMAX()∆VPEAK
()2πfCM
()
4.5
fCLK
------------=
VPEAK
∆VEMAX()fCLK
()
2πfCM
()4.5()
--------------------------------------------------=
VPEAK
510
3–
×()640 103
×()
6.28()60()4.5()
---------------------------------------------------------- 1.9V≅=
ADC0803, ADC0804
10
Such an adjusted reference voltage can accommodate a
reduced span or dynamic voltage range of the analog input
voltage. If the analog input voltage were to range from 0.5V to
3.5V, instead of 0V to 5V, the span would be 3V. With 0.5V
applied to the VlN(-) pin to absorb the offset, the reference
voltage can be made equal to 1/2 of the 3V span or 1.5V. The
A/D now will encode the VlN(+) signal from 0.5V to 3.5V with
the 0.5V input corresponding to zero and the 3.5V input
corresponding to full scale. The full 8 bits of resolution are
therefore applied over this reduced analog input voltage
range. The requisite connections are shown in Figure 13. For
expanded scale inputs, the circuits of Figures 14 and 15 can
be used.
Reference Accuracy Requirements
The converter can be operated in a pseudo-ratiometric mode
or an absolute mode. In ratiometric converter applications,
the magnitude of the reference voltage is a factor in both the
output of the source transducer and the output of the A/D
converter and therefore cancels out in the final digital output
code. In absolute conversion applicatIons, both the initial
value and the temperature stability of the reference voltage
are important accuracy factors in the operation of the A/D
converter. For VREF/2 voltages of 2.5V nominal value, initial
errors of ±10mV will cause conversion errors of ±1 LSB due
to the gain of 2 of the VREF/2 input. In reduced span
applications, the initial value and the stability of the VREF/2
input voltage become even more important. For example, if
the span is reduced to 2.5V, the analog input LSB voltage
value is correspondingly reduced from 20mV (5V span) to
10mV and 1 LSB at the VREF/2 input becomes 5mV. As can
be seen, this reduces the allowed initial tolerance of the
reference voltage and requires correspondingly less
absolute change with temperature variations. Note that
spans smaller than 2.5V place even tighter requirements on
the initial accuracy and stability of the reference source.
In general, the reference voltage will require an initial
adjustment. Errors due to an improper value of reference
voltage appear as full scale errors in the A/D transfer
FIGURE 12. THE VREFERENCE DESIGN ON THE IC
FIGURE 13. OFFSETTING THE ZERO OF THE ADC080X AND
PERFORMING AN INPUT RANGE (SPAN)
ADJUSTMENT
V+
DGND
VREF/2
AGND
(VREF)
R
R
DIGITAL
CIRCUITS
ANALOG
CIRCUITS
9
810
20
DECODE
300
TO VREF/2
TO VIN(-)
ZERO SHIFT VOLTAGE
0.1µF
5V
-
+
VREF
(5V)
FS
ADJ.
“SPAN”/2
ICL7611
FIGURE 14. HANDLING ±10V ANALOG INPUT RANGE
FIGURE 15. HANDLING ±5V ANALOG INPUT RANGE
VIN(-)
2R
5V
2R
VIN ± 10V
R
VIN(+)
(VREF)
V+ 20
10µF
6
7
+
ADC0803-
ADC0804
VIN(-)
R
5V
VIN ±5V
R
VIN(+)
(VREF)
V+ 20
10µF
6
7
+
ADC0803-
ADC0804
ADC0803, ADC0804
11
function. IC voltage regulators may be used for references if
the ambient temperature changes are not excessive.
Zero Error
The zero of the A/D does not require adjustment. If the
minimum analog input voltage value, VlN(MlN), is not ground, a
zero offset can be done. The converter can be made to output
0000 0000 digital code for this minimum input voltage by
biasing the A/D VIN(-) input at this VlN(MlN) value (see
Applications section). This utilizes the differential mode
operation of the A/D.
The zero error of the A/D converter relates to the location of
the first riser of the transfer function and can be measured by
grounding the VIN(-) input and applying a small magnitude
positive voltage to the VIN(+) input. Zero error is the difference
between the actual DC input voltage which is necessary to
just cause an output digital code transition from 0000 0000 to
0000 0001 and the ideal 1/2 LSB value (1/2 LSB = 9.8mV for
VREF/2 = 2.500V).
Full Scale Adjust
The full scale adjustment can be made by applying a
differential input voltage which is 11/2 LSB down from the
desired analog full scale voltage range and then adjusting
the magnitude of the VREF/2 input (pin 9) for a digital output
code which is just changing from 1111 1110 to 1111 1111.
When offsetting the zero and using a span-adjusted VREF/2
voltage, the full scale adjustment is made by inputting VMlN
to the VIN(-) input of the A/D and applying a voltage to the
VIN(+) input which is given by:
,
where:
VMAX = the high end of the analog input range, and
VMIN = the low end (the offset zero) of the analog range.
(Both are ground referenced.)
Clocking Option
The clock for the A/D can be derived from an external source
such as the CPU clock or an external RC network can be
added to provIde self-clocking. The CLK IN (pin 4) makes
use of a Schmitt trigger as shown in Figure 16.
Heavy capacitive or DC loading of the CLK R pin should be
avoided as this will disturb normal converter operation.
Loads less than 50pF, such as driving up to 7 A/D converter
clock inputs from a single CLK R pin of 1 converter, are
allowed. For larger clock line loading, a CMOS or low power
TTL buffer or PNP input logic should be used to minimize the
loading on the CLK R pin (do not use a standard TTL buffer).
Restart During a Conversion
If the A/D is restarted (CS and WR go low and return high)
during a conversion, the converter is reset and a new
conversion is started. The output data latch is not updated if
the conversion in progress is not completed. The data from
the previous conversion remain in this latch.
Continuous Conversions
In this application, the CS input is grounded and the WR
input is tied to the INTR output. This WR and INTR node
should be momentarily forced to logic low following a power-
up cycle to insure circuit operation. See Figure 17 for details.
Driving the Data Bus
This CMOS A/D, like MOS microprocessors and memories,
will require a bus driver when the total capacitance of the
data bus gets large. Other circuItry, which is tied to the data
bus, will add to the total capacitive loading, even in three-
state (high-impedance mode). Back plane busing also
greatly adds to the stray capacitance of the data bus.
There are some alternatives available to the designer to
handle this problem. Basically, the capacitive loading of the
data bus slows down the response time, even though DC
specifications are still met. For systems operating with a
relatively slow CPU clock frequency, more time is available
in which to establish proper logic levels on the bus and
therefore higher capacitive loads can be driven (see Typical
Performance Curves).
At higher CPU clock frequencies time can be extended for
I/O reads (and/or writes) by inserting wait states (8080) or
using clock-extending circuits (6800).
VIN +()
fSADJ VMAX 1.5 VMAX VMIN
–()
256
-----------------------------------------–=
CLK R
4
CLK IN
CLK
ADC0803-
ADC0804
fCLK ≅
19
R
C
1
1.1 RC
R ≅ 10kΩ
FIGURE 16. SELF-CLOCKING THE A/D
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
ADC0803 - ADC0804
WR
RD
CS
INTR
CLK IN
VIN (-)
VIN (+)
DGND
VREF/2
AGND
DB1
DB0
DB4
DB3
DB2
DB7
DB6
DB5
CLK R
V+
10K 5V (VREF)
10µF
+
DATA
START
ANALOG
INPUTS
150pF
OUTPUTS
N.O.
MSB
LSB
FIGURE 17. FREE-RUNNING CONNECTION
ADC0803, ADC0804
12
Finally, if time is short and capacitive loading is high, external
bus drivers must be used. These can be three-state buffers
(low power Schottky is recommended, such as the 74LS240
series) or special higher-drive-current products which are
designed as bus drivers. High-current bipolar bus drivers
with PNP inputs are recommended.
Power Supplies
Noise spikes on the V+ supply line can cause conversion
errors as the comparator will respond to this noise. A
low-inductance tantalum filter capacitor should be used
close to the converter V+ pin, and values of 1µF or greater
are recommended. If an unregulated voltage is available in
the system, a separate 5V voltage regulator for the converter
(and other analog circuitry) will greatly reduce digital noise
on the V+ supply. An lCL7663 can be used to regulate such
a supply from an input as low as 5.2V.
Wiring and Hook-Up Precautions
Standard digital wire-wrap sockets are not satisfactory for
breadboarding with this A/D converter. Sockets on PC
boards can be used. All logic signal wires and leads should
be grouped and kept as far away as possible from the
analog signal leads. Exposed leads to the analog inputs can
cause undesired digital noise and hum pickup; therefore,
shielded leads may be necessary in many applications.
A single-point analog ground should be used which is
separate from the logic ground points. The power supply
bypass capacitor and the self-clockIng capacitor (if used)
should both be returned to digital ground. Any VREF/2
bypass capacitors, analog input filter capacitors, or input
signal shielding should be returned to the analog ground
point. A test for proper grounding is to measure the zero
error of the A/D converter. Zero errors in excess of 1/4 LSB
can usually be traced to improper board layout and wiring
(see Zero Error for measurement). Further information can
be found in Application Note AN018.
Testing the A/D Converter
There are many degrees of complexity associated with testing
an A/D converter. One of the simplest tests is to apply a
known analog input voltage to the converter and use LEDs to
display the resulting digital output code as shown in Figure 18.
For ease of testing, the VREF/2 (pin 9) should be supplied
with 2.560V and a V+ supply voltage of 5.12V should be
used. This provides an LSB value of 20mV.
If a full scale adjustment is to be made, an analog input
voltage of 5.090V (5.120 - 11/2 LSB) should be applied to
the VIN(+) pin with the VIN(-) pin grounded. The value of the
VREF/2 input voltage should be adjusted until the digital
output code is just changing from 1111 1110 to 1111 1111.
This value of VREF/2 should then be used for all the tests.
The digital-output LED display can be decoded by dividing
the 8 bits into 2 hex characters, one with the 4 most-
significant bits (MS) and one with the 4 least-significant bits
(LS). The output is then interpreted as a sum of fractions
times the full scale voltage:
.
For example, for an output LED display of 1011 0110, the
MS character is hex B (decimal 11) and the LS character is
hex (and decimal) 6, so:
.
Figures 19 and 20 show more sophisticated test circuits.
Typical Applications
Interfacing 8080/85 or Z-80 Microprocessors
VOUT
MS
16
---------LS
256
----------+
5.12()V=
START
VIN (+)
DGND
2.560V
AGND
10µF
150pF
N.O.
0.1µF
0.1µF
TANTALUM
5.120V
5V
1.3kΩLEDs
(8) (8)
MSB
LSB
10kΩ
VREF/2
+
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
ADC0803-
ADC0804
FIGURE 18. BASIC TESTER FOR THE A/D
VOUT 11
16
------ 6
256
----------+
5.12()3.64V==
ANALOG
INPUTS
“A”
R
“B”
R
R
R
“C”
100R
-
+A2
8-BIT
A/D UNDER
TEST
10-BIT
DAC
VANALOG OUTPUT
100X ANALOG
-
+
A1
ERROR VOLTAGE
FIGURE 19. A/D TESTER WITH ANALOG ERROR OUTPUT. THIS
CIRCUIT CAN BE USED TO GENERATE “ERROR
PLOTS” OF FIGURE 11.
A/D UNDER
TEST
10-BIT
DAC
DIGITAL
VANALOG
INPUTS
DIGITAL
OUTPUTS
FIGURE 20. BASIC “DIGITAL” A/D TESTER
ADC0803, ADC0804
13
This converter has been designed to directly interface with
8080/85 or Z-80 Microprocessors. The three-state output
capability of the A/D eliminates the need for a peripheral
interface device, although address decoding is still required
to generate the appropriate CS for the converter. The A/D
can be mapped into memory space (using standard
memory-address decoding for CS and the MEMR and
MEMW strobes) or it can be controlled as an I/O device by
using the I/OR and I/OW strobes and decoding the address
bits A0 → A7 (or address bits A8 → A15, since they will
contain the same 8-bit address information) to obtain the CS
input. Using the I/O space provides 256 additional
addresses and may allow a simpler 8-bit address decoder,
but the data can only be input to the accumulator. To make
use of the additional memory reference instructions, the A/D
should be mapped into memory space. See AN020 for more
discussion of memory-mapped vs I/O-mapped interfaces. An
example of an A/D in I/O space is shown in Figure 21.
The standard control-bus signals of the 8080 (CS, RD and
WR) can be directly wired to the digital control inputs of the
A/D, since the bus timing requirements, to allow both starting
the converter, and outputting the data onto the data bus, are
met. A bus driver should be used for larger microprocessor
systems where the data bus leaves the PC board and/or
must drive capacitive loads larger than 100pF.
It is useful to note that in systems where the A/D converter is
1 of 8 or fewer I/O-mapped devices, no address-decoding
circuitry is necessary. Each of the 8 address bits (A0 to A7)
can be directly used as CS inputs, one for each I/O device.
Interfacing the Z-80 and 8085
The Z-80 and 8085 control buses are slightly different from
that of the 8080. General RD and WR strobes are provided
and separate memory request, MREQ, and I/O request,
IORQ, signals have to be combined with the generalized
strobes to provide the appropriate signals. An advantage of
operating the A/D in I/O space with the Z-80 is that the CPU
will automatically insert one wait state (the RD and WR
strobes are extended one clock period) to allow more time
for the I/O devices to respond. Logic to map the A/D in I/O
space is shown in Figure 22. By using MREQ in place of
IORQ, a memory-mapped configuration results.
Additional I/O advantages exist as software DMA routines are
available and use can be made of the output data transfer
which exists on the upper 8 address lines (A8 to A15) during
I/O input instructions. For example, MUX channel selection for
the A/D can be accomplished with this operating mode.
The 8085 also provides a generalized RD and WR strobe, with
an IO/M line to distinguish I/O and memory requests. The circuit
of Figure 22 can again be used, with IO/M in place of IORQ for
a memory-mapped interface, and an extra inverter (or the logic
equivalent) to provide IO/M for an I/O-mapped connection.
Interfacing 6800 Microprocessor Derivatives (6502,
etc.)
The control bus for the 6800 microprocessor derivatives does
not use the RD and WR strobe signals. Instead it employs a
single R/W line and additional timing, if needed, can be derived
from the φ2 clock. All I/O devices are memory-mapped in the
6800 system, and a special signal, VMA, indicates that the
current address is valid. Figure 23 shows an interface
schematic where the A/D is memory-mapped in the 6800
system. For simplicity, the CS decoding is shown using 1/2
DM8092. Note that in many 6800 systems, an already decoded
4/5 line is brought out to the common bus at pin 21. This can be
tied directly to the CS pin of the A/D, provided that no other
devices are addressed at HEX ADDR: 4XXX or 5XXX.
In Figure 24 the ADC080X series is interfaced to the MC6800
microprocessor through (the arbitrarily chosen) Port B of the
MC6820 or MC6821 Peripheral Interface Adapter (PlA). Here
the CS pin of the A/D is grounded since the PlA is already
memory-mapped in the MC6800 system and no CS decoding
is necessary. Also notice that the A/D output data lines are
connected to the microprocessor bus under program control
through the PlA and therefore the A/D RD pin can be grounded.
Application Notes
NOTE # DESCRIPTION
AN016 “Selecting A/D Converters”
AN018 “Do’s and Don’ts of Applying A/D Converters”
AN020 “A Cookbook Approach to High Speed Data Acquisition
and Microprocessor Interfacing”
AN030 “The ICL7104 - A Binary Output A/D Converter for
Microprocessors”
ADC0803, ADC0804
14
NOTE: Pin numbers for 8228 System Controller: Others are 8080A.
FIGURE 21. ADC080X TO 8080A CPU INTERFACE
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
ADC0803 - ADC0804
WR
RD
CS
INTR
CLK IN
VIN (-)
VIN (+)
DGND
VREF/2
AGND
DB1
DB0
DB4
DB3
DB2
DB7
DB6
DB5
CLK R
V+
10K
5V 10µF
+
ANALOG
INPUTS
150pF
MSB
LSB
DB1 (16) (NOTE)
DB0 (13) (NOTE)
DB4 (5) (NOTE)
DB3 (9) (NOTE)
DB2 (11) (NOTE)
DB7 (7) (NOTE)
DB6 (20) (NOTE)
DB5 (18) (NOTE)
5V
AD15 (36)
AD14 (39)
AD13 (38)
AD12 (37)
AD11 (40)
AD10 (1)
8131
BUS
COMPARATOR
INT (14)
I/O RD (25) (NOTE)
I/O WR (27) (NOTE)
T5
T4
T3
T2
T1
T0
B5
B4
B3
B2
B1
B0
V+OUT
ADC0803, ADC0804
15
FIGURE 22. MAPPING THE A/D AS AN
I/O DEVICE FOR USE
WITH THE Z-80 CPU
FIGURE 23. ADC080X TO MC6800 CPU INTERFACE
FIGURE 24. ADC080X TO MC6820 PIA INTERFACE
WR
RD
IORQ
RD
WR
74C32
ADC0803-
ADC0804
3
2
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
ADC0803 - ADC0804
WR
RD
CS
INTR
CLK IN
VIN (-)
VIN (+)
DGND
VREF/2
AGND
DB1
DB0
DB4
DB3
DB2
DB7
DB6
DB5
CLK R
V+
10K
5V (8)
10µF
+
ANALOG
INPUTS
150pF
MSB
LSB
D1 (32) [29]
D0 (33) [31]
D4 (29) [32]
D3 (30) [H]
D2 (31) [K]
D7 (26) [J]
D6 (27) [L]
D5 (28) [30]
A12 (22) [34]
A13 (23) [N]
A14 (24) [M]
A15 (25) [33]
VMA (5) [F]
IRQ (4)† [D] ††
R/W (34) [6]
1
2
3
4
5
6
1/2 DM8092
A B C
1 2 3
†Numbers in parentheses refer to MC6800 CPU Pinout.
†† Numbers or letters in brackets refer to standard MC6800 System Common Bus Code.
11
12
13
14
15
16
17
18
20
19
10
9
8
7
6
5
4
3
2
1
ADC0803 - ADC0804
WR
RD
CS
INTR
CLK IN
VIN (-)
VIN (+)
DGND
VREF/2
AGND
DB1
DB0
DB4
DB3
DB2
DB7
DB6
DB5
CLK R
V+
10K
5V
ANALOG
INPUTS
150pF
MSB
LSB
11
10
14
13
12
17
16
15
PB1
PB0
PB4
PB3
PB2
PB7
PB6
PB5
MC6820
(MCS6520)
PIA
CB2
CB1
19
18
ADC0803, ADC0804
16
Die Characteristics
DIE DIMENSIONS
101 mils x 93 mils
METALLIZATION
Type: Al
Thickness: 10kÅ ±1kÅ
PASSIVATION
Type: Nitride over Silox
Nitride Thickness: 8kÅ
Silox Thickness: 7kÅ
Metallization Mask Layout
ADC0803, ADC0804
WR
RD
CS
CLK ININTRVIN (-) VIN (+)
DGND
VREF/2
AGND
V+ OR VREF
CLK R
DB0
DB1
DB2
DB3
DB4
DB5
DB6
DB7 (MSB)
V+ OR VREF
ADC0803, ADC0804
17
All Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at website www.intersil.com/quality/iso.asp.
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice.
Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. How-
ever, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No
license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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ADC0803, ADC0804
Dual-In-Line Plastic Packages (PDIP)
NOTES:
1. Controlling Dimensions: INCH. In case of conflict between English
and Metric dimensions, the inch dimensions control.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication No. 95.
4. Dimensions A, A1 and L are measured with the package seated in
JEDEC seating plane gauge GS-3.
5. D, D1, and E1 dimensions do not include mold flash or protrusions.
Mold flash or protrusions shall not exceed 0.010 inch (0.25mm).
6. E and are measured with the leads constrained to be perpen-
dicular to datum .
7. eB and eC are measured at the lead tips with the leads uncon-
strained. eC must be zero or greater.
8. B1 maximum dimensions do not include dambar protrusions. Dam-
bar protrusions shall not exceed 0.010 inch (0.25mm).
9. N is the maximum number of terminal positions.
10. Corner leads (1, N, N/2 and N/2 + 1) for E8.3, E16.3, E18.3, E28.3,
E42.6 will have a B1 dimension of 0.030 - 0.045 inch (0.76 - 1.14mm).
eA-C-
C
L
E
eA
C
eB
eC
-B-
E1
INDEX 12 3 N/2
N
AREA
SEATING
BASE
PLANE
PLANE
-C-
D1
B1
B
e
D
D1
A
A2
L
A1
-A-
0.010 (0.25) C AMBS
E20.3 (JEDEC MS-001-AD ISSUE D)
20 LEAD DUAL-IN-LINE PLASTIC PACKAGE
SYMBOL
INCHES MILLIMETERS
NOTESMIN MAX MIN MAX
A - 0.210 - 5.33 4
A1 0.015 - 0.39 - 4
A2 0.115 0.195 2.93 4.95 -
B 0.014 0.022 0.356 0.558 -
B1 0.045 0.070 1.55 1.77 8
C 0.008 0.014 0.204 0.355 -
D 0.980 1.060 24.89 26.9 5
D1 0.005 - 0.13 - 5
E 0.300 0.325 7.62 8.25 6
E1 0.240 0.280 6.10 7.11 5
e 0.100 BSC 2.54 BSC -
eA0.300 BSC 7.62 BSC 6
eB- 0.430 - 10.92 7
L 0.115 0.150 2.93 3.81 4
N20 209
Rev. 0 12/93
