THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 D D D D D D D THS3001 D AND DGN PACKAGE (TOP VIEW) High Speed - 420 MHz Bandwidth (G = 1, -3 dB) - 6500 V/s Slew Rate - 40-ns Settling Time (0.1%) High Output Drive, IO = 100 mA Excellent Video Performance - 115 MHz Bandwidth (0.1 dB, G = 2) - 0.01% Differential Gain - 0.02 Differential Phase Low 3-mV (max) Input Offset Voltage Very Low Distortion - THD = -96 dBc at f = 1 MHz - THD = -80 dBc at f = 10 MHz Wide Range of Power Supplies - VCC = 4.5 V to 16 V Evaluation Module Available NULL IN - IN + VCC- 1 8 2 7 3 6 4 5 THS3002 D AND DGN PACKAGE (TOP VIEW) NULL VCC+ OUT NC 1OUT 1IN - 1IN + -VCC 1 8 2 7 3 6 4 5 VCC+ 2OUT 2IN- 2IN+ NC - No internal connection The THS3001 implemented in the DGN package is in the product preview stage of development. Contact your local TI sales office for availability. OUTPUT AMPLITUDE vs FREQUENCY 8 VCC = 15 V RF = 680 7 6 The THS300x is a high-speed current-feedback operational amplifier, ideal for communication, imaging, and high-quality video applications. This device offers a very fast 6500-V/s slew rate, a 420-MHz bandwidth, and 40-ns settling time for large-signal applications requiring excellent transient response. In addition, the THS300x operates with a very low distortion of - 96 dBc, making it well suited for applications such as wireless communication basestations or ultrafast ADC or DAC buffers. Output Amplitude - dB description 5 VCC = 5 V RF = 750 4 3 2 1 0 G=2 RL = 150 VI = 200 mV RMS -1 100k 1M 10M 100M 1G f - Frequency - Hz HIGH-SPEED AMPLIFIER FAMILY SUPPLY VOLTAGE ARCHITECTURE DEVICE VFB THS4001 THS4011/12 THS4031/32 THS4061/62 CFB 5V * THS3001/02 * * * * * 5 V 15 V * * * * * * * * * * BW (MHz) SR (V/s) THD f = 1 MHz (dB) ts 0.1% (ns) DIFF. GAIN DIFF. PHASE Vn (nV/Hz) 420 6500 -96 40 0.01% 0.02 1.6 270 400 -72 40 0.04% 0.15 12.5 290 310 -80 37 0.006% 0.01 7.5 100 100 -72 60 0.02% 0.03 1.6 180 400 -72 40 0.02% 0.02 14.5 CAUTION: The THS300x provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss of functionality. Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. Copyright 1999, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 1 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 AVAILABLE OPTIONS PACKAGED DEVICE SOIC (D) TA EVALUATION MODULE MSOP (DGN) 0C to 70C DEVICE THS3001CDGN THS3002CDGN SYMBOL THS3001CD THS3002CD TIADP TIADI THS3001EVM THS3002EVM - 40C to 85C THS3001ID THS3002ID THS3001IDGN THS3002IDGN TIADQ TIADJ -- The D package is available taped and reeled. Add an R suffix to the device type (i.e., THS3001CDR) Product Preview absolute maximum ratings over operating free-air temperature range (unless otherwise noted) Supply voltage, VCC+ to VCC- . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VCC Output Current, IO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175 mA Differential input voltage, VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table Operating free-air temperature, TA, THS300xC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to 70C THS300xI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . - 40C to 85C Storage temperature, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -65C to 125C Lead temperature, 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300C Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE PACKAGE TA 25C POWER RATING DERATING FACTOR ABOVE TA = 25C TA = 70C POWER RATING TA = 85C POWER RATING D 740 mW 6 mW/C 470 mW 380 mW recommended operating conditions MIN Supply voltage voltage, VCC+ CC and VCC- CC Operating free-air free air temperature, temperature TA 2 MAX 16 Single supply 9 32 THS300xC 0 70 -40 85 THS300xI POST OFFICE BOX 655303 NOM 4.5 Split supply * DALLAS, TEXAS 75265 UNIT V C THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 electrical characteristics, TA = 25C, RL = 150 , RF = 1 k (unless otherwise noted) TEST CONDITIONS PARAMETER VCC Split supply Power supply operating range Single supply VCC = 5 V ICC Quiescent current VCC = 15 V VCC = 5 V VO Output voltage swing VCC = 15 V IO Output current (see Note 1) VCC = 5 V, VCC = 15 V, VIO Input offset voltage VCC = 5 V or 15 V Input offset voltage drift VCC = 5 V or 15 V VCC = 5 V or 15 V Input In ut bias current +Input VICR Open O en loop loo transresistance CMRR PSRR 33 5.5 TA = full range TA = 25C 6.6 TA = full range RL = 150 Common-mode rejection ratio 3.2 3.3 RL = 150 12.1 12.8 RL = 1 k 12.8 13.1 RL = 20 9 mA V 100 85 TA = 25C TA = full range mA 120 1 3 4 2 TA = full range 10 15 TA = 25C TA = full range 1 mV V/C 5 10 A 15 3 3.2 12.9 13.2 VCC = 5 V, RL = 1 k VO = 2.5 V, 1.3 VCC = 15 V, RL = 1 k VO = 7.5 V, 2.4 V M VCC = 5 V, VCM = 2.5 V 62 70 VCC = 15 V, VCM = 10 V 65 73 76 VCC = 5 V TA = 25C TA = full range 65 VCC = 15 V TA = 25C TA = full range 69 Power supply rejection ratio V 10 3 RL = 75 UNIT 7.5 8.5 2.9 RL = 1 k MAX 9 TA = 25C VCC = 5 V VCC = 15 V Common mode input voltage range Common-mode TYP 16.5 TA = 25C -Input -In ut IIB MIN 4.5 63 76 67 dB dB dB +Input 1.5 M -Input 15 7.5 pF 10 1.6 nV/Hz RI Input resistance CI Differential input capacitance RO Output resistance Open loop at 5 MHz Vn Input voltage noise VCC = 5 V or 15 V, G=2 f = 10 kHz, In Input current noise VCC = 5 V or 15 V,, G=2 f = 10 kHz,, Positive (IN+) Negative (IN-) 13 16 pA/Hz Full range = 0C to 70C for the THS300xC and - 40C to 85C for the THS300xI. NOTE 1: Observe power dissipation ratings to keep the junction temperature below absolute maximum when the output is heavily loaded or shorted. See absolute maximum ratings section. POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 3 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 operating characteristics, TA = 25C, RL = 150 , RF = 1 k (unless otherwise noted) PARAMETER SR TEST CONDITIONS AD D MAX UNIT VCC = 5 V,, VO(PP) = 4 V G=5 1300 VCC = 15 V,, VO(PP) = 20 V G = -5 6500 G=5 6300 Settling time to 0.1% VCC = 15 V, 0 V to 10 V Step Gain = -1, Settling time to 0.1% VCC = 5 V, 0 V to 2 V Step, Gain = -1, Total harmonic distortion VCC = 15 V, fc = 10 MHz, VO(PP) = 2 V, G=2 VCC = 5 V 0.015% Differential gain error G = 2, 40 IRE modulation,, 100 IRE Ramp, NTSC and PAL VCC = 15 V 0.01% G = 2, 40 IRE modulation,, 100 IRE Ramp, NTSC and PAL VCC = 5 V 0.01 VCC = 15 V 0.02 VCC = 5 V, VCC = 15 V, 330 MHz 420 MHz VCC = 5 V VCC = 15 V 300 350 G = 2, RF = 750 , VCC = 15 V VCC = 5 V G = 2, RF = 680 , VCC = 15 V 115 VCC = 5 V, VO(PP) = 4 V V, RL = 500 G = -5 65 MHz G=5 62 MHz VCC = 15 V, VO(PP) = 20 V RL = 500 G = -5 32 MHz G=5 31 MHz Slew rate (see Note 2) Differential phase error G=1 1, RF = 1 k k, G = 2, RF = 750 , Small signal bandwidth (-3 dB) G = 2, RF = 680 , G = 5, RF = 560 , BW Bandwidth for 0.1 0 1 dB flatness Full power ower bandwidth (see Note 3) Crosstalk (THS3002 only) PARAMETER MEASUREMENT INFORMATION RG RF VCC+ - VO + VI 50 VCC- RL Figure 1. Test Circuit, Gain = 1 + (RF/RG) POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 V/s 40 ns 25 - 80 385 85 TBD NOTES: 2. Slew rate is measured from an output level range of 25% to 75%. 3. Full power bandwidth is defined as the frequency at which the output has 3% THD. 4 TYP 1700 ts THD MIN G = -5 dBc MHz MHz dB THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS Table of Graphs FIGURE |VO| Output voltage swing vs Free-air temperature 2 ICC IIB Current supply vs Free-air temperature 3 Input bias current vs Free-air temperature 4 VIO Input offset voltage vs Free-air temperature 5 vs Common-mode input voltage 6 vs Common-mode input voltage 7 vs Frequency 8 CMRR Vn In PSRR SR Common-mode rejection ratio Transresistance vs Free-air temperature 9 Closed-loop output impedance vs Frequency 10 Voltage noise vs Frequency 11 Current noise vs Frequency 11 vs Frequency 12 vs Free-air temperature 13 Power supply rejection ratio vs Supply voltage Slew rate vs Output step peak-to-peak Normalized slew rate vs Gain 14 15, 16 17 vs Peak-to-peak output voltage swing 18, 19 vs Frequency 20, 21 Differential gain vs Loading 22, 23 Differential phase vs Loading 24, 25 Output amplitude vs Frequency 26-30 Normalized output response vs Frequency 31-34 Harmonic distortion Small and large signal frequency response 35, 36 Small signal pulse response 37, 38 Large signal pulse response 39 - 46 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 5 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS OUTPUT VOLTAGE SWING vs FREE-AIR TEMPERATURE 14 9 VCC = 15 V No Load 8 13 VCC = 15 V RL = 150 12.5 I CC - Supply Current - mA VO - Output Voltage Swing - V 13.5 CURRENT SUPPLY vs FREE-AIR TEMPERATURE 12 4 3.5 VCC = 5 V No Load 3 VCC = 5 V RL = 150 2.5 2 -40 -20 0 20 40 60 VCC = 15 V 7 6 VCC = 10 V 5 VCC = 5 V 4 80 3 -40 100 -20 TA - Free-Air Temperature - C 0 Figure 2 80 100 0 VCC = 5 V -1 IIB+ -0.2 VIO - Input Offset Voltage - mV I IB - Input Bias Current - A 60 INPUT OFFSET VOLTAGE vs FREE-AIR TEMPERATURE -0.5 VCC = 15 V IIB+ -1.5 VCC = 5 V -2 IIB- VCC = 15 V -2.5 IIB- -20 0 20 40 60 80 VCC = 5 V -0.4 -0.6 VCC = 15 V -0.8 -1 Gain = 1 RF = 1 k 100 -1.2 -40 -20 TA - Free-Air Temperature - C 0 20 Figure 5 POST OFFICE BOX 655303 40 60 TA - Free-Air Temperature - C Figure 4 6 40 Figure 3 INPUT BIAS CURRENT vs FREE-AIR TEMPERATURE -3 -40 20 TA - Free-Air Temperature - C * DALLAS, TEXAS 75265 80 100 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS COMMON-MODE REJECTION RATIO vs COMMON-MODE INPUT VOLTAGE COMMON-MODE REJECTION RATIO vs COMMON-MODE INPUT VOLTAGE 80 TA = -40C 70 CMRR - Common-Mode Rejection Ratio - dB CMRR - Common-Mode Rejection Ratio - dB 80 TA = 85C TA = 25C 60 50 40 VCC = 15 V 30 0 2 4 6 8 10 12 TA = -40C 70 TA = 85C TA = 25C 60 50 40 30 VCC = 5 V 20 14 0 |VIC| - Common-Mode Input Voltage - V 0.5 1 Figure 6 3.5 4 2.6 VCC = 15 V 2.4 60 Transresistance - M CMRR - Common-Mode Rejection Ratio - dB 3 2.8 VCC = 15 V VCC = 5 V 50 40 30 1 k 2.2 2 VCC = 10 V 1.8 1.6 1 k 20 - + VI 0 1k 2.5 TRANSRESISTANCE vs FREE-AIR TEMPERATURE 70 10 2 Figure 7 COMMON-MODE REJECTION RATIO vs FREQUENCY 80 1.5 |VIC| - Common-Mode Input Voltage - V 1 k 10k 1.4 VO 1 k 100k 1.2 1M 10M 100M 1 -40 VCC = 5 V VO = VCC/2 RL = 1 k -20 0 20 40 60 80 100 TA - Free-Air Temperature - C f - Frequency - Hz Figure 8 Figure 9 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 7 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY 10 1000 VCC = 15 V RF = 750 Gain = +2 TA = 25C VI(PP) = 2 V Vn - Voltage Noise - nV/ Hz and I n - Current Noise - pA/ Hz Closed-Loop Output Impedance - 100 VOLTAGE NOISE AND CURRENT NOISE vs FREQUENCY 1 VO 750 750 1 k - 0.1 + 50 VI THS300x 1000 VO Zo = -1 VI ( 0.01 100k 1M VCC = 15 V and 5 V TA = 25C 100 In- 10 In+ ) 10M 100M f - Frequency - Hz Vn 1 1G 10 100 Figure 10 PSRR - Power Supply Rejection Ratio - dB PSRR - Power Supply Rejection Ratio - dB 90 VCC = 5 V VCC = 15 V VCC = 15 V 60 50 VCC = 5 V -PSRR 40 +PSRR 30 20 10 0 1k G=1 RF = 1 k 10k 100k 1M 10M 100M 85 VCC = -5 V VCC = -15 V 80 VCC = +5 V 75 VCC = +15 V 70 -40 -20 0 20 Figure 12 Figure 13 POST OFFICE BOX 655303 40 60 TA - Free-Air Temperature - C f - Frequency - Hz 8 100k POWER SUPPLY REJECTION RATIO vs FREE-AIR TEMPERATURE 80 70 10k Figure 11 POWER SUPPLY REJECTION RATIO vs FREQUENCY 90 1k f - Frequency - Hz * DALLAS, TEXAS 75265 80 100 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS SLEW RATE vs SUPPLY VOLTAGE SLEW RATE vs OUTPUT STEP 7000 +SR SR - Slew Rate - V/ s 6000 SR - Slew Rate - V/ s 10000 G = +5 RL = 150 tr/tf = 300 ps RF = 1 k 5000 4000 3000 +SR 2000 -SR 1000 VCC = 15 V G = +5 RL = 150 tr/tf = 300 ps RF = 1 k -SR 1000 5 7 9 11 13 100 15 0 5 |VCC| - Supply Voltage - V 10 15 20 VO(PP) - Output Step - V Figure 14 Figure 15 SLEW RATE vs OUTPUT STEP NORMALIZED SLEW RATE vs GAIN 2000 1.5 VCC = 5 V VO(PP) = 4 V RL = 150 RF = 1 k tr/tf = 300 ps +SR SR - Normalized Slew Rate - V/s 1.4 SR - Slew Rate - V/ s 1000 -SR VCC = 5 V G = +5 RL = 150 tr/tf = 300 ps RF= 1 k 1 2 3 4 1.2 -Gain 1.1 1 +Gain 0.9 0.8 100 0 1.3 5 0.7 1 2 VO(PP) - Output Step - V 3 4 5 6 7 8 9 10 G - Gain - V/V Figure 16 Figure 17 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 9 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS HARMONIC DISTORTION vs PEAK-TO-PEAK OUTPUT VOLTAGE SWING HARMONIC DISTORTION vs PEAK-TO-PEAK OUTPUT VOLTAGE SWING -50 -50 8 MHz Gain = 2 VCC = 15 V RL = 150 RF = 750 -60 Harmonic Distortion - dBc Harmonic Distortion - dBc -55 4 MHz Gain = 2 VCC = 15 V RL = 150 RF = 750 -55 3rd Harmonic -65 -70 2nd Harmonic -75 -60 3rd Harmonic -65 -70 -75 -80 2nd Harmonic -85 -80 -90 -85 0 2 4 6 8 10 12 14 16 18 -95 20 0 VO(PP) - Peak-to-Peak Output Voltage Swing - V 2 4 6 Figure 18 12 14 16 18 20 HARMONIC DISTORTION vs FREQUENCY -70 -60 Gain = 2 VCC = 15 V VO = 2 VPP RL = 150 RF = 750 -65 Harmonic Distortion - dBc Harmonic Distortion - dBc 10 Figure 19 HARMONIC DISTORTION vs FREQUENCY -75 8 VO(PP) - Peak-to-Peak Output Voltage Swing - V -80 3rd Harmonic -85 -90 2nd Harmonic -70 Gain = 2 VCC = 5 V VO = 2 VPP RL = 150 RF = 750 -75 -80 -85 2nd Harmonic -90 -95 -95 3rd Harmonic -100 100k 1M 10M -100 100k f - Frequency - Hz Figure 20 10 1M f - Frequency - Hz Figure 21 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 10M THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS DIFFERENTIAL GAIN vs LOADING DIFFERENTIAL GAIN vs LOADING 0.04 0.04 Gain = 2 RF = 750 40 IRE NTSC Modulation Worst Case: 100 IRE Ramp Gain = 2 RF = 750 40 IRE PAL Modulation Worst Case: 100 IRE Ramp 0.03 Differential Gain - % Differential Gain - % 0.03 VCC = 15 V 0.02 VCC = 5 V 0.01 VCC = 15 V 0.02 VCC = 5 V 0.01 0 1 2 3 4 5 6 7 0 8 1 2 Number of 150 Loads 3 Figure 22 5 6 7 8 Figure 23 DIFFERENTIAL PHASE vs LOADING DIFFERENTIAL PHASE vs LOADING 0.3 0.35 Gain = 2 RF = 750 40 IRE NTSC Modulation Worst Case: 100 IRE Ramp Gain = 2 RF = 750 40 IRE PAL Modulation Worst Case: 100 IRE Ramp 0.3 Differential Phase - Degrees 0.25 Differential Phase - Degrees 4 Number of 150 Loads 0.2 0.15 VCC = 15 V 0.1 VCC = 5 V 0.25 0.2 0.15 VCC = 15 V 0.1 VCC = 5 V 0.05 0.05 0 1 2 3 4 5 6 7 8 0 1 2 Number of 150 Loads 3 4 5 6 7 8 Number of 150 Loads Figure 24 Figure 25 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 11 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS OUTPUT AMPLITUDE vs FREQUENCY OUTPUT AMPLITUDE vs FREQUENCY 3 1 RF = 750 1 0 -1 -2 RF = 1 k -3 RF = 1.5 k -1 -2 RF = 1.5 k -5 -5 100M 10M RF = 1 k -3 -4 1M -6 100k 1G 1M f - Frequency - Hz OUTPUT AMPLITUDE vs FREQUENCY 9 8 RF = 560 7 Output Amplitude - dB Output Amplitude - dB 9 Gain = 2 VCC = 15 V RL = 150 VI = 200 mV RMS 6 5 4 RF = 680 3 RF = 1 k 2 4 RF = 1 k 2 0 100M 1G -1 100k f - Frequency - Hz 1M 10M f - Frequency - Hz Figure 28 12 RF = 750 3 0 10M RF = 560 5 1 1M Gain = 2 VCC = 5 V RL = 150 VI = 200 mV RMS 6 1 -1 100k 1G Figure 27 OUTPUT AMPLITUDE vs FREQUENCY 7 100M 10M f - Frequency - Hz Figure 26 8 RF = 750 0 -4 -6 100k Gain = 1 VCC = 5 V RL = 150 VI = 200 mV RMS 2 Output Amplitude - dB 2 Output Amplitude - dB 3 Gain = 1 VCC = 15 V RL = 150 VI = 200 mV RMS Figure 29 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 100M 1G THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS OUTPUT AMPLITUDE vs FREQUENCY 70 Output Amplitude - dB 60 50 VCC = 15 V 40 30 VCC = 5 V 20 10 0 G = +1000 RF = 10 k RL = 150 VO = 200 mV RMS -10 100k 10M 1M 1G 100M f - Frequency - Hz Figure 30 NORMALIZED OUTPUT RESPONSE vs FREQUENCY NORMALIZED OUTPUT RESPONSE vs FREQUENCY 3 1 2 RF = 560 Normalized Output Response - dB Normalized Output Response - dB 2 3 Gain = -1 VCC = 15 V RL = 150 VI = 200 mV RMS 0 -1 RF = 680 -2 -3 RF = 1 k -4 -5 -6 100k 1 Gain = -1 VCC = 5 V RL = 150 VI = 200 mV RMS RF = 560 0 -1 RF = 750 -2 -3 RF = 1 k -4 -5 1M 10M 100M 1G -6 100k f - Frequency - Hz 1M 10M 100M 1G f - Frequency - Hz Figure 32 Figure 31 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 13 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS NORMALIZED OUTPUT RESPONSE vs FREQUENCY NORMALIZED OUTPUT RESPONSE vs FREQUENCY 3 4 RF = 390 RF = 390 0 Normalized Output Response - dB Normalized Output Response - dB 2 RF = 560 -3 RF = 1 k -6 -9 Gain = +5 VCC = 15 V RL = 150 VO = 200 mV RMS -12 -15 100k 1M 0 -2 RF = 620 -4 -6 RF = 1 k -8 -10 Gain = +5 VCC = 5 V RL = 150 VO = 200 mV RMS -12 100M 10M -14 100k 1G 1M 10M f - Frequency - Hz Figure 33 SMALL AND LARGE SIGNAL FREQUENCY RESPONSE 3 -3 VI = 500 mV VI = 500 mV -6 0 -3 -9 VI = 250 mV Output Level - dBV Output Level - dBV 1G Figure 34 SMALL AND LARGE SIGNAL FREQUENCY RESPONSE -12 -15 VI = 125 mV -18 -21 VI = 250 mV -6 -9 VI = 125 mV -12 -15 VI = 62.5 mV -24 -27 -30 100k 14 100M f - Frequency - Hz VI = 62.5 mV -18 Gain = 1 VCC = 15 V RF = 1 k RL = 150 1M -21 100M 10M 1G -24 100k Gain = 2 VCC = 15 V RF = 680 RL = 150 1M 10M f - Frequency - Hz f - Frequency - Hz Figure 35 Figure 36 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 100M 1G THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS VI - Input Voltage - mV SMALL SIGNAL PULSE RESPONSE 100 -100 -200 200 VO- Output Voltage - mV VO - Output Voltage - V VI - Input Voltage - mV SMALL SIGNAL PULSE RESPONSE 300 100 0 Gain = 1 VCC = 5 V RL = 150 RF = 1 k tr/tf = 300 ps -100 -200 -300 0 10 20 30 40 50 60 70 80 60 20 -20 -60 200 100 0 Gain = 5 VCC = 5 V RL = 150 RF = 1 k tr/tf = 300 ps -100 -200 -300 90 100 0 10 20 30 t - Time - ns Figure 37 VI - Input Voltage - V 1 -1 -3 2 1 Gain = +1 VCC = 15 V RL = 150 RF = 1 k tr/tf= 2.5 ns -2 -3 60 70 80 90 100 LARGE SIGNAL PULSE RESPONSE VO - Output Voltage - V VO - Output Voltage - V VI - Input Voltage - V LARGE SIGNAL PULSE RESPONSE -1 50 Figure 38 3 0 40 t - Time - ns 3 1 -1 -3 2 1 0 Gain = 1 VCC = 5 V RL = 150 RF = 1 k tr/tf= 2.5 ns -1 -2 -3 0 10 20 30 40 50 60 70 80 90 100 0 10 t - Time - ns 20 30 40 50 60 70 80 90 100 t - Time - ns Figure 39 Figure 40 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 15 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS VI - Input Voltage - mV LARGE SIGNAL PULSE RESPONSE 1 -1 -3 10 VO - Output Voltage - V VO - Output Voltage - V VI - Input Voltage - V LARGE SIGNAL PULSE RESPONSE 3 5 0 Gain = +5 VCC = 15 V RL = 150 RF = 1 k tr/tf= 300 ps -5 -10 -15 600 200 -200 -600 2 1 0 Gain = 5 VCC = 5 V RL = 150 RF = 1 k tr/tf= 300 ps -1 -2 -3 0 10 20 30 40 50 60 70 80 90 100 0 10 20 30 t - Time - ns Figure 41 VI - Input Voltage - V 1 -1 2 1 Gain = -1 VCC = 15 V RL = 150 RF = 1 k tr/tf= 2.5 ns -2 -3 70 80 90 100 3 1 -1 2 1 Gain = -1 VCC = 5 V RL = 150 RF = 1 k tr/tf= 300 ps 0 -1 -2 -3 0 10 20 30 40 50 60 70 80 90 100 0 10 t - Time - ns 20 30 40 50 60 t - Time - ns Figure 43 16 60 LARGE SIGNAL PULSE RESPONSE VO - Output Voltage - V VO - Output Voltage - V VI - Input Voltage - V LARGE SIGNAL PULSE RESPONSE -1 50 Figure 42 3 0 40 t - Time - ns Figure 44 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 70 80 90 100 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 TYPICAL CHARACTERISTICS LARGE SIGNAL PULSE RESPONSE VI - Input Voltage - V 200 -200 -600 2 VO - Output Voltage - V VO - Output Voltage - V VI - Input Voltage - mV LARGE SIGNAL PULSE RESPONSE 600 1 0 Gain = -5 VCC = 5 V RL = 150 RF = 1 k tr/tf= 300 ps -1 -2 -3 0 10 20 30 40 50 60 70 80 90 100 3 1 -1 -2 10 5 0 Gain = -5 VCC = 15 V RL = 150 RF = 1 k tr/tf= 300 ps -5 -10 -15 0 10 t - Time - ns 20 30 40 50 60 70 80 90 100 t - Time - ns Figure 45 Figure 46 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 17 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION theory of operation The THS300x is a high-speed, operational amplifier configured in a voltage-feedback architecture. The device is built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing fTs of several GHz. This configuration implements an exceptionally high-performance amplifier that has a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic is shown in Figure 47. VCC+ 7 IIB IN+ 3 2 IN- 6 IIB 4 VCC- Figure 47. Simplified Schematic 18 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 OUT THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION recommended feedback and gain resistor values The THS300x is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This process provides the excellent isolation and extremely high slew rates that result in superior distortion characteristics. As with all current-feedback amplifiers, the bandwidth of the THS300x is an inversely proportional function of the value of the feedback resistor (see Figures 26 to 34). The recommended resistors for the optimum frequency response are shown in Table 1. These should be used as a starting point and once optimum values are found, 1% tolerance resistors should be used to maintain frequency response characteristics. For most applications, a feedback resistor value of 1 k is recommended - a good compromise between bandwidth and phase margin that yields a very stable amplifier. Consistent with current-feedback amplifiers, increasing the gain is best accomplished by changing the gain resistor, not the feedback resistor. This is because the bandwidth of the amplifier is dominated by the feedback resistor value and internal dominant-pole capacitor. The ability to control the amplifier gain independent of the bandwidth constitutes a major advantage of current-feedback amplifiers over conventional voltage-feedback amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value of the gain resistor to increase or decrease the overall amplifier gain. Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance decreases the loop gain and increases the distortion. It is also important to know that decreasing load impedance increases total harmonic distortion (THD). Typically, the third-order harmonic distortion increases more than the second-order harmonic distortion. Table 1. Recommended Resistor Values for Optimum Frequency Response GAIN RF for VCC = 15 V RF for VCC = 5 V 1 1 k 1 k 2, -1 680 750 -2 620 620 5 560 620 offset voltage The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times the corresponding gains. The following schematic and formula can be used to calculate the output offset voltage: RF IIB- RG + - VIO VO + RS IIB+ V OO + VIO 1 ) R R F G " IIB) RS 1 ) R R F G " IIB- RF Figure 48. Output Offset Voltage Model POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 19 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION noise calculations and noise figure Noise can cause errors on very small signals. This is especially true for amplifying small signals coming over a transmission line or an antenna. The noise model for current-feedback amplifiers (CFB) is the same as for voltage feedback amplifiers (VFB). The only difference between the two is that CFB amplifiers generally specify different current-noise parameters for each input, while VFB amplifiers usually only specify one noise-current parameter. The noise model is shown in Figure 49. This model includes all of the noise sources as follows: * * * * en = amplifier internal voltage noise (nV/Hz) IN+ = noninverting current noise (pA/Hz) IN- = inverting current noise (pA/Hz) eRx = thermal voltage noise associated with each resistor (eRx = 4 kTRx ) eRs RS en Noiseless + _ eni IN+ eno eRf RF eRg IN- RG Figure 49. Noise Model The total equivalent input noise density (eni) is calculated by using the following equation: e + ni Where: en 2 ) IN ) ) 2 R S IN- R o RG ) 4 kTRs ) 4 kT RF o RG F 2 k = Boltzmann's constant = 1.380658 x 10-23 T = temperature in degrees Kelvin (273 +C) RF || RG = parallel resistance of RF and RG To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the overall amplifier gain (AV). e no 20 + eni AV + e ni 1 ) RR F (Noninverting Case) G POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION noise calculations and noise figure (continued) As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain is increased (by reducing RG), the input noise is reduced considerably because of the parallel resistance term. This leads to the general conclusion that the most dominant noise sources are the source resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier. This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be defined and is typically 50 in RF applications. NF + 10log e 2 ni e 2 Rs Because the dominant noise components are generally the source resistance and the internal amplifier noise voltage, we can approximate noise figure as: ) e NF + 10log 1 ) ) 2 n IN 4 kTR 2 R S S The Figure 50 shows the noise figure graph for the THS300x. NOISE FIGURE vs SOURCE RESISTANCE 20 18 f = 10 kHz TA = 25C 16 Noise Figure - dB 14 12 10 8 6 4 2 0 10 100 1k 10k RS - Source Resistance - Figure 50. Noise Figure vs Source Resistance POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 21 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION slew rate The slew rate performance of a current-feedback amplifier, like the THS300x, is affected by many different factors. Some of these factors are external to the device, such as amplifier configuration and PCB parasitics, and others are internal to the device, such as available currents and node capacitance. Understanding some of these factors should help the PCB designer arrive at a more optimum circuit with fewer problems. Whether the THS300x is used in an inverting amplifier configuration or a noninverting configuration can impact the output slew rate. As can be seen from the specification tables as well as some of the figures in this data sheet, slew-rate performance in the inverting configuration is faster than in the noninverting configuration. This is because in the inverting configuration the input terminals of the amplifier are at a virtual ground and do not significantly change voltage as the input changes. Consequently, the time to charge any capacitance on these input nodes is less than for the noninverting configuration, where the input nodes actually do change in voltage an amount equal to the size of the input step. In addition, any PCB parasitic capacitance on the input nodes degrades the slew rate further simply because there is more capacitance to charge. Also, if the supply voltage (VCC ) to the amplifier is reduced, slew rate decreases because there is less current available within the amplifier to charge the capacitance on the input nodes as well as other internal nodes. Internally, the THS300x has other factors that impact the slew rate. The amplifier's behavior during the slew-rate transition varies slightly depending upon the rise time of the input. This is because of the way the input stage handles faster and faster input edges. Slew rates (as measured at the amplifier output) of less than about 1500 V/s are processed by the input stage in a very linear fashion. Consequently, the output waveform smoothly transitions between initial and final voltage levels. This is shown in Figure 51. For slew rates greater than 1500 V/s, additional slew-enhancing transistors present in the input stage begin to turn on to support these faster signals. The result is an amplifier with extremely fast slew-rate capabilities. Figures 41 and 52 show waveforms for these faster slew rates. The additional aberrations present in the output waveform with these faster-slewing input signals are due to the brief saturation of the internal current mirrors. This phenomenon, which typically lasts less than 20 ns, is considered normal operation and is not detrimental to the device in any way. If for any reason this type of response is not desired, then increasing the feedback resistor or slowing down the input-signal slew rate reduces the effect. SLEW RATE 4 VI - Input Voltage - V VI - Input Voltage - V SLEW RATE 2 0 10 2 0 -2 5 VO - Output Voltage - V VO - Output Voltage - V 5 4 SR = 1500 V/s Gain = 5 VCC = 15 V RL = 150 RF = 1 k tr/tf = 10 ns 0 -5 -10 -15 0 20 40 60 80 100 120 140 160 180 200 SR = 2400 V/s Gain = 5 VCC = 15 V RL = 150 RF = 1 k tr/tf = 5 ns 0 -5 -10 -15 0 20 t - Time - ns 60 80 100 120 140 160 180 200 t - Time - ns Figure 51 22 40 Figure 52 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION driving a capacitive load Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are taken. The first is to realize that the THS300x has been internally compensated to maximize its bandwidth and slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the output will decrease the device's phase margin leading to high-frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of the amplifier, as shown in Figure 53. A minimum value of 20 should work well for most applications. For example, in 75- transmission systems, setting the series resistor value to 75 both isolates any capacitance loading and provides the proper line impedance matching at the source end. 1 k 1 k Input _ 20 Output THS300x + CLOAD Figure 53. Driving a Capacitive Load PCB design considerations Proper PCB design techniques in two areas are important to assure proper operation of the THS300x. These areas are high-speed layout techniques and thermal-management techniques. Because the THS300x is a high-speed part, the following guidelines are recommended. D D Ground plane - It is essential that a ground plane be used on the board to provide all components with a low inductive ground connection. Although a ground connection directly to a terminal of the THS300x is not necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves two functions: it provides a low inductive ground to the device substrate to minimize internal crosstalk, and it provides the path for heat removal. Input stray capacitance - To minimize potential problems with amplifier oscillation, the capacitance at the inverting input of the amplifiers must be kept to a minimum. To do this, PCB trace runs to the inverting input must be as short as possible, the ground plane must be removed under any etch runs connected to the inverting input, and external components should be placed as close as possible to the inverting input. This is especially true in the noninverting configuration. An example of this can be seen in Figure 54, which shows what happens when a 1-pF capacitor is added to the inverting input terminal. The bandwidth increases at the expense of peaking. This is because some of the error current is flowing through the stray capacitor instead of the inverting node of the amplifier. Although, while the device is in the inverting mode, stray capacitance at the inverting input has a minimal effect. This is because the inverting node is at a virtual ground and the voltage does not fluctuate nearly as much as in the noninverting configuration. This can be seen in Figure 55, where a 10-pF capacitor adds only 0.35 dB of peaking. In general, as the gain of the system increases, the output peaking due to this capacitor decreases. While this can initially look like a faster and better system, overshoot and ringing are more likely to occur under fast transient conditions. So proper analysis of adding a capacitor to the inverting input node should be performed for stable operation. POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 23 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION PCB design considerations (continued) OUTPUT AMPLITUDE vs FREQUENCY OUTPUT AMPLITUDE vs FREQUENCY 7 1 1 k Cin Output Amplitude - dB 5 Vin 0 Vout - RL = 150 50 3 2 1 0 -1 -2 -3 10M 1M 100M Cin -2 Vin -3 Vout - 50 -4 + -5 -7 RL = 150 1G Gain = -1 VCC = 15 V VO = 200 mV RMS -8 100k f - Frequency - Hz 1M 10M 100M 1G f - Frequency - Hz Figure 54 D 1 k 1 k -6 CI = 0 pF (Stray C Only) Gain = 1 VCC = 15 V VO = 200 mV RMS -4 100k CI = Stray C Only -1 + 4 CI = 10 pF CI = 1 pF Output Amplitude - dB 6 Figure 55 Proper power-supply decoupling - Use a minimum 6.8-F tantalum capacitor in parallel with a 0.1-F ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers depending on the application, but a 0.1-F ceramic capacitor should always be used on the supply terminal of every amplifier. In addition, the 0.1-F capacitor should be placed as close as possible to the supply terminal. As this distance increases, the inductance in the connecting etch makes the capacitor less effective. The designer should strive for distances of less than 0.1 inches between the device power terminal and the ceramic capacitors. thermal information The THS300x incorporates output-current-limiting protection. Should the output become shorted to ground, the output current is automatically limited to the value given in the data sheet. While this protects the output against excessive current, the device internal power dissipation increases due to the high current and large voltage drop across the output transistors. Continuous output shorts are not recommended and could damage the device. Additionally, connection of the amplifier output to one of the supply rails (VCC) is not recommended. Failure of the device is possible under this condition and should be avoided. But, the THS300x does not incorporate thermal-shutdown protection. Because of this, special attention must be paid to the device's power dissipation or failure may result. 24 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION thermal information (continued) The thermal coefficient JA is approximately 169C/W for the SOIC 8-pin D package. For a given JA, the maximum power dissipation, shown in Figure 56, is calculated by the following formula: P + D T -T MAX A q JA Where: PD = Maximum power dissipation of THS300x (watts) TMAX = Absolute maximum junction temperature (150C) TA = Free-ambient air temperature (C) JA = Thermal coefficient from die junction to ambient air (C/W) MAXIMUM POWER DISSIPATION vs FREE-AIR TEMPERATURE PD - Maximum Power Dissipation - W 1.5 SOIC-D Package: JA = 169C/W TJ = 150C No Airflow 1 0.5 0 -40 -20 0 20 40 60 80 100 TA - Free-Air Temperature - C Figure 56. Maximum Power Dissipation vs Free-Air Temperature POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 25 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION general configurations A common error for the first-time CFB user is the creation of a unity gain buffer amplifier by shorting the output directly to the inverting input. A CFB amplifier in this configuration will oscillate and is not recommended. The THS300x, like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing capacitors directly from the output to the inverting input is not recommended. This is because, at high frequencies, a capacitor has a very low impedance. This results in an unstable amplifier and should not be considered when using a current-feedback amplifier. Because of this, integrators and simple low-pass filters, which are easily implemented on a VFB amplifier, have to be designed slightly differently. If filtering is required, simply place an RC-filter at the noninverting terminal of the operational-amplifier (see Figure 57). RG RF f O V I VO + VI R1 + V - 1 + 2pR1C1 -3dB 1 ) RRF G 1 ) sR1C1 1 C1 Figure 57. Single-Pole Low-Pass Filter If a multiple-pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is because the filtering elements are not in the negative feedback loop and stability is not compromised. Because of their high slew-rates and high bandwidths, CFB amplifiers can create very accurate signals and help minimize distortion. An example is shown in Figure 58. C1 + _ VI R1 R1 = R2 = R C1 = C2 = C Q = Peaking Factor (Butterworth Q = 0.707) R2 f C2 RG RF -3dB RG = + 2p1RC ( RF 1 2- Q ) Figure 58. 2-Pole Low-Pass Sallen-Key Filter There are two simple ways to create an integrator with a CFB amplifier. The first, shown in Figure 59, adds a resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant and the feedback impedance never drops below the resistor value. The second, shown in Figure 60, uses positive feedback to create the integration. Caution is advised because oscillations can occur due to the positive feedback. 26 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION general configurations (continued) C1 RF V RG - VI VO + O V I + THS300x R R ) S F G 1 R C1 F S Figure 59. Inverting CFB Integrator RG RF For Stable Operation: R2 R1 || RA - THS300x VO + VO VI R1 R2 ( RF RG RF RG sR1C1 1+ ) VI RA C1 Figure 60. Noninverting CFB Integrator The THS300x may also be employed as a very good video distribution amplifier. One characteristic of distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are compromised as the number of lines increases and the closed-loop gain increases (see Figures 22 to 25 for more information). Be sure to use termination resistors throughout the distribution system to minimize reflections and capacitive loading. 750 750 75 - 75- Transmission Line VO1 + VI 75 75 THS300x N Lines 75 VON 75 Figure 61. Video Distribution Amplifier Application POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 27 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION evaluation board Evaluation boards are available for the THS3001 (literature #SLOP130) and the THS3002 (literature #SLOP241). The boards have been configured for very low parasitic capacitance in order to realize the full performance of the amplifier. Schematics of the evaluation boards are shown in Figures 62 and 63. The circuitry has been designed so that the amplifier may be used in either an inverting or noninverting configuration. To order the evaluation board contact your local TI sales office or distributor. For more detailed information, refer to the THS3001 EVM User's Manual (literature #SLOV021) or the THS3002 EVM User's Guide (literature #SLOVxxx). To order the evaluation board, contact your local TI sales office or distributor. VCC+ + C2 0.1 F C1 6.8 F R1 1 k IN + R2 49.9 + R3 49.9 OUT THS3001 _ R5 1 k + C4 0.1 F C3 6.8 F IN - R4 49.9 VCC - Figure 62. THS3001 Evaluation Board Schematic 28 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 APPLICATION INFORMATION evaluation board (continued) VCC+ C5 0.1 F C3 R6 301 R5 R4 100 R1 100 R3 100 2 3 8 - THS3002 U1:A R7 49.9 OUT1 1 C4 0.1 F + 4 R2 0 VCC- C6 R8 R14 301 R13 R9 100 R12 100 R11 100 6 - THS3002 U1:B 7 5 R15 49.9 OUT2 + R10 0 Figure 63. THS3002 Evaluation Board Schematic POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 29 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 MECHANICAL INFORMATION D (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE 14 PIN SHOWN PINS ** 0.050 (1,27) 8 14 16 A MAX 0.197 (5,00) 0.344 (8,75) 0.394 (10,00) A MIN 0.189 (4,80) 0.337 (8,55) 0.386 (9,80) DIM 0.020 (0,51) 0.014 (0,35) 14 0.010 (0,25) M 8 0.244 (6,20) 0.228 (5,80) 0.008 (0,20) NOM 0.157 (4,00) 0.150 (3,81) 1 Gage Plane 7 A 0.010 (0,25) 0- 8 0.044 (1,12) 0.016 (0,40) Seating Plane 0.069 (1,75) MAX 0.010 (0,25) 0.004 (0,10) 0.004 (0,10) 4040047 / D 10/96 NOTES: A. B. C. D. 30 All linear dimensions are in inches (millimeters). This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15). Falls within JEDEC MS-012 POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS SLOS217A - JULY 1998 - REVISED JUNE 1999 MECHANICAL INFORMATION DGN (S-PDSO-G8) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE 0,38 0,25 0,65 8 0,25 M 5 Thermal Pad (See Note D) 0,15 NOM 3,05 2,95 4,98 4,78 Gage Plane 0,25 1 0- 6 4 3,05 2,95 0,69 0,41 Seating Plane 1,07 MAX 0,15 0,05 0,10 4073271/A 01/98 NOTES: A. B. C. D. All linear dimensions are in millimeters. This drawing is subject to change without notice. Body dimensions include mold flash or protrusions. The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad. This pad is electrically and thermally connected to the backside of the die and possibly selected leads. E. Falls within JEDEC MO-187 PowerPAD is a trademark of Texas Instruments Incorporated. POST OFFICE BOX 655303 * DALLAS, TEXAS 75265 31 IMPORTANT NOTICE Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with TI's standard warranty. Testing and other quality control techniques are utilized to the extent TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed, except those mandated by government requirements. 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