THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
D
High Speed
– 420 MHz Bandwidth (G = 1, –3 dB)
– 6500 V/µs Slew Rate
– 40-ns Settling Time (0.1%)
D
High Output Drive, IO = 100 mA
D
Excellent Video Performance
– 115 MHz Bandwidth (0.1 dB, G = 2)
– 0.01% Differential Gain
– 0.02° Differential Phase
D
Low 3-mV (max) Input Offset Voltage
D
Very Low Distortion
– THD = –96 dBc at f = 1 MHz
– THD = –80 dBc at f = 10 MHz
D
Wide Range of Power Supplies
– VCC = ±4.5 V to ±16 V
D
Evaluation Module Available
description
The THS300x is a high-speed current-feedback
operational amplifier, ideal for communication,
imaging, and high-quality video applications. This
device offers a very fast 6500-V/µs slew rate, a
420-MHz bandwidth, and 40-ns settling time for
large-signal applications requiring excellent tran-
sient response. In addition, the THS300x
operates with a very low distortion of – 96 dBc,
making it well suited for applications such as
wireless communication basestations or ultrafast
ADC or DAC buffers.
HIGH-SPEED AMPLIFIER FAMILY
DEVICE ARCHITECTURE SUPPLY
VOLTAGE BW
(MHz)
SR
(V/µs)
THD
f = 1 MHz ts
0.1% DIFF.
GAIN
DIFF.
PHASE
Vn
(nV/√Hz)
VFB CFB 5 V ±5 V ±15 V
(MHz)
(V/µs)
(dB) (ns)
GAIN
PHASE
(
n
V/√H
z
)
THS3001/02 • • • 420 6500 –96 40 0.01% 0.02°1.6
THS4001 • • • • 270 400 –72 40 0.04% 0.15°12.5
THS4011/12 • • • 290 310 –80 37 0.006% 0.01°7.5
THS4031/32 • • • 100 100 –72 60 0.02% 0.03°1.6
THS4061/62 • • • 180 400 –72 40 0.02% 0.02°14.5
CAUTION: The THS300x provides ESD protection circuitry. However , permanent damage can still occur if this device is subjected
to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss
of functionality.
Copyright 1999, Texas Instruments Incorporated
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
5
3
1
–1 1M 100M
6
4
2
0
10M 1G100k
7
8
Output Amplitude – dB
G = 2
RL = 150 Ω
VI = 200 mV RMS
VCC = ±5 V
RF = 750 Ω
VCC = ±15 V
RF = 680 Ω
THS3002
D AND DGN PACKAGE
(TOP VIEW)
1
2
3
4
8
7
6
5
1OUT
1IN–
1IN+
–VCC
VCC+
2OUT
2IN–
2IN+
1
2
3
4
8
7
6
5
NULL
IN–
IN+
VCC–
NULL
VCC+
OUT
NC
THS3001
D AND DGN† PACKAGE
(TOP VIEW)
NC – No internal connection
†The THS3001 implemented in the DGN package is in the
product preview stage of development. Contact your local TI
sales office for availability.
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
2POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
AVAILABLE OPTIONS
PACKAGED DEVICE
EVALUATION
TASOIC
†
MSOP (DGN) EVALUATION
MODULE
(D) DEVICE SYMBOL
MODULE
0°C to 70°CTHS3001CD
THS3002CD‡THS3001CDGN‡
THS3002CDGN‡TIADP
TIADI THS3001EVM
THS3002EVM‡
–40°C to 85°CTHS3001ID
THS3002ID‡THS3001IDGN‡
THS3002IDGN‡TIADQ
TIADJ —
†The D package is available taped and reeled. Add an R suffix to the device type (i.e.,
THS3001CDR)
‡Product Preview
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage, VCC+ to VCC– 33 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input voltage, VI ±VCC
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output Current, IO 175 mA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Differential input voltage, VID ±6 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Continuous total power dissipation See Dissipation Rating Table. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Operating free-air temperature, TA, THS300xC 0°C to 70°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
THS300xI –40°C to 85°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Storage temperature, Tstg –65°C to 125°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Lead temperature, 1,6 mm (1/16 inch) from case for 10 seconds 300°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
†Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only , and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may af fect device reliability.
DISSIPATION RATING TABLE
PACKAGE
T
A
≤ 25°CDERATING FACTOR T
A
= 70°C T
A
= 85°C
PACKAGE
A
POWER RATING ABOVE TA = 25°C
A
POWER RATING
A
POWER RATING
D740 mW 6 mW/°C470 mW 380 mW
recommended operating conditions
MIN NOM MAX UNIT
Su
pp
ly voltage VCC and VCC
Split supply ±4.5 ±16
V
S
u
ppl
y v
oltage
,
V
CC+
and
V
CC– Single supply 9 32
V
O
p
erating free air tem
p
erature TA
THS300xC 0 70 °
C
Operating
free
-
air
temperat
u
re
,
T
ATHS300xI –40 85
°C
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
electrical characteristics, TA = 25°C, RL = 150 Ω, RF = 1 kΩ (unless otherwise noted)
PARAMETER TEST CONDITIONS†MIN TYP MAX UNIT
VCC
Power su
pp
ly o
p
erating range
Split supply ±4.5 ±16.5
V
V
CC
Po
w
er
s
u
ppl
y
operating
range
Single supply 9 33
V
VCC =±5V
TA = 25°C 5.5 7.5
ICC
Quiescent current
V
CC =
±5
V
TA = full range 8.5
mA
I
CC
Q
u
iescent
c
u
rrent
VCC =±15 V
TA = 25°C 6.6 9
mA
V
CC =
±15
V
TA = full range 10
VCC =±5V
RL = 150 Ω±2.9 ±3.2
VO
Out
p
ut voltage swing
V
CC =
±5
V
RL = 1 kΩ±3±3.3
V
V
O
O
u
tp
u
t
v
oltage
s
w
ing
VCC =±15 V
RL = 150 Ω±12.1 ±12.8
V
V
CC =
±15
V
RL = 1 kΩ±12.8 ±13.1
IO
Out
p
ut current (see Note 1)
VCC = ±5 V, RL = 20 Ω100
mA
I
O
O
u
tp
u
t
c
u
rrent
(see
Note
1)
VCC = ±15 V, RL = 75 Ω85 120
mA
VIO
In
p
ut offset voltage
VCC =±5Vor±15 V
TA = 25°C 1 3
mV
V
IO
Inp
u
t
offset
v
oltage
V
CC =
±5
V
or
±15
V
TA = full range 4
mV
Input offset voltage drift VCC = ±5 V or ±15 V 5 µV/°C
–In
p
ut
TA = 25°C 2 10
IIB
In
p
ut bias current
–
In ut
VCC =±5Vor±15 V
TA = full range 15
µA
IIB
In ut
bias
current
+In
p
ut
VCC
=
±5
V
or
±15
V
TA = 25°C 1 10
µA
+
Inp
u
t
TA = full range 15
VICR
Common mode in
p
ut voltage range
VCC = ±5 V ±3±3.2
V
V
ICR
Common
-
mode
inp
u
t
v
oltage
range
VCC = ±15 V ±12.9 ±13.2
V
O
p
en loo
p
transresistance
VCC = ±5 V,
RL = 1 kΩVO = ±2.5 V, 1.3
MΩ
Oen
loo
transresistance
VCC = ±15 V,
RL = 1 kΩVO = ±7.5 V, 2.4
MΩ
CMRR
Common-mode rejection ratio
VCC = ±5 V, VCM = ±2.5 V 62 70
dB
CMRR
Common
-
mode
rejection
ratio
VCC = ±15 V, VCM = ±10 V 65 73
dB
VCC =±5V
TA = 25°C 65 76
dB
PSRR
Power su
pp
ly rejection ratio
V
CC =
±5
V
TA = full range 63
dB
PSRR
Po
w
er
s
u
ppl
y
rejection
ratio
VCC =±15 V
TA = 25°C 69 76
dB
V
CC =
±15
V
TA = full range 67
dB
RI
In
p
ut resistance
+Input 1.5 MΩ
R
I
Inp
u
t
resistance
–Input 15 Ω
CIDifferential input capacitance 7.5 pF
ROOutput resistance Open loop at 5 MHz 10 Ω
VnInput voltage noise VCC = ±5 V or ±15 V, f = 10 kHz,
G = 2 1.6 nV/√Hz
I
In
p
ut current noise
Positive (IN+) V
CC
= ±5 V or ±15 V, f = 10 kHz, 13
pA/√Hz
I
n
Inp
u
t
c
u
rrent
noise
Negative (IN–)
CC ,
G = 2
,
16 p
A/√H
z
†Full range = 0°C to 70°C for the THS300xC and –40°C to 85°C for the THS300xI.
NOTE 1: Observe power dissipation ratings to keep the junction temperature below absolute maximum when the output is heavily loaded or
shorted. See absolute maximum ratings section.
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
4POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
operating characteristics, TA = 25°C, RL = 150 Ω, RF = 1 kΩ (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
V
CC
= ±5 V, G = –5 1700
SR
Slew rate (see Note 2)
CC ,
VO(PP) = 4 V G = 5 1300
V/µs
SR
Sle
w
rate
(see
Note
2)
V
CC
= ±15 V, G = –5 6500
V/
µ
s
CC ,
VO(PP) = 20 V G = 5 6300
t
Settling time to 0.1% VCC = ±15 V,
0 V to 10 V Step Gain = –1, 40
ns
t
sSettling time to 0.1% VCC = ±5 V,
0 V to 2 V Step, Gain = –1, 25
ns
THD Total harmonic distortion VCC = ±15 V,
fc = 10 MHz, VO(PP) = 2 V,
G = 2 –80 dBc
AD
Differential gain error
G = 2,
40 IRE modulation, VCC = ±5 V 0.015%
A
D
Differential
gain
error
,
±100 IRE Ramp,
NTSC and PAL VCC = ±15 V 0.01%
θD
Differential
p
hase error
G = 2,
40 IRE modulation, VCC = ±5 V 0.01°
θ
D
Differential
phase
error
,
±100 IRE Ramp,
NTSC and PAL VCC = ±15 V 0.02°
G=1 R
F=1kΩ
VCC = ±5 V, 330 MHz
G
=
1
,
R
F =
1
kΩ
,VCC = ±15 V, 420 MHz
Small signal bandwidth (–3 dB) G = 2, RF = 750 Ω, VCC = ±5 V 300
BW G = 2, RF = 680 Ω, VCC = ±15 V 385 MHz
G = 5, RF = 560 Ω, VCC = ±15 V 350
Bandwidth for 0 1 dB flatness
G = 2, RF = 750 Ω, VCC = ±5 V 85
MHz
Band
w
idth
for
0
.
1
dB
flatness
G = 2, RF = 680 Ω, VCC = ±15 V 115
MH
z
VCC = ±5 V,
VO(PP) =4V
G = –5 65 MHz
Full
p
ower bandwidth (see Note 3)
VO(PP)
=
4
V
,
RL = 500 ΩG = 5 62 MHz
Full
ower
bandwidth
(see
Note
3)
VCC = ±15 V,
VO(PP) =20V
G = –5 32 MHz
VO(PP)
=
20
V
RL = 500 ΩG = 5 31 MHz
Crosstalk (THS3002 only) TBD dB
NOTES: 2. Slew rate is measured from an output level range of 25% to 75%.
3. Full power bandwidth is defined as the frequency at which the output has 3% THD.
PARAMETER MEASUREMENT INFORMATION
VIVO
+
–
RGRF
RL
50 ΩVCC–
VCC+
Figure 1. Test Circuit, Gain = 1 + (RF/RG)
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
|VO|Output voltage swing vs Free-air temperature 2
ICC Current supply vs Free-air temperature 3
IIB Input bias current vs Free-air temperature 4
VIO Input offset voltage vs Free-air temperature 5
vs Common-mode input voltage 6
CMRR Common-mode rejection ratio vs Common-mode input voltage 7
vs Frequency 8
Transresistance vs Free-air temperature 9
Closed-loop output impedance vs Frequency 10
VnVoltage noise vs Frequency 11
InCurrent noise vs Frequency 11
PSRR
Power su
pp
ly rejection ratio
vs Frequency 12
PSRR
Po
w
er
s
u
ppl
y
rejection
ratio
vs Free-air temperature 13
Slew rate
vs Supply voltage 14
SR
Sle
w
rate
vs Output step peak-to-peak 15, 16
Normalized slew rate vs Gain 17
Harmonic distortion
vs Peak-to-peak output voltage swing 18, 19
Harmonic
distortion
vs Frequency 20, 21
Differential gain vs Loading 22, 23
Differential phase vs Loading 24, 25
Output amplitude vs Frequency 26–30
Normalized output response vs Frequency 31–34
Small and large signal frequency response 35, 36
Small signal pulse response 37, 38
Large signal pulse response 39 – 46
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
6POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 2
TA – Free-Air Temperature – °C
OUTPUT VOLTAGE SWING
vs
FREE-AIR TEMPERATURE
12
2–20 20
14
4
0 40 100–40 60 80
O – Output Voltage Swing – VV
12.5
13
13.5
3.5
3
2.5
VCC = ±5 V
RL = 150 Ω
VCC = ±5 V
No Load
VCC = ±15 V
RL = 150 Ω
VCC = ±15 V
No Load
Figure 3
TA – Free-Air Temperature – °C
CURRENT SUPPLY
vs
FREE-AIR TEMPERATURE
9
7
5
3–20 20
8
6
4
0 40 100–40 60 80
VCC = ±5 V
VCC = ±15 V
VCC = ±10 V
ICC – Supply Current – mA
Figure 4
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
–40 –20 0 20 80 100
TA – Free-Air Temperature – °C
6040
IIB – Input Bias Current –
–1
–2
–3
–0.5
–1.5
–2.5
Aµ
VCC = ±5 V
VCC = ±15 V
VCC = ±5 V
VCC = ±15 V
IIB–
IIB–
IIB+
IIB+
Figure 5
TA – Free-Air Temperature – °C
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
0
–0.4
–0.8
–1.2 –20 20
–0.2
–0.6
–1
0 40 100–40 60 80
VCC = ±5 V
VCC = ±15 V
Gain = 1
RF = 1 kΩ
VIO – Input Offset Voltage – mV
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 6
|VIC| – Common-Mode Input Voltage – V
COMMON-MODE REJECTION RATIO
vs
COMMON-MODE INPUT VOLTAGE
60
50
40
30 2648 14
80
010 12
70
TA = –40°C
CMRR – Common-Mode Rejection Ratio – dB
TA = 85°CTA = 25°C
VCC = ±15 V
Figure 7
|VIC| – Common-Mode Input Voltage – V
COMMON-MODE REJECTION RATIO
vs
COMMON-MODE INPUT VOLTAGE
50
40
30
20 0.5 1.512 4
80
02.5 3
60
TA = –40°C
CMRR – Common-Mode Rejection Ratio – dB
TA = 85°C
TA = 25°C
VCC = ±5 V
3.5
70
Figure 8
f – Frequency – Hz
COMMON-MODE REJECTION RATIO
vs
FREQUENCY
1k 10k 10M 100M1M100k
60
40
20
0
50
30
10
80
70 VCC = ±5 V
VCC = ±15 V
CMRR – Common-Mode Rejection Ratio – dB
1 kΩ
1 kΩ
VI+
–VO
1 kΩ
1 kΩ
Figure 9
TA – Free-Air Temperature – °C
TRANSRESISTANCE
vs
FREE-AIR TEMPERATURE
2.2
1.8
1.4
1–20 20
2.4
2
1.6
1.2
0 40 100
VO = VCC/2
RL = 1 kΩ
2.8
–40 60 80
2.6
Transresistance – MΩ
VCC = ±5 V
VCC = ±15 V
VCC = ±10 V
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
8POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 10
CLOSED-LOOP OUTPUT IMPEDANCE
vs
FREQUENCY
10
1
0.1
0.01 1M f – Frequency – Hz
100k 10M 100M
100
Closed-Loop Output Impedance –
VCC = ±15 V
RF = 750 Ω
Gain = +2
TA = 25°C
VI(PP) = 2 V
1G
Ω
VO
+
–
50 Ω
750 Ω1 kΩVI
THS300x
750 Ω
(VO
VI
=
1000
Zo)
– 1
Figure 11
f – Frequency – Hz
VOLTAGE NOISE AND CURRENT NOISE
vs
FREQUENCY
100
10
1100 10k1k 100k10
1000 VCC = ±15 V and ±5 V
TA = 25°C
In–
nV/ Hz– Voltage Noise –VnIn– Current Noise – pA/ Hz
and
In+
Vn
Figure 12
f – Frequency – Hz
PSRR – Power Supply Rejection Ratio – dB
POWER SUPPLY REJECTION RATIO
vs
FREQUENCY
1k 10k 10M 100M1M100k
60
40
20
0
50
30
10
80
90
70
VCC = ±5 V
VCC = ±15 V
G = 1
RF = 1 kΩ
VCC = ±5 V
VCC = ±15 V
–PSRR
+PSRR
Figure 13
TA – Free-Air Temperature – °C
POWER SUPPLY REJECTION RATIO
vs
FREE-AIR TEMPERATURE
80
70 –20 20
85
75
0 40 100–40 60 80
90
PSRR – Power Supply Rejection Ratio – dB
VCC = +5 V
VCC = +15 V
VCC = –5 V
VCC = –15 V
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 14
|VCC| – Supply Voltage – V
SLEW RATE
vs
SUPPLY VOLTAGE
4000
3000
2000
1000 711913
6000
5 15
5000
G = +5
RL = 150 Ω
tr/tf = 300 ps
RF = 1 kΩ
SR – Slew Rate – V/µs
–SR
+SR
7000
Figure 15
VO(PP) – Output Step – V
SLEW RATE
vs
OUTPUT STEP
10000
100 515
1000
10 200
VCC = ±15 V
G = +5
RL = 150 Ω
tr/tf = 300 ps
RF = 1 kΩ
SR – Slew Rate – V/µs
+SR
–SR
Figure 16
VO(PP) – Output Step – V
SLEW RATE
vs
OUTPUT STEP
2000
100 13
1000
240 5
V
CC = ±5 V
G = +5
RL = 150 Ω
tr/tf = 300 ps
RF= 1 kΩ
–SR
+SR
SR – Slew Rate – V/µs
Figure 17
G – Gain – V/V
NORMALIZED SLEW RATE
vs
GAIN
1.3
1.1
0.9
0.7 24
1.2
1
0.8
35 101 67
–Gain
+Gain
89
1.5
1.4
VCC = ±5 V
VO(PP) = 4 V
RL = 150 Ω
RF = 1 kΩ
tr/tf = 300 ps
SR – Normalized Slew Rate – V/µs
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
10 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 18
VO(PP) – Peak-to-Peak Output Voltage Swing – V
HARMONIC DISTORTION
vs
PEAK-TO-PEAK OUTPUT VOLTAGE SWING
0 2 4 6 12 14108 16
–55
–65
–75
–85
–60
–70
–80
–50
18
2nd Harmonic
3rd Harmonic
Harmonic Distortion – dBc
20
8 MHz
Gain = 2
VCC = ±15 V
RL = 150 Ω
RF = 750 Ω
Figure 19
VO(PP) – Peak-to-Peak Output Voltage Swing – V
HARMONIC DISTORTION
vs
PEAK-TO-PEAK OUTPUT VOLTAGE SWING
0 2 4 6 12 14108 16
–65
–75
–85
–95
–70
–80
–90
–50
18
2nd Harmonic
3rd Harmonic
Harmonic Distortion – dBc
20
4 MHz
Gain = 2
VCC = ±15 V
RL = 150 Ω
RF = 750 Ω
–55
–60
Figure 20
HARMONIC DISTORTION
vs
FREQUENCY
–70
–80
–90
–100
–75
–85
–95
2nd Harmonic
3rd Harmonic
Harmonic Distortion – dBc
Gain = 2
VCC = ±15 V
VO = 2 VPP
RL = 150 Ω
RF = 750 Ω
100k 1M 10M
f – Frequency – Hz
Figure 21
HARMONIC DISTORTION
vs
FREQUENCY
–70
–80
–90
–100
–75
–85
–95
2nd Harmonic
3rd Harmonic
Harmonic Distortion – dBc
Gain = 2
VCC = ±5 V
VO = 2 VPP
RL = 150 Ω
RF = 750 Ω
100k 1M 10M
f – Frequency – Hz
–60
–65
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
11
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 22
DIFFERENTIAL GAIN
vs
LOADING
1234 78
Number of 150 Ω Loads
65
VCC = ±15 V
0.04
0.02
0
0.03
0.01
VCC = ±5 V
Differential Gain – %
Gain = 2
RF = 750 Ω
40 IRE NTSC Modulation
Worst Case: ±100 IRE Ramp
Figure 23
DIFFERENTIAL GAIN
vs
LOADING
1234 78
Number of 150 Ω Loads
65
VCC = ±15 V
0.04
0.02
0
0.03
0.01
VCC = ±5 V
Differential Gain – %
Gain = 2
RF = 750 Ω
40 IRE PAL Modulation
Worst Case: ±100 IRE Ramp
Figure 24
DIFFERENTIAL PHASE
vs
LOADING
1234 78
Number of 150 Ω Loads
65
VCC = ±15 V
0.3
0.1
0
0.15
0.05
VCC = ±5 V
Differential Phase – Degrees
Gain = 2
RF = 750 Ω
40 IRE NTSC Modulation
Worst Case: ±100 IRE Ramp
0.2
0.25
Figure 25
DIFFERENTIAL PHASE
vs
LOADING
1234 78
Number of 150 Ω Loads
65
VCC = ±15 V
0.35
0.1
0
0.15
0.05
VCC = ±5 V
Differential Phase – Degrees
Gain = 2
RF = 750 Ω
40 IRE PAL Modulation
Worst Case: ±100 IRE Ramp
0.2
0.25
0.3
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
12 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 26
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
0
–2
–4
–6 1M 100M
1
–1
–3
–5
10M 1G100k
2
3
RF = 750 Ω
Output Amplitude – dB
RF = 1.5 kΩ
Gain = 1
VCC = ±15 V
RL = 150 Ω
VI = 200 mV RMS
RF = 1 kΩ
Figure 27
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
0
–2
–4
–6 1M 100M
1
–1
–3
–5
10M 1G100k
2
3
RF = 750 Ω
Output Amplitude – dB
RF = 1.5 kΩ
Gain = 1
VCC = ±5 V
RL = 150 Ω
VI = 200 mV RMS
RF = 1 kΩ
Figure 28
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
5
3
1
–1 1M 100M
6
4
2
0
10M 1G100k
7
8RF = 560 Ω
Output Amplitude – dB
RF = 1 kΩ
Gain = 2
VCC = ±15 V
RL = 150 Ω
VI = 200 mV RMS
RF = 680 Ω
9
Figure 29
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
5
3
1
–1 1M 100M
6
4
2
0
10M 1G100k
7
8RF = 560 Ω
Output Amplitude – dB
RF = 1 kΩ
Gain = 2
VCC = ±5 V
RL = 150 Ω
VI = 200 mV RMS
RF = 750 Ω
9
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
13
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
50
30
10
–10 1M 100M
60
40
20
0
10M 1G100k
70
G = +1000
RF = 10 kΩ
RL = 150 Ω
VO = 200 mV RMS
VCC = ±5 V
VCC = ±15 V
Output Amplitude – dB
Figure 30
Figure 31
f – Frequency – Hz
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
0
–2
–4
–6 1M 100M
1
–1
–3
–5
10M 1G100k
2
3
RF = 560 Ω
Normalized Output Response – dB
RF = 680 Ω
RF = 1 kΩ
Gain = –1
VCC = ±15 V
RL = 150 Ω
VI = 200 mV RMS
Figure 32
f – Frequency – Hz
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
0
–2
–4
–6 1M 100M
1
–1
–3
–5
10M 1G100k
2
3
RF = 560 Ω
Normalized Output Response – dB
RF = 750 Ω
RF = 1 kΩ
Gain = –1
VCC = ±5 V
RL = 150 Ω
VI = 200 mV RMS
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
14 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 33
f – Frequency – Hz
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
3
–3
–9
–15 1M 100M
0
–6
–12
10M 1G100k
Gain = +5
VCC = ±15 V
RL = 150 Ω
VO = 200 mV RMS
RF = 390 Ω
Normalized Output Response – dB
RF = 560 Ω
RF = 1 kΩ
Figure 34
f – Frequency – Hz
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
–2
–6
–10
–14 1M 100M
0
–4
–8
–12
10M 1G100k
2
4
Gain = +5
VCC = ±5 V
RL = 150 Ω
VO = 200 mV RMS
RF = 390 Ω
Normalized Output Response – dB
RF = 620 Ω
RF = 1 kΩ
Figure 35
f – Frequency – Hz
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
–12
–18
–24
–30 1M 100M
–9
–15
–21
–27
10M 1G100k
–6
–3
Gain = 1
VCC = ±15 V
RF = 1 kΩ
RL = 150 Ω
VI = 500 mV
VI = 250 mV
VI = 125 mV
VI = 62.5 mV
Output Level – dBV
Figure 36
f – Frequency – Hz
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
–6
–12
–18
–24 1M 100M
–3
–9
–15
–21
10M 1G100k
0
3
Output Level – dBV
Gain = 2
VCC = ±15 V
RF = 680 Ω
RL = 150 Ω
VI = 500 mV
VI = 250 mV
VI = 125 mV
VI = 62.5 mV
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
15
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 37
t – Time – ns
Gain = 1
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf = 300 ps
SMALL SIGNAL PULSE RESPONSE
–200
0
100
–100
100
–100
0302010 40 50 7060 80 90 100
300
–200
–300
– Output Voltage – V
VOVI– Input Voltage – mV
200
Figure 38
t – Time – ns
Gain = 5
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf = 300 ps
SMALL SIGNAL PULSE RESPONSE
–60
0
100
–100
20
–20
0302010 40 50 7060 80 90 100
60
–200
–300
– Output Voltage – mV
VOVI– Input Voltage – mV
200
Figure 39
t – Time – ns
Gain = +1
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 2.5 ns
LARGE SIGNAL PULSE RESPONSE
–3
0
1
–1
1
–1
0302010 40 50 7060 80 90 100
3
–2
–3
– Output Voltage – V
VOVI– Input Voltage – V
2
Figure 40
t – Time – ns
Gain = 1
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 2.5 ns
LARGE SIGNAL PULSE RESPONSE
–3
0
1
–1
1
–1
0302010 40 50 7060 80 90 100
3
–2
–3
– Output Voltage – V
VOVI– Input Voltage – V
2
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
16 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 41
t – Time – ns
Gain = +5
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 300 ps
LARGE SIGNAL PULSE RESPONSE
–3
0
5
–5
1
–1
0302010 40 50 7060 80 90 100
3
–10
–15
– Output Voltage – V
VOVI– Input Voltage – V
10
Figure 42
t – Time – ns
Gain = 5
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 300 ps
LARGE SIGNAL PULSE RESPONSE
–600
0
1
–1
200
–200
0302010 40 50 7060 80 90 100
600
–2
–3
– Output Voltage – V
VOVI– Input Voltage – mV
2
Figure 43
t – Time – ns
Gain = –1
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 2.5 ns
LARGE SIGNAL PULSE RESPONSE
2
0
1
–1
1
–1
0302010 40 50 7060 80 90 100
3
–2
–3
– Output Voltage – V
VOVI– Input Voltage – V
Figure 44
t – Time – ns
Gain = –1
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 300 ps
LARGE SIGNAL PULSE RESPONSE
2
0
1
–1
1
–1
0302010 40 50 7060 80 90 100
3
–2
–3
– Output Voltage – V
VOVI– Input Voltage – V
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
17
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 45
t – Time – ns
Gain = –5
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 300 ps
LARGE SIGNAL PULSE RESPONSE
–600
0
1
–1
200
–200
0302010 40 50 7060 80 90 100
600
–2
–3
– Output Voltage – V
VOVI– Input Voltage – mV
2
Figure 46
t – Time – ns
Gain = –5
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf= 300 ps
LARGE SIGNAL PULSE RESPONSE
–2
0
5
–5
1
–1
0302010 40 50 7060 80 90 100
3
–10
–15
– Output Voltage – V
VOVI– Input Voltage – V
10
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
18 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
theory of operation
The THS300x is a high-speed, operational amplifier configured in a voltage-feedback architecture. The device
is built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors
possessing fTs of several GHz. This configuration implements an exceptionally high-performance amplifier that
has a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic is shown in
Figure 47.
IN+ IN–
VCC+
VCC–
OUT
32 6
7
4
I
IB
IIB
Figure 47. Simplified Schematic
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
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APPLICATION INFORMATION
recommended feedback and gain resistor values
The THS300x is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This
process provides the excellent isolation and extremely high slew rates that result in superior distortion
characteristics.
As with all current-feedback amplifiers, the bandwidth of the THS300x is an inversely proportional function of
the value of the feedback resistor (see Figures 26 to 34). The recommended resistors for the optimum frequency
response are shown in Table 1. These should be used as a starting point and once optimum values are found,
1% tolerance resistors should be used to maintain frequency response characteristics. For most applications,
a feedback resistor value of 1 kΩ is recommended – a good compromise between bandwidth and phase margin
that yields a very stable amplifier.
Consistent with current-feedback amplifiers, increasing the gain is best accomplished by changing the gain
resistor , not the feedback resistor . This is because the bandwidth of the amplifier is dominated by the feedback
resistor value and internal dominant-pole capacitor . The ability to control the amplifier gain independent of the
bandwidth constitutes a major advantage of current-feedback amplifiers over conventional voltage-feedback
amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value
of the gain resistor to increase or decrease the overall amplifier gain.
Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance
decreases the loop gain and increases the distortion. It is also important to know that decreasing load
impedance increases total harmonic distortion (THD). Typically, the third-order harmonic distortion increases
more than the second-order harmonic distortion.
Table 1. Recommended Resistor Values for Optimum Frequency Response
GAIN RF for VCC = ±15 V RF for VCC = ±5 V
11 kΩ1 kΩ
2, –1 680 Ω750 Ω
–2 620 Ω620 Ω
5 560 Ω620 Ω
offset voltage
The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times
the corresponding gains. The following schematic and formula can be used to calculate the output offset
voltage:
VOO
+
VIO
ǒ
1
)ǒ
RF
RG
ǓǓ"
IIB
)
RS
ǒ
1
)ǒ
RF
RG
ǓǓ"
IIB– RF
+
–
VIO +
RG
RS
RF
IIB–
VO
IIB+
Figure 48. Output Offset Voltage Model
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
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APPLICATION INFORMATION
noise calculations and noise figure
Noise can cause errors on very small signals. This is especially true for amplifying small signals coming over
a transmission line or an antenna. The noise model for current-feedback amplifiers (CFB) is the same as for
voltage feedback amplifiers (VFB). The only difference between the two is that CFB amplifiers generally specify
different current-noise parameters for each input, while VFB amplifiers usually only specify one noise-current
parameter. The noise model is shown in Figure 49. This model includes all of the noise sources as follows:
•en = amplifier internal voltage noise (nV/√Hz)
•IN+ = noninverting current noise (pA/√Hz)
•IN– = inverting current noise (pA/√Hz)
•eRx = thermal voltage noise associated with each resistor (eRx = 4 kTRx)
_
+
RF
RS
RG
eRg
eRf
eRs en
IN+
Noiseless
IN–
eni eno
Figure 49. Noise Model
The total equivalent input noise density (eni) is calculated by using the following equation:
eni
+ǒ
en
Ǔ
2
)ǒ
IN
)
RS
Ǔ
2
)ǒ
IN–
ǒ
RF
ø
RG
ǓǓ
2
)
4kTR
s
)
4kT
ǒ
R
F
ø
R
G
Ǔ
Ǹ
Where: k = Boltzmann’s constant = 1.380658 × 10–23
T = temperature in degrees Kelvin (273 +°C)
RF || RG = parallel resistance of RF and RG
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the
overall amplifier gain (AV).
eno
+
eni AV
+
eni
ǒ
1
)
RF
RG
Ǔ
(Noninverting Case)
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
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APPLICATION INFORMATION
noise calculations and noise figure (continued)
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the
closed-loop gain is increased (by reducing RG), the input noise is reduced considerably because of the parallel
resistance term. This leads to the general conclusion that the most dominant noise sources are the source
resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares
method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly
simplify the formula and make noise calculations much easier.
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise
figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be
defined and is typically 50 Ω in RF applications.
NF
+
10log
ȧ
ȱ
Ȳ
e2
ni
eRs2
ȧ
ȳ
ȴ
Because the dominant noise components are generally the source resistance and the internal amplifier noise
voltage, we can approximate noise figure as:
NF
+
10log
ȧ
ȧ
ȧ
ȧ
ȧ
ȱ
Ȳ
1
)
ȧ
ȡ
Ȣǒ
en
Ǔ
2
)ǒ
IN
)
RS
Ǔ
2
ȧ
ȣ
Ȥ
4kTR
S
ȧ
ȧ
ȧ
ȧ
ȧ
ȳ
ȴ
The Figure 50 shows the noise figure graph for the THS300x.
NOISE FIGURE
vs
SOURCE RESISTANCE
10 100 10k
RS – Source Resistance – Ω
1k
f = 10 kHz
TA = 25°C
12
8
4
0
10
6
2
20
16
18
14
Noise Figure – dB
Figure 50. Noise Figure vs Source Resistance
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
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APPLICATION INFORMATION
slew rate
The slew rate performance of a current-feedback amplifier, like the THS300x, is affected by many different
factors. Some of these factors are external to the device, such as amplifier configuration and PCB parasitics,
and others are internal to the device, such as available currents and node capacitance. Understanding some
of these factors should help the PCB designer arrive at a more optimum circuit with fewer problems.
Whether the THS300x is used in an inverting amplifier configuration or a noninverting configuration can impact
the output slew rate. As can be seen from the specification tables as well as some of the figures in this data sheet,
slew-rate performance in the inverting configuration is faster than in the noninverting configuration. This is
because in the inverting configuration the input terminals of the amplifier are at a virtual ground and do not
significantly change voltage as the input changes. Consequently, the time to charge any capacitance on these
input nodes is less than for the noninverting configuration, where the input nodes actually do change in voltage
an amount equal to the size of the input step. In addition, any PCB parasitic capacitance on the input nodes
degrades the slew rate further simply because there is more capacitance to charge. Also, if the supply voltage
(VCC) to the amplifier is reduced, slew rate decreases because there is less current available within the amplifier
to charge the capacitance on the input nodes as well as other internal nodes.
Internally, the THS300x has other factors that impact the slew rate. The amplifier’s behavior during the slew-rate
transition varies slightly depending upon the rise time of the input. This is because of the way the input stage
handles faster and faster input edges. Slew rates (as measured at the amplifier output) of less than about
1500 V/µs are processed by the input stage in a very linear fashion. Consequently, the output waveform
smoothly transitions between initial and final voltage levels. This is shown in Figure 51. For slew rates greater
than 1500 V/µs, additional slew-enhancing transistors present in the input stage begin to turn on to support
these faster signals. The result is an amplifier with extremely fast slew-rate capabilities. Figures 41 and 52 show
waveforms for these faster slew rates. The additional aberrations present in the output waveform with these
faster-slewing input signals are due to the brief saturation of the internal current mirrors. This phenomenon,
which typically lasts less than 20 ns, is considered normal operation and is not detrimental to the device in any
way . If for any reason this type of response is not desired, then increasing the feedback resistor or slowing down
the input-signal slew rate reduces the effect.
Figure 51
t – Time – ns
SR = 1500 V/µs
Gain = 5
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf = 10 ns
SLEW RATE
10
0
5
–5
2
0
0604020 80 100 140120 160 180 200
4
–10
–15
– Output Voltage – V
VOVI– Input Voltage – V
Figure 52
t – Time – ns
SR = 2400 V/µs
Gain = 5
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
tr/tf = 5 ns
SLEW RATE
–2
0
5
–5
2
0
0604020 80 100 140120 160 180 200
4
–10
–15
– Output Voltage – V
VOVI– Input Voltage – V
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
23
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
driving a capacitive load
Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS300x has been internally compensated to maximize its bandwidth and
slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high-frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 53. A minimum value of 20 Ω should work well for most applications. For
example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
+
_
THS300x
CLOAD
1 kΩ
Input
Output
1 kΩ
20 Ω
Figure 53. Driving a Capacitive Load
PCB design considerations
Proper PCB design techniques in two areas are important to assure proper operation of the THS300x. These
areas are high-speed layout techniques and thermal-management techniques. Because the THS300x is a
high-speed part, the following guidelines are recommended.
D
Ground plane – It is essential that a ground plane be used on the board to provide all components with a
low inductive ground connection. Although a ground connection directly to a terminal of the THS300x is not
necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves
two functions: it provides a low inductive ground to the device substrate to minimize internal crosstalk, and
it provides the path for heat removal.
D
Input stray capacitance – To minimize potential problems with amplifier oscillation, the capacitance at the
inverting input of the amplifiers must be kept to a minimum. To do this, PCB trace runs to the inverting input
must be as short as possible, the ground plane must be removed under any etch runs connected to the
inverting input, and external components should be placed as close as possible to the inverting input. This
is especially true in the noninverting configuration. An example of this can be seen in Figure 54, which shows
what happens when a 1-pF capacitor is added to the inverting input terminal. The bandwidth increases at
the expense of peaking. This is because some of the error current is flowing through the stray capacitor
instead of the inverting node of the amplifier. Although, while the device is in the inverting mode, stray
capacitance at the inverting input has a minimal effect. This is because the inverting node is at a
virtual
ground
and the voltage does not fluctuate nearly as much as in the noninverting configuration. This can be
seen in Figure 55, where a 10-pF capacitor adds only 0.35 dB of peaking. In general, as the gain of the
system increases, the output peaking due to this capacitor decreases. While this can initially look like a
faster and better system, overshoot and ringing are more likely to occur under fast transient conditions. So
proper analysis of adding a capacitor to the inverting input node should be performed for stable operation.
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
24 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
PCB design considerations (continued)
Figure 54
OUTPUT AMPLITUDE
vs
FREQUENCY
2
0
–2
–4 1M f – Frequency – Hz
3
1
–1
–3
100k 10M 100M
4
6
Output Amplitude – dB
5
7
1G
Gain = 1
VCC = ±15 V
VO = 200 mV RMS
CI = 1 pF
CI = 0 pF
(Stray C Only)
+
–
1 kΩ
50 ΩRL =
150 Ω
Cin
Vin Vout
Figure 55
OUTPUT AMPLITUDE
vs
FREQUENCY
–2
–4
–6
–8 1M f – Frequency – Hz
–1
–3
–5
–7
100k 10M 100M
0
Output Amplitude – dB
1
1G
Gain = –1
VCC = ±15 V
VO = 200 mV RMS
CI = 10 pF
CI = Stray C Only
+
–
1 kΩ
RL =
150 Ω
Cin
Vin
Vout
1 kΩ
50 Ω
D
Proper power-supply decoupling – Use a minimum 6.8-µF tantalum capacitor in parallel with a 0.1-µF
ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several
amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the
supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible
to the supply terminal. As this distance increases, the inductance in the connecting etch makes the capacitor
less effective. The designer should strive for distances of less than 0.1 inches between the device power
terminal and the ceramic capacitors.
thermal information
The THS300x incorporates output-current-limiting protection. Should the output become shorted to ground, the
output current is automatically limited to the value given in the data sheet. While this protects the output against
excessive current, the device internal power dissipation increases due to the high current and large voltage drop
across the output transistors. Continuous output shorts are not recommended and could damage the device.
Additionally, connection of the amplifier output to one of the supply rails (±VCC) is not recommended. Failure
of the device is possible under this condition and should be avoided. But, the THS300x does not incorporate
thermal-shutdown protection. Because of this, special attention must be paid to the device’s power dissipation
or failure may result.
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
25
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
thermal information (continued)
The thermal coefficient θJA is approximately 169°C/W for the SOIC 8-pin D package. For a given θJA, the
maximum power dissipation, shown in Figure 56, is calculated by the following formula:
PD
+ǒ
TMAX–TA
q
JA
Ǔ
Where: PD= Maximum power dissipation of THS300x (watts)
TMAX= Absolute maximum junction temperature (150°C)
TA= Free-ambient air temperature (°C)
θJA = Thermal coefficient from die junction to ambient air (°C/W)
TA – Free-Air Temperature – °C
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
1
0–20 20
1.5
0.5
0 40 100–40 60 80
PD– Maximum Power Dissipation – W
SOIC-D Package:
θJA = 169°C/W
TJ = 150°C
No Airflow
Figure 56. Maximum Power Dissipation vs Free-Air Temperature
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
26 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
general configurations
A common error for the first-time CFB user is the creation of a unity gain buffer amplifier by shorting the output
directly to the inverting input. A CFB amplifier in this configuration will oscillate and is not recommended. The
THS300x, like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing
capacitors directly from the output to the inverting input is not recommended. This is because, at high
frequencies, a capacitor has a very low impedance. This results in an unstable amplifier and should not be
considered when using a current-feedback amplifier. Because of this, integrators and simple low-pass filters,
which are easily implemented on a VFB amplifier , have to be designed slightly differently. If filtering is required,
simply place an RC-filter at the noninverting terminal of the operational-amplifier (see Figure 57).
VIVO
C1
+
–
RGRF
R1
f–3dB
+
1
2
p
R1C1
VO
VI
+ǒ
1
)
RF
RG
Ǔǒ
1
1
)
sR1C1
Ǔ
Figure 57. Single-Pole Low-Pass Filter
If a multiple-pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is
because the filtering elements are not in the negative feedback loop and stability is not compromised. Because
of their high slew-rates and high bandwidths, CFB amplifiers can create very accurate signals and help minimize
distortion. An example is shown in Figure 58.
VI
C2
R2R1
C1
RF
RG
R1 = R2 = R
C1 = C2 = C
Q = Peaking Factor
(Butterworth Q = 0.707)
(
=1
Q
2 – )
RGRF
_
+f–3dB
+
1
2
p
RC
Figure 58. 2-Pole Low-Pass Sallen-Key Filter
There are two simple ways to create an integrator with a CFB amplifier. The first, shown in Figure 59, adds a
resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant
and the feedback impedance never drops below the resistor value. The second, shown in Figure 60, uses
positive feedback to create the integration. Caution is advised because oscillations can occur due to the positive
feedback.
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
27
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
general configurations (continued)
+
–
C1
RF
RG
VO
VI
THS300x
VO
VI
+ǒ
RF
RG
Ǔȧ
ȡ
Ȣ
S
)
1
RFC1
S
ȧ
ȣ
Ȥ
Figure 59. Inverting CFB Integrator
+
–
RF
VO
RG
R2R1
C1
RA
VI
THS300x
For Stable Operation:
R2
R1 || RA≥RF
RG
sR1C1
()
RF
RG
1 +
VO ≅ VI
Figure 60. Noninverting CFB Integrator
The THS300x may also be employed as a very good video distribution amplifier. One characteristic of
distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are compromised
as the number of lines increases and the closed-loop gain increases (see Figures 22 to 25 for more information).
Be sure to use termination resistors throughout the distribution system to minimize reflections and capacitive
loading.
+
–
750 Ω750 Ω
75 Ω75 Ω
75 Ω
75 Ω
75 Ω
N Lines
VO1
VON
THS300x
75-Ω Transmission Line
VI
Figure 61. Video Distribution Amplifier Application
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
28 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
evaluation board
Evaluation boards are available for the THS3001 (literature #SLOP130) and the THS3002 (literature
#SLOP241). The boards have been configured for very low parasitic capacitance in order to realize the full
performance of the amplifier. Schematics of the evaluation boards are shown in Figures 62 and 63. The circuitry
has been designed so that the amplifier may be used in either an inverting or noninverting configuration. T o order
the evaluation board contact your local TI sales office or distributor. For more detailed information, refer to the
THS3001 EVM User’s Manual
(literature #SLOV021) or the
THS3002 EVM User’s Guide
(literature #SLOVxxx).
To order the evaluation board, contact your local TI sales office or distributor.
_
+
THS3001
VCC–
VCC+
C3
6.8 µF
C4
0.1 µF
C1
6.8 µF
C2
0.1 µF
R1
1 kΩ
R5
1 kΩ
R3
49.9 Ω
R2
49.9 Ω
R4
49.9 Ω
IN–
IN+
OUT
+
+
Figure 62. THS3001 Evaluation Board Schematic
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
29
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
evaluation board (continued)
R3
100 Ω
–
+1
2
3
R4
100 Ω
R5
R1
100 Ω
R2
0 Ω
R6
301 Ω
C3
C5
0.1 µF
VCC+
C4
0.1 µF
VCC–
R11
100 Ω
–
+
7
6
5
R12
100 Ω
R13
R9
100 Ω
R10
0 Ω
R14
301 Ω
C6
4
8R7
49.9 Ω
R15
49.9 Ω
OUT1
OUT2
R8
THS3002
U1:B
THS3002
U1:A
Figure 63. THS3002 Evaluation Board Schematic
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
30 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL INFORMATION
D (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE
14 PIN SHOWN
4040047/D 10/96
0.228 (5,80)
0.244 (6,20)
0.069 (1,75) MAX 0.010 (0,25)
0.004 (0,10)
1
14
0.014 (0,35)
0.020 (0,51)
A
0.157 (4,00)
0.150 (3,81)
7
8
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.010 (0,25)
PINS **
0.008 (0,20) NOM
A MIN
A MAX
DIM
Gage Plane
0.189
(4,80)
(5,00)
0.197
8
(8,55)
(8,75)
0.337
14
0.344
(9,80)
16
0.394
(10,00)
0.386
0.004 (0,10)
M
0.010 (0,25)
0.050 (1,27)
0°–8°
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
D. Falls within JEDEC MS-012
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
31
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL INFORMATION
DGN (S-PDSO-G8) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
0,69
0,41
0,25
Thermal Pad
(See Note D)
0,15 NOM
Gage Plane
4073271/A 01/98
4,98
0,25
5
3,05 4,78
2,95
8
4
3,05
2,95
1
0,38
0,15
0,05
1,07 MAX
Seating Plane
0,10
0,65 M
0,25
0°–6°
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions include mold flash or protrusions.
D. The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad. This pad is electrically
and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MO-187
PowerPAD is a trademark of Texas Instruments Incorporated.
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CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
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Copyright 1999, Texas Instruments Incorporated
