R3
D1
C1
R1
VOUT
R2
VIN
C4
C3
C2
L1
R4
TPS61175
VIN
EN
PGND
AGND
Syn
FREQ
COMP
SS
NC
FB
PGND
PGND
SW
SW
TPS61175
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SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
3-A High Voltage Boost Converter with Soft-start and Programmable Switching Frequency
Check for Samples: TPS61175
1FEATURES DESCRIPTION
The TPS61175 is a monolithic switching regulator
2• 2.9-V to 18-V Input Voltage Range with integrated 3-A, 40-V power switch. It can be
• 3A, 40V Internal Switch configured in several standard switching-regulator
• High Efficiency Power Conversion: Up to 93% topologies, including boost, SEPIC and flyback. The
device has a wide input voltage range to support
• Frequency Set by External Resistor: 200-kHz application with input voltage from multi-cell batteries
to 2.2-MHz or regulated 5-V, 12-V power rails.
• Synchronous External Switching Frequency The TPS61175 regulates the output voltage with
• User Defined Soft Start into Full Load current mode PWM (pulse width modulation) control.
• Skip-Switching Cycle for Output Regulation at The switching frequency of PWM is either set by an
Light Load external resistor or an external clock signal. The user
can program the switching frequency from 200-kHz to
• 14-pin HTSSOP Package with PowerPad™ 2.2-MHz.
APPLICATIONS The device features a programmable soft-start
function to limit inrush current during start-up, and
• 5V to 12V, 24V power conversion has built-in other protection features, such as pulse-
• Supports SEPIC, Flyback topology by-pulse over current limit and thermal shutdown. The
• ADSL Modems TPS61175 is available in 14-pin HTSSOP package
with Powerpad.
• TV Tuner
TYPICAL APPLICATION FOR BOOST CONVERTER
Figure 1. Typical Application
ORDERING INFORMATION(1)
TAPART NUMBER PACKAGE(2)
–40°C to 85°C TPS61175PWP HTSSOP-14
(1) For the most current package and ordering information, see the TI Web site at www.ti.com.
(2) The PWP package is available in tape and reel. Add R suffix (TPS61175PWPR) to order quantities of 2000 parts per reel. Without suffix,
the TPS61175PWP is shipped in tubes with 90 parts per tube.
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2PowerPad is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date. Copyright © 2008–2012, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS(1)
over operating free-air temperature range (unless otherwise noted) VALUE UNIT
Supply Voltages on pin VIN (2) –0.3 to 20 V
Voltages on pins EN(2) –0.3 to 20 V
Voltage on pin FB, FREQ and COMP(2) –0.3 to 3 V
Voltage on pin SYNC, SS(2) –0.3 to 7 V
Voltage on pin SW(2) –0.3 to 40 V
Continuous Power Dissipation See the Thermal Information Table
Operating Junction Temperature Range –40 to +150 °C
Storage Temperature Range –65 to +150 °C
(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
THERMAL INFORMATION TPS61175
THERMAL METRIC(1) UNITS
PWP 14 PINS
θJA Junction-to-ambient thermal resistance 45.2
θJCtop Junction-to-case (top) thermal resistance 34.9
θJB Junction-to-board thermal resistance 30.1 °C/W
ψJT Junction-to-top characterization parameter 1.5
ψJB Junction-to-board characterization parameter 29.9
θJCbot Junction-to-case (bottom) thermal resistance 5.8
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted) MIN NOM MAX UNIT
VIN Input voltage range 2.9 18 V
VOOutput voltage range VIN 38 V
L Inductor(1) 4.7 47 μH
fSW Switching frequency 200 2200 kHz
CIInput Capacitor 4.7 μF
COOutput Capacitor 4.7 μF
VSYN External Switching Frequency Logic 5 V
TAOperating ambient temperature –40 85 °C
TJOperating junction temperature –40 125 °C
(1) The inductance value depends on the switching frequency and end application. While larger values may be used, values between 4.7-
μH and 47-μH have been successfully tested in various applications. Refer to the Inductor Selection for detail.
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SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
ELECTRICAL CHARACTERISTICS
FSW = 1.2 MHz (Rfreq=80 kΩ), Vin=3.6V, TA= –40°C to +85°C, typical values are at TA= 25°C (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
SUPPLY CURRENT
VIN Input voltage range 2.9 18 V
IQOperating quiescent current into Vin Device PWM switching without load 3.5 mA
ISD Shutdown current EN=GND 1.5 μA
VUVLO Under-voltage lockout threshold 2.5 2.7 V
Vhys Under-voltage lockout hysteresis 130 mV
ENABLE AND REFERENCE CONTROL
Venh EN logic high voltage Vin = 2.9 V to 18 V 1.2 V
Venl EN logic low voltage Vin = 2.9 V to 18 V 0.4 V
VSYNh SYN logic high voltage 1.2
VSYNl SYN logic low voltage 0.4 V
Ren EN pull down resistor 400 800 1600 kΩ
Toff Shutdown delay, SS discharge EN high to low 10 ms
VOLTAGE AND CURRENT CONTROL
VREF Voltage feedback regulation voltage 1.204 1.229 1.254 V
IFB Voltage feedback input bias current 200 nA
Isink Comp pin sink current VFB = VREF + 200 mV, VCOMP = 1 V 50 μA
Isource Comp pin source current VFB = VREF –200 mV, VCOMP = 1 V 130 μA
VCCLP Comp pin Clamp Voltage High Clamp, VFB = 1 V 3 V
Low Clamp, VFB = 1.5 V 0.75
VCTH Comp pin threshold Duty cycle = 0% 0.95 V
Gea Error amplifier transconductance 240 340 440 μmho
Rea Error amplifier output resistance 10 MΩ
fea Error amplifier crossover frequency 500 KHz
FREQUENCY
Rfreq = 480 kΩ0.16 0.21 0.26
fSOscillator frequency Rfreq = 80 kΩ1.0 1.2 1.4 MHz
Rfreq = 40 kΩ1.76 2.2 2.64
Dmax Maximum duty cycle VFB = 1.0 V, Rfreq = 80 kΩ89% 93%
VFREQ FREQ pin voltage 1.229 V
Tmin_on Minimum on pulse width Rfreq = 80 kΩ60 ns
POWER SWITCH
RDS(ON) N-channel MOSFET on-resistance VIN = VGS = 3.6 V 0.13 0.25 Ω
VIN = VGS = 3.0 V 0.13 0.3
ILN_NFET N-channel leakage current VDS = 40 V, TA= 25°C 1 μA
OC, OVP AND SS
ILIM N-Channel MOSFET current limit D = Dmax 3 3.8 5 A
ISS Soft start bias current Vss = 0 V 6 μA
THERMAL SHUTDOWN
Tshutdown Thermal shutdown threshold 160 °C
Thysteresis Thermal shutdown threshold hysteresis 15 °C
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 3
Product Folder Link(s): TPS61175
1
2
3
4
5
6
7
14
13
12
11
10
9
8
SW
SW
VIN
EN
SS
SYNC
AGND
PGND
PGND
PGND
NC
FREQ
FB
COMP
TSSOP 14-pin
(TOP VIEW)
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
DEVICE INFORMATION
PIN ASSIGNMENTS
PIN FUNCTIONS
PIN DESCRIPTION
I/O
NAME NO.
VIN 3 I The input supply pin for the IC. Connect VIN to a supply voltage between 2.9V and 18V. It is acceptable for
the voltage on the pin to be different from the boost power stage input for applications requiring voltage
beyond VIN range.
SW 1,2 I This is the switching node of the IC. Connect SW to the switched side of the inductor.
FB 9 I Feedback pin for positive voltage regulation. Connect to the center tap of a resistor divider to program the
output voltage.
EN 4 I Enable pin. When the voltage of this pin falls below the enable threshold for more than 10ms, the IC turns
off.
COMP 8 O Output of the internal transconductance error amplifier. An external RC network is connected to this pin to
compensate the regulator.
SS 5 O Soft start programming pin. A capacitor between the SS pin and GND pin programs soft start timing. See
application section for information on how to size the SS capacitor.
FREQ 10 O Switch frequency program pin. An external resistor is connected to this pin to set switch frequency. See
application section for information on how to size the FREQ resistor.
AGND 7 I Signal ground of the IC
PGND 12,13,14 I Power ground of the IC. It is connected to the source of the PWM switch.
SYNC 6 I Switch frequency synchronous pin. Customers can use an external signal to set the IC switch frequency
between 200-kHz and 2.2-MHz. If not used, this pin should be tied to AGND as short as possbile to avoid
noise coupling.
NC 11 I Reserved pin. Must connect this pin to ground.
Thermal Pad The thermal pad should be soldered to the analog ground. If possible, use thermal via to connect to top and
internal ground plane layers for ideal power dissipation.
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SW
Ramp
Generator
Oscillator
Current
Sensor
EN
PGND
C4
L1
+
EA
R2
C2
VIN
PWM Control
COMP
R1
R3
R4
C3
SS FREQ SYNC
FB
FB
Gate
Driver
C1
AGND
D1
1.229 V
Reference
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
FUNCTIONAL BLOCK DIAGRAM
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
Circuit of Figure 1; L1 = D104C2-10μH; D1 = SS3P6L-E3/86A, R4 = 80kΩ, R3 = 10kΩ, C4 = 22nF, FIGURE
C2 = 10μF;Vin = 5V, Vout = 24V, Iout = 200mA; unless otherwise noted
Efficiency VIN = 5V, Vout = 12V, 24V, 35V 2
Efficiency VIN = 5V, 12V; Vout = 24V 3
Error amplifier transconductance vs Temperature 4
Switch current limit vs Temperature 5
Switch current limit vs Duty cycle 6
FB accuracy vs Temperature 7
Line transient response Vin = 4.5 V to 5 V 8
Load transient response Iout = 100 mA to 300 mA; refer to 'compensating the control loop' for optimization 9
PWM Operation 10
Pulse skipping No load 11
Start-up C3 = 47 nF 12
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Product Folder Link(s): TPS61175
50
60
70
80
90
100
Efficiency-%
0 0.2 0.4 0.6 0.8 1 1.2
I -OutputCurrent- A
O
V =5V
I
V =12V
I
V =24V
O
0 0.2 0.4 0.6 0.8 1 1.2
I -OutputCurrent- A
O
50
60
70
80
90
100
Efficiency-%
V =5V
I
V =12V
O
V =24V
O
V =35V
O
320
340
360
380
400
-40 -20 0
20
40 60 80 100 120
T -Free-AirTemperature-°C
A
EA Transconductance-mhos
3
3.5
4
4.5
5
0.2 0.4 0.6 0.8 1
DutyCycle-%
OvercurrentLimit- A
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
EFFICIENCY EFFICIENCY
vs vs
OUTPUT CURRENT OUTPUT CURRENT
Figure 2. Figure 3.
ERROR AMPLIFIER TRANSCONDUCTANCE OVERCURRENT LIMIT
vs vs
FREE-AIR TEMPERATURE DUTY CYCLE
Figure 4. Figure 5.
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3.5
3.6
3.7
3.8
3.9
4
-40 -20 0 20 40 60 80 100 120
T -Free-AirTemperature-°C
A
OvercurrentLimit- A
1220
1225
1230
1235
1240
-40 -20 0
20
40 60 80 100 120
T -Free-AirTemperature-°C
A
FBVoltage-mV
VOUT
500mV/div
AC
I
200mA/div
LOAD
t-100 s/divm
VIN
1V/div
AC
VOUT
100mV/div
AC
t-200 s/divm
SW
20V/div
VOUT
100mV/div
AC
t-400ns/div
IL
500mA/div
VOUT
1V/div
20Voffset
I
100mA/div
L
t-400 s/divm
VOUT
20mV/div
AC
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
OVERCURRENT LIMIT FB VOLTAGE
vs vs
FREE-AIR TEMPERATURE FREE-AIR TEMPERATURE
Figure 6. Figure 7.
Line Transient Response Load Transient Response
Figure 8. Figure 9.
PWM Operation Pulse Skipping
Figure 10. Figure 11.
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 7
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EN
2V/div
VOUT
5V/div
I
500mA/div
L
t-1ms/div
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
Soft Startup
Figure 12.
DETAILED DESCRIPTION
OPERATION
The TPS61175 integrates a 40-V low side switch FET for up to 38-V output. The device regulates the output with
current mode PWM (pulse width modulation) control. The PWM control circuitry turns on the switch at the
beginning of each switching cycle. The input voltage is applied across the inductor and stores the energy as
inductor current ramps up. During this portion of the switching cycle, the load current is provided by the output
capacitor. When the inductor current rises to the threshold set by the error amplifier output, the power switch
turns off and the external Schottky diode is forward biased. The inductor transfers stored energy to replenish the
output capacitor and supply the load current. This operation repeats each every switching cycle. As shown in the
block diagram, the duty cycle of the converter is determined by the PWM control comparator which compares the
error amplifier output and the current signal. The switching frequency is programmed by the external resistor or
synchronized to an external clock signal.
A ramp signal from the oscillator is added to the current ramp to provide slope compensation. Slope
compensation is necessary to avoid subharmonic oscillation that is intrinsic to the current mode control at duty
cycle higher than 50%. If the inductor value is lower than 4.7μH, the slope compensation may not be adequate.
The feedback loop regulates the FB pin to a reference voltage through a transconductance error amplifier. The
output of the error amplifier is connected to the COMP pin. An external RC compensation network is connected
to the COMP pin to optimize the feedback loop for stability and transient response.
SWITCHING FREQUENCY
The switch frequency is set by a resistor (R4) connected to the FREQ pin of the TPS61175. Do not leave this pin
open. A resistor must always be connected for proper operation. See Table 1 and Figure 13 for resistor values
and corresponding frequencies.
Table 1. Switching Frequency vs External Resistor
R4 (kΩ) fSW (kHz)
443 240
256 400
176 600
80 1200
51 2000
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0
500
1000
1500
2000
2500
3000
3500
10 100 1000
ExternalResistor-kW
f-Frequency-kHz
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
Figure 13. Switching Frequency vs External Resistor
Alternatively, the TPS61175 switching frequency will synchronize to an external clock signal that is applied to the
SYNC pin. The logic level of the external clock is shown in the specification table. The duty cycle of the clock is
recommended in the range of 10% to 90%. The resistor also must be connected to the FREQ pin when IC is
switching by the external clock. The external clock frequency must be within ±20% of the corresponding
frequency set by the resistor. For example, if the corresponding frequency as set by a resistor on the FREQ pin
is 1.2-MHz, the external clock signal should be in the range of 0.96-MHz to 1.44-MHz.
If the external clock signal is higher than the frequency per the resistor on the FREQ pin, the maximum duty
cycle specification (DMAX) should be lowered by 2%. For instance, if the resistor set value is 2.5MHz, and the
external clock is 3MHz, DMAX is 87% instead of 89%.
SOFT START
The TPS61175 has a built-in soft start circuit which significantly reduces the start-up current spike and output
voltage overshoot. When the IC is enabled, an internal bias current (6-μA typically) charges a capacitor (C3) on
the SS pin. The voltage at the capacitor clamps the output of the internal error amplifier that determines the duty
cycle of PWM control, thereby the input inrush current is eliminated. Once the capacitor reaches 1.8-V, the soft
start cycle is completed and the soft start voltage no longer clamps the error amplifier output. Refer to Figure 13
for the soft start waveform. See Table 2 for C3 and corresponding soft start time. A 47-nF capacitor eliminates
the output overshoot and reduces the peak inductor current for most applications.
Table 2. Soft Start Time vs C3
Vin (V) Vout (V) Load (A) Cout (μF) fSW (MHz) C3 (nF) tSS(ms) Overshot (mV)
47 4 none
5 24 0.4 10 1.2 10 0.8 210
100 6.5 none
12 35 0.6 10 2 10 0.4 300
When the EN is pulled low for 10-ms, the IC enters shutdown and the SS capacitor discharges through a 5kΩ
resistor for the next soft start.
OVERCURRENT PROTECTION
The TPS61175 has a cycle-by-cycle overcurrent limit protection that turns off the power switch once the inductor
current reaches the overcurrent limit threshold. The PWM circuitry resets itself at the beginning of the next switch
cycle. During an overcurrent event, the output voltage begins to droop as a function of the load on the output.
When the FB voltage drops lower than 0.9-V, the switching frequency is automatically reduced to 1/4 of the set
value. The switching frequency does not reset until the overcurrent condition is removed. This feature is disabled
during soft start.
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SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
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ENABLE AND THERMAL SHUTDOWN
The TPS61175 enters shutdown when the EN voltage is less than 0.4-V for more than 10-ms. In shutdown, the
input supply current for the device is less than 1.5-μA (max). The EN pin has an internal 800-kΩpull down
resistor to disable the device when it is floating.
An internal thermal shutdown turns off the device when the typical junction temperature of 160°C is exceeded.
The IC restarts when the junction temperature drops by 15°C.
UNDER VOLTAGE LOCKOUT (UVLO)
An under voltage lockout circuit prevents mis-operation of the device at input voltages below 2.5-V (typical).
When the input voltage is below the under voltage threshold, the device remains off and the internal switch FET
is turned off. The under voltage lockout threshold is set below minimum operating voltage of 2.9V to avoid any
transient VIN dip triggering the UVLO and causing the device to reset. For the input voltages between UVLO
threshold and 2.9V, the device attempts to operate, but the specifications are not ensured.
MINIMUM ON TIME and PULSE SKIPPING
Once the PWM switch is turned on, the TPS61175 has minimum ON pulse width of 60-ns. This sets the limit of
the minimum duty cycle of the PWM switch, and it is independent of the set switching frequency. When operating
conditions result in the TPS61175 having a minimum ON pulse width less than 60-ns, the IC enters pulse-
skipping mode. In this mode, the device keeps the power switch off for several switching cycles to keep the
output voltage from rising above the regulated voltage. This operation typically occurs in light load condition
when the PWM operates in discontinuous mode. Pulse skipping increases the output voltage ripple, see the
Figure 14.
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Product Folder Link(s): TPS61175
R3
D1
C1
R1
VOUT
R2
VIN
C4
C3
C2
L1
R4
TPS61175
VIN
EN
PGND
AGND
Syn
FREQ
COMP
SS
NC
FB
PGND
PGND
SW
SW
OUT D IN
OUT D
V + V V
D =
V + V
-
OUT D OUT SW
2
IN
2 (V + V ) I L
D =
V
´ ´ ´ ´ ¦
2
OUT D IN IN
OUT(crit) 2
OUT D SW
(V + V V ) V
I =
2 (V + V ) L
- ´
´ ´ ¦ ´
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
APPLICATION INFORMATION
The following section provides a step-by-step design approach for configuring the TPS61175 as a voltage
regulating boost converter, as shown in Figure 14. When configured as SEPIC or flyback converter, a different
design approach is required.
Figure 14. Boost Converter Configuration
DETERMINING THE DUTY CYCLE
The TPS61175 has a maximum worst case duty cycle of 89% and a minimum on time of 60 ns. These two
constraints place limitations on the operating frequency that can be used for a given input to output conversion
ratio. The duty cycle at which the converter operates is dependent on the mode in which the converter is running.
If the converter is running in discontinuous conduction mode (DCM), where the inductor current ramps to zero at
the end of each cycle, the duty cycle varies with changes to the load much more than it does when running in
continuous conduction mode (CCM). In continuous conduction mode, where the inductor maintains a dc current,
the duty cycle is related primarily to the input and output voltages as computed below:
(1)
In discontinuous mode the duty cycle is a function of the load, input and output voltages, inductance and
switching frequency as computed below:
(2)
All converters using a diode as the freewheeling or catch component have a load current level at which they
transition from discontinuous conduction to continuous conduction. This is the point where the inductor current
just falls to zero. At higher load currents, the inductor current does not fall to zero but remains flowing in a
positive direction and assumes a trapezoidal wave shape as opposed to a triangular wave shape. This load
boundary between discontinuous conduction and continuous conduction can be found for a set of converter
parameters as follows.
(3)
For loads higher than the result of the equation above, the duty cycle is given by Equation 1 and for loads less
that the results of Equation 2, the duty cycle is given Equation 3. For Equation 1 through Equation 3, the variable
definitions are as follows.
• VOUT is the output voltage of the converter in V
• VDis the forward conduction voltage drop across the rectifier or catch diode in V
• VIN is the input voltage to the converter in V
• IOUT is the output current of the converter in A
• L is the inductor value in H
• fSW is the switching frequency in Hz
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OUT D ININ
L
SW S W
SW
O UT D IN IN
O UT
IN est
(V + V V ) (1 D)V D 1
I = = =
L L 1 1
L +
V + V V V
P
RPL% V η
- ´ -´
D´ ¦ ´ ¦ é ù
æ ö
´ ¦ ´
ê úç ÷
-
ê ú
è ø
ë û
£ ´ ´
est IN
S W OUT
OUT D IN IN
η V
L
1 1
+ RPL% P
V + V V V
´
£é ù
æ ö
¦ ´
ê ú
ç ÷
-
ê úè ø
ë û
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
Unless otherwise stated, the design equations that follow assume that the converter is running in continuous
mode.
SELECTING THE INDUCTOR
The selection of the inductor affects steady state operation as well as transient behavior and loop stability. These
factors make it the most important component in power regulator design. There are three important inductor
specifications, inductor value, DC resistance and saturation current. Considering inductor value alone is not
enough.
Inductor values can have ±20% tolerance with no current bias. When the inductor current approaches saturation
level, its inductance can fall to some percentage of its 0-A value depending on how the inductor vendor defines
saturation current. For CCM operation, the rule of thumb is to choose the inductor so that its inductor ripple
current (ΔIL) is no more than a certain percentage (RPL% = 20–40%) of its average DC value (IIN(AVG) = IL(AVG))
(4)
Rearranging and solving for L gives
(5)
Choosing the inductor ripple current to closer to 20% of the average inductor current results in a larger
inductance value, maximizes the converter’s potential output current and minimizes EMI. Choosing the inductor
ripple current closer to 40% of IL(AVG) results in a smaller inductance value, and a physically smaller inductor,
improves transient response but results in potentially higher EMI and lower efficiency if the DCR of the smaller
packaged inductor is significantly higher. Using an inductor with a smaller inductance value than computed
above may result in the converter operating in DCM. This reduces the boost converter’s maximum output current,
causes larger input voltage and output ripple and typically reduces efficiency. Table 3 lists the recommended
inductor for the TPS61175.
Table 3. Recommended Inductors for TPS61175
L DCR MAX SATURATION CURRENT SIZE
PART NUMBER VENDOR
(μH) (mΩ) (A) (L × W × H mm)
D104C2 10 44 3.6 10.4x10.4x4.8 TOKO
VLF10040 15 42 3.1 10.0x9.7x4.0 TDK
CDRH105RNP 22 61 2.9 10.5x10.3x5.1 Sumida
MSS1038 15 50 3.8 10.0x10.2x3.8 Coilcraft
The device has built-in slope compensation to avoid subharmonic oscillation associated with current mode
control. If the inductor value is lower than 4.7μH, the slope compensation may not be adequate, and the loop can
be unstable. Applications requiring inductors above 47μH have not been evaluated. Therefore, the user is
responsible for verifying operation if they select an inductor that is outside the 4.7μH–47μH recommended range.
COMPUTING THE MAXIMUM OUTPUT CURRENT
The over-current limit for the integrated power FET limits the maximum input current and thus the maximum input
power for a given input voltage. Maximum output power is less than maximum input power due to power
conversion losses. Therefore, the current limit setting, input voltage, output voltage and efficiency can all change
the maximum current output (IOUT(MAX)). The current limit clamps the peak inductor current, therefore the ripple
has to be subtracted to derive maximum DC current.
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Product Folder Link(s): TPS61175
IN(min) est IN(min) LIM est
OUT(max)
OUT OUT
V η V I (1 %RPL/2) η
I = =
V V
´ ´ ´ - ´
R1
Vout = 1.229 V + 1
R2
Vout
R1 = R2 1
1.229V
æ ö
´ç ÷
è ø
æ ö
´ -
ç ÷
è ø
( )
OUT IN out
out
OUT ripple
V V I
C = V Fs V
-
´ ´
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
(6)
where
ILIM = over current limit
ηest= efficiency estimate based on similar applications or computed above
For instance, when VIN = 12-V is boosted to VOUT = 24-V, the inductor is 10-uH, the Schottky forward voltage is
0.4-V and the switching frequency is 1.2-MHz; then the maximum output current is 1.2-A in typical condition,
assuming 90% efficiency and a %RPL = 20%.
SETTING OUTPUT VOLTAGE
To set the output voltage in either DCM or CCM, select the values of R1 and R2 according to the following
equation.
(7)
Considering the leakage current through the resistor divider and noise decoupling into FB pin, an optimum value
for R2 is around 10k. The output voltage tolerance depends on the VFB accuracy and the tolerance of R1 and
R2.
SETTING THE SWITCHING FREQUENCY
Choose the appropriate resistor from the resistance versus frequency table Table 1 or graph Figure 13. A resistor
must be placed from the FREQ pin to ground, even if an external oscillation is applied for synchronization.
Increasing switching frequency reduces the value of external capacitors and inductors, but also reduces the
power conversion efficiency. The user should set the frequency for the minimum tolerable efficiency.
SETTING THE SOFT START TIME
Choose the appropriate capacitor from the soft start table Table 2. Increasing the soft start time reduces the
overshoot during start-up.
SELECTING THE SCHOTTKY DIODE
The high switching frequency of the TPS61175 demands a high-speed rectification for optimum efficiency.
Ensure that the diode’s average and peak current rating exceed the average output current and peak inductor
current. In addition, the diode’s reverse breakdown voltage must exceed the switch FET rating voltage of 40V.
So, the VISHAY SS3P6L-E3/86A is recommended for TPS61175. The power dissipation of the diode's package
must be larger than IOUT(max) x VD
SELECTING THE INPUT AND OUTPUT CAPACITORS
The output capacitor is mainly selected to meet the requirements for the output ripple and load transient. Then
the loop is compensated for the output capacitor selected. The output ripple voltage is related to the capacitor’s
capacitance and its equivalent series resistance (ESR). Assuming a capacitor with zero ESR, the minimum
capacitance needed for a given ripple can be calculated by
(8)
where, Vripple= peak to peak output ripple. The additional output ripple component caused by ESR is calculated
using:
Vripple_ESR = I × RESR
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Link(s): TPS61175
TRAN
OUT
LOOP-BW TRAN
ΔI
C =
2 f ΔV´ p ´ ´
_
+
(1-D)
RSENSE
R1
C2 R
2
O
RESR
R2
Vref
C5
C4
R3
(optional)
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
Due to its low ESR, Vripple_ESR can be neglected for ceramic capacitors, but must be considered if tantalum or
electrolytic capacitors are used.
The minimum ceramic output capacitance needed to meet a load transient requirement can be estimated by the
equation below:
(9)
Where
ΔITRAN is the transient load current step
ΔVTRAN is the allowed voltage dip for the load current step
fLOOP-BW is the control loop bandwidth (i.e., the frequency where the control loop gain crosses zero).
Care must be taken when evaluating a ceramic capacitor’s derating under dc bias, aging and AC signal. For
example, larger form factor capacitors (in 1206 size) have their self resonant frequencies in the range of the
switching frequency. So the effective capacitance is significantly lower. The DC bias can also significantly reduce
capacitance. Ceramic capacitors can loss as much as 50% of its capacitance at its rated voltage. Therefore, one
must add margin on the voltage rating to ensure adequate capacitance at the required output voltage.
For a typical boost converter implementation, at least 4.7μF of ceramic input and output capacitance is
recommended. Additional input and output capacitance may be required to meet ripple and/or transient
requirements.
The popular vendors for high value ceramic capacitors are:
TDK (http://www.component.tdk.com/components.php)
Murata (http://www.murata.com/cap/index.html)
COMPENSATING THE SMALL SIGNAL CONTROL LOOP
All continuous mode boost converters have a right half plane zero (ƒRHPZ) due to the inductor being removed
from the output during charging. In a traditional voltage mode controlled boost converter, the inductor and output
capacitor form a small signal double pole. For a negative feedback system to be stable, the fed back signal must
have a gain less than 1 before having 180 degrees of phase shift. With its double pole and RHPZ all providing
phase shift, voltage mode boost converters are a challenge to compensate. In a converter with current mode
control, there are essentially two loops, an inner current feedback loop created by the inductor current
information sensed across RSENSE (40mΩ) and the output voltage feedback loop. The inner current loop allows
the switch, inductor and modulator to be lumped together into a small signal variable current source controlled by
the error amplifier, as shown in Figure 15.
Figure 15. Small Signal Model of a Current Mode Boost in CCM
The new power stage, including the slope compensation, small signal model becomes:
14 Submit Documentation Feedback Copyright © 2008–2012, Texas Instruments Incorporated
Product Folder Link(s): TPS61175
æ öæ ö
+ -
ç ÷ç ÷
´ p´ ¦ ´ p´ ¦
´ - è ø
è ø
= ´ ´
´+´ p´ ¦
ESR RHPZ
OUT
PS
SENSE
P
s s
1 1
2 2
R (1 D)
G (s) He(s)
s
2 R 12
P
O
2
2 R C2
¦ = p ´ ´
¦ » p ´ ´
ESR
ESR
1
2 R C2
2
O IN
RHPZ
OU T
R V
=
2 L V
æ ö
¦ ´ ç ÷
p ´ è ø
2
2
S W SW
1
He(s)
Se
s 1 (1 D) 0.5
Sn s
1
( )
=é ù
æ ö
´ + ´ - -
ê úç ÷
è ø
ë û
+ +
¦p ´ ¦
-
´
OUT D IN
SENSE
V + V V
Sn = R
L
( )
m
+
´ - ´
0.32 V / R4 0.5 A
Se = 16 1 D 6pF 6 pF
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
(10)
Where
(11)
(12)
(13)
And
(14)
He(s) models the inductor current sampling effect as well as the slope compensation effect on the small signal
response. Note that if Sn > Se, e.g., when L is smaller than recommended, the converter operates as a voltage
mode converter and the above model no longer holds.
The slope compensation in TPS61175 is shown as follow
(15)
Where R4 is the frequency setting resistor (16)
Figure 16 shows a bode plot of a typical CCM boost converter power stage
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Link(s): TPS61175
f − Frequency − kHz
Gain − dB
Phase – °
–180
–120
fP
–60
0
60
120
180
Phase
Gain
+´ p´ ¦
= ´ ´ ´
+æ ö æ ö
+ ´ +
ç ÷ ç ÷
´ p´ ¦ ´ p´ ¦
è ø è ø
Z
EA EA EA
P1 P2
s
1
2
R2
H G R
R2 R1 s s
1 1
2 2
¦ = p ´ ´
P1
EA
1
2 R C 4
¦ = p ´ ´
P2
1(optional)
2 R3 C5
¦p ´ ´
Z
1
=
2 R3 C4
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
Figure 16. Bode Plot of Power Stage Gain and Phase
The TPS61175 COMP pin is the output of the internal trans-conductance amplifier. Equation 17 shows the
equation for feedback resistor network and the error amplifier.
(17)
where GEA and REA are the amplifier’s trans-conductance and output resistance located in the ELECTRICAL
CHARACTERISTICS table.
(18)
C5 is optional and can be modeled as 10 pF stray capacitance. (19)
and
(20)
Figure 17 shows a typical bode plot for transfer function H(s).
16 Submit Documentation Feedback Copyright © 2008–2012, Texas Instruments Incorporated
Product Folder Link(s): TPS61175
f − Frequency − kHz
Gain − dB
Phase – °
–180
–90
fZfC
0
90
180
f 2
p
<–f 1
p
Phase
Gain
Kcomp
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
Figure 17. Bode Plot of Feedback Resistors and Compensated Amplifier Gain and Phase
The next step is to choose the loop crossover frequency, fC. The higher in frequency that the loop gain stays
above zero before crossing over, the faster the loop response will be and therefore the lower the output voltage
will droop during a step load. It is generally accepted that the loop gain cross over no higher than the lower of
either 1/5 of the switching frequency, fSW, or 1/3 of the RHPZ frequency, fRHPZ. To approximate a single pole roll-
off up to fP2, select R3 so that the compensation gain, KCOMP, at fCon Figure 17 is the reciprocal of the gain, KPW,
read at frequency fCfrom the Figure 16 bode plot or more simply
KCOMP(fC) = 20 × log(GEA × R3 × R2/(R2+R1)) = 1/KPW(fC)
This makes the total loop gain, T(s) = GPS(s) × HEA(s), zero at the fC. Then, select C4 so that fZ≅fC/10 and
optional fP2> fC*10. Following this method should lead to a loop with a phase margin near 45 degrees. Lowering
R3 while keeping fZ≅fC/10 increases the phase margin and therefore increases the time it takes for the output
voltage to settle following a step load.
In the TPS61175, if the FB pin voltage changes suddenly due to a load step on the output voltage, the error
amplifier increases its transconductance for 8-ms in an effort to speed up the IC’s transient response and reduce
output voltage droop due to the load step. For example, if the FB voltage decreases 10-mV due to load change,
the error amplifier increases its source current through COMP by 5 times; if FB voltage increases 11-mV, the
sink current through COMP is increased to 3.5 times normal value. This feature often results in saw tooth ringing
on the output voltage, shown as Figure 9. Designing the loop for greater than 45 degrees of phase margin and
greater than 10db gain margin minimizes the amplitude of this ringing. This feature is disabled during soft start.
LAYOUT CONSIDERATIONS
As for all switching power supplies, especially those running at high switching frequency and high currents,
layout is an important design step. If layout is not carefully done, the regulator could suffer from instability as well
as noise problems. To maximize efficiency, switch rise and fall times are very fast. To prevent radiation of high
frequency noise (eg. EMI), proper layout of the high frequency switching path is essential. Minimize the length
and area of all traces connected to the SW pin and always use a ground plane under the switching regulator to
minimize interplane coupling. The high current path including the switch, Schottky diode, and output capacitor,
contains nanosecond rise and fall times and should be kept as short as possible. The input capacitor needs not
only to be close to the VIN pin, but also to the GND pin in order to reduce the Iinput supply ripple.
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Link(s): TPS61175
Minimize thearea
ofSWtrace
SW
AGND
EN
Thermal Pad
SW
VIN
SS
SYNC
PGND
COMP
NC
FREQ
FB
PGND
PGND
OUTPUT
CAPACITOR
PGND
VOUT
SCHOTTKEY
Place enough
VIAs around
thermalpadto
enhance thermal
performance
COMPESNATION
NETWORK
FEEDBACK
SW
AGND
INDUCTOR
VIN
INPUT
CAPACITOR
A
D(max)
JA
125 C T
P = Rq
° -
TPS61175
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
www.ti.com
Figure 18. TPS61175 Layout
THERMAL CONSIDERATIONS
The maximum IC junction temperature should be restricted to 125°C under normal operating conditions. This
restriction limits the power dissipation of the TPS61175. Calculate the maximum allowable dissipation, PD(max),
and keep the actual dissipation less than or equal to PD(max). The maximum-power-dissipation limit is
determined using the following equation:
(21)
where, TAis the maximum ambient temperature for the application. RθJA is the thermal resistance junction-to-
ambient given in Power Dissipation Table.
The TPS61175 comes in a thermally enhanced TSSOP package. This package includes a thermal pad that
improves the thermal capabilities of the package. The RθJA of the TSSOP package greatly depends on the PCB
layout and thermal pad connection. The thermal pad must be soldered to the analog ground on the PCB. Using
thermal vias underneath the thermal pad.
18 Submit Documentation Feedback Copyright © 2008–2012, Texas Instruments Incorporated
Product Folder Link(s): TPS61175
TPS61175
www.ti.com
SLVS892B –DECEMBER 2008–REVISED FEBRUARY 2012
REVISION HISTORY
Changes from Original (December 2008) to Revision A Page
• Changed the Ordering Information table - Part Number From: TPS61175 To: TPS61175PWP; Removed the
Package Marking column ..................................................................................................................................................... 1
Changes from Revision A (October 2010) to Revision B Page
• Replaced the Dissipation Ratings Table with the Thermal Information Table ...................................................................... 2
Copyright © 2008–2012, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Link(s): TPS61175
PACKAGE OPTION ADDENDUM
www.ti.com 16-Feb-2012
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status (1) Package Type Package
Drawing Pins Package Qty Eco Plan (2) Lead/
Ball Finish MSL Peak Temp (3) Samples
(Requires Login)
TPS61175PWP ACTIVE HTSSOP PWP 14 90 Green (RoHS
& no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR
TPS61175PWPR ACTIVE HTSSOP PWP 14 2000 Green (RoHS
& no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
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