LT3790 60V Synchronous 4-Switch Buck-Boost Controller FEATURES DESCRIPTION 4-Switch Single Inductor Architecture Allows VIN Above, Below or Equal to VOUT nn Synchronous Switching: Up to 98.5% Efficiency nn Wide V Range: 4.7V to 60V IN nn 2% Output Voltage Accuracy: 1.2V V OUT < 60V nn 6% Output Current Accuracy: 0V V OUT < 60V nn Input and Output Current Regulation with Current Monitor Outputs nn No Top FET Refresh in Buck or Boost nn V OUT Disconnected from VIN During Shutdown nn C/10 Charge Termination and Output Shorted Flags nn Capable of 100W or Greater per IC nn Easy Parallel Capability to Extend Output Power nn 38-Lead TSSOP with Exposed Pad The LT(R)3790 is a synchronous 4-switch buck-boost voltage/current regulator controller. The LT3790 can regulate output voltage, output current, or input current with input voltages above, below, or equal to the output voltage. The constant-frequency, current mode architecture allows its frequency to be adjusted or synchronized from 200kHz to 700kHz. No top FET refresh switching cycle is needed in buck or boost operation. With 60V input, 60V output capability and seamless transitions between operating regions, the LT3790 is ideal for voltage regulator, battery/super-capacitor charger applications in automotive, industrial, telecom, and even battery-powered systems. nn The LT3790 provides input current monitor, output current monitor, and various status flags, such as C/10 charge termination and shorted output flag. APPLICATIONS L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Automotive, Telecom, Industrial Systems nn High Power Battery-Powered System nn TYPICAL APPLICATION 120W (24V 5A) Buck-Boost Voltage Regulator 0.003 VIN 1F 51 IVINP 0.1F TG1 0.1F EN/UVLO SWI OVLO 56.2k BG1 INTVCC 27.4k 100k LT3790 200k 47F 80V 220F 35V 0.009 4.7F 50V x2 3.83k SNSN PGND RT 3.3k 33nF 33nF SGND 95 90 85 80 75 VIN = 12V VIN = 24V VIN = 54V 65 SW2 CTRL SS SYNC V C 100 70 BG2 TG2 ISP ISN FB Efficiency vs Load Current VOUT 24V 5A 73.2k 10H 0.004 PWM 100k + SNSP SHORT C/10 CCM IVINMON ISMON CLKOUT PWMOUT VREF 0.1F + BST1 499k 4.7F 100V 4.7F BST2 IVINN 470nF 499k INTVCC EFFICIENCY (%) VIN 12V TO 58V 60 0 1 2 3 LOAD CURRENT (A) 4 5 3790 TA01b 3790 TA01a 147k 200kHz 3790fa For more information www.linear.com/LT3790 1 LT3790 ABSOLUTE MAXIMUM RATINGS (Note 1) Supply Voltages Input Supply (VIN)......................................................60V SW1, SW2...................................................... -5V to 60V C/10, SHORT..............................................................15V EN/UVLO, IVINP, IVINN, ISP, ISN...............................60V INTVCC, (BST1-SW1), (BST2-SW2)..............................6V CCM, SYNC, RT, CTRL, OVLO, PWM...........................6V IVINMON, ISMON, FB, SS, VC, VREF............................6V IVINP-IVINN, ISP-ISN, SNSP-SNSN........................0.5V SNSP, SNSN............................................................0.3V Operating Junction Temperature (Notes 2, 3) LT3790E/LT3790I............................... -40C to 125C LT3790H............................................. -40C to 150C LT3790MP.......................................... -55C to 150C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec).................... 300C PIN CONFIGURATION TOP VIEW CTRL 1 38 OVLO SS 2 37 FB PWM 3 36 VC C/10 4 35 RT SHORT 5 34 SYNC VREF 6 33 CLKOUT ISMON 7 32 CCM IVINMON 8 31 PWMOUT EN/UVLO 9 IVINP 10 30 SGND 39 SGND 29 TEST1 IVINN 11 28 SNSN VIN 12 27 SNSP INTVCC 13 26 ISN TG1 14 25 ISP BST1 15 24 TG2 SW1 16 23 NC PGND 17 22 BST2 BG1 18 21 SW2 BG2 19 20 PGND FE PACKAGE 38-LEAD PLASTIC TSSOP JA = 28C/W EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3790EFE#PBF LT3790EFE#TRPBF LT3790FE 38-Lead Plastic TSSOP -40C to 125C LT3790IFE#PBF LT3790IFE#TRPBF LT3790FE 38-Lead Plastic TSSOP -40C to 125C LT3790HFE#PBF LT3790HFE#TRPBF LT3790FE 38-Lead Plastic TSSOP -40C to 150C LT3790MPFE#PBF LT3790MPFE#TRPBF LT3790FE 38-Lead Plastic TSSOP -55C to 150C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS Input VIN Operating Voltage 4.7 60 V VIN Shutdown IQ VEN/UVLO = 0V 0.1 1 A VIN Operating IQ (Not Switching) FB = 1.3V, RT = 59.0k 3.0 4 mA 2 3790fa For more information www.linear.com/LT3790 LT3790 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX 1.16 1.2 1.24 UNITS Logic Inputs EN/UVLO Falling Threshold l EN/UVLO Rising Hysteresis 15 EN/UVLO Input Low Voltage IVIN Drops Below 1A EN/UVLO Pin Bias Current Low VEN/UVLO = 1V EN/UVLO Pin Bias Current High VEN/UVLO = 1.6V 2 CCM Threshold Voltage 0.3 V 3 4 A 10 100 nA 1.5 V 0.3 CTRL Input Bias Current VCTRL = 1V CTRL Latch-Off Threshold Rising 45 CTRL Latch-Off Hysteresis 20 50 nA 50 55 mV 13 OVLO Rising Shutdown Voltage l 2.85 OVLO Falling Hysteresis V mV 3 mV 3.15 75 V mV Regulation VREF Voltage VREF Line Regulation 4.7V < VIN < 60V V(ISP-ISN) Threshold VCTRL = 2V, VISP = 12V/0.1V l 1.96 2.00 2.04 V l 58 56 0.002 0.04 %/V 60 60 62 64 mV mV l 48 46 50 50 52 54 mV mV l 28 26 30 30 32 34 mV mV l 1 0.2 4.2 4.2 7.4 8.2 mV mV VCTRL = 1V, VISP = 12V/0.1V VCTRL = 600mV, VISP = 12V/0.1V VCTRL = 100mV, VISP = 12V/0.1V ISP Bias Current ISN Bias Current VISP = 12V, VISN = 11.9V VISP = 12V, VISN = 11.9V 110 A 20 Output Current Sense Common Mode Range 0 Output Current Sense Amplifier gm A 60 1650 V S ISMON Monitor Voltage V(ISP-ISN) = 60mV l 1.14 Input Current Sense Threshold V(IVINP-IVINN) 3V VIVINP 60V l 46.5 IVINP Bias Current VIVINP = VIVINN = 12V 90 A IVINN Bias Current VIVINP = VIVINN = 12V 20 A Input Current Sense Common Mode Range 50 54 60 2.12 V(IVINP-IVINN) = 50mV 1 1.04 V l 1.2 1.2 1.206 1.220 V V 0.002 0.025 %/V 565 VC Standby Input Bias Current PWM = 0V VSENSE(MAX) (VSNSP-SNSN) Boost Buck V mS 0.96 FB Amplifier gm FB in Regulation mV 1.194 1.176 4.7V < VIN < 60V FB Pin Input Bias Current V l FB Regulation Voltage FB Line Regulation 1.26 3 Input Current Sense Amplifier gm IVINMON Monitor Voltage 1.2 200 20 nA 51 -47.5 60 -39 mV mV -20 l l 42 -56 S 100 nA Fault SS Pull-Up Current VSS = 0V SS Discharge Current 14 A 1.4 A 3790fa For more information www.linear.com/LT3790 3 LT3790 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS C/10 Falling Threshold (V(ISP-ISN)) VFB = 1.2V MIN TYP MAX UNITS 1 5 9 mV 380 400 450 mV C/10 Pin Output Impedance 1.1 2.0 k SHORT Pin Output Impedance 1.1 2.0 k SS Latch-Off Threshold 1.75 V SS Reset Threshold 0.2 V SHORT Falling Threshold (VFB) Oscillator Switching Frequency RT = 147k RT = 59.0k RT = 29.1k 190 380 665 SYNC Frequency 200 400 700 200 SYNC Pin Resistance to GND 210 420 735 kHz kHz kHz 700 kHz 90 SYNC Threshold Voltage 0.3 k 1.5 V Internal VCC Regulator INTVCC Regulation Voltage Dropout (VIN - INTVCC) 4.8 IINTVCC = -10mA, VIN = 5V INTVCC Undervoltage Lockout INTVCC Current Limit 3.1 VINTVCC = 4V 5 5.2 V 240 350 mV 3.5 3.9 V 67 mA PWM PWM Threshold Voltage 0.3 1.5 V PWM Pin Resistance to GND 90 k PWMOUT Pull-Up Resistance 10 20 PWMOUT Pull-Down Resistance 5 10 NMOS Drivers TG1, TG2 Gate Driver On-Resistance Gate Pull-Up Gate Pull-Down VBST - VSW = 5V BG1, BG2 Gate Driver On-Resistance Gate Pull-Up Gate Pull-Down VINTVCC = 5V TG Off to BG On Delay CL = 3300pF 2.6 1.7 3 1.2 60 ns BG Off to TG On Delay CL = 3300pF 60 TG1, TG2, tOFF(MIN) RT = 59.0k 240 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3790E is guaranteed to meet performance from 0C to 125C junction temperature. Specification over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3790I is guaranteed to meet performance specifications over the -40C to 125C operating junction temperature range. The LT3790H is guaranteed to meet performance specifications over the -40C to 150C 4 ns 320 ns operating junction temperature range. The LT3790MP is guaranteed to meet performance specifications over the -55C to 150C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125C. Note 3: The LT3790 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature when overtemperature protection is active. Continuous operation above the specified absolute maximum operating junction temperature may impair device reliability. 3790fa For more information www.linear.com/LT3790 LT3790 TYPICAL PERFORMANCE CHARACTERISTICS INTVCC Dropout Voltage vs Current, Temperature VIN-VINTVCC (V) 90 5.20 TA = 150C TA = 25C TA = -50C 2.0 INTVCC Current Limit vs Temperature INTVCC Voltage vs Temperature 80 5.15 INTVCC CURRENT LIMIT (mA) 2.5 TA = 25C, unless otherwise noted. 5.10 INTVCC (V) 1.5 1.0 5.05 VIN = 60V 5.00 VIN = 12V 4.95 4.90 0.5 4.85 0 20 10 0 30 LDO CURRENT (mA) 0 3790 G01 50 40 30 20 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) VREF Voltage vs Temperature VREF Load Regulation 5.75 2.03 2.15 5.50 2.02 2.10 5.25 2.01 2.05 2.00 4.75 1.99 4.50 1.98 4.25 0 10 20 30 40 50 60 0 1.85 1.80 25 50 75 100 125 150 TEMPERATURE (C) V(ISP-ISN) Threshold vs VCTRL 70 VIN = 12V 68 VCTRL = 2V 66 66 64 64 V(ISP-ISN) (mV) V(ISP-ISN) (mV) 68 62 60 58 56 20 10 RISING FALLING 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 VCTRL (V) 3790 G07 100 150 200 250 300 350 400 IREF (A) V(ISP-ISN) Threshold vs Temperature 70 30 50 3790 G06 V(ISP-ISN) Threshold vs VISP 40 0 3790 G05 50 V(ISP-ISN) (mV) VIN = 60V VIN = 12V VIN = 4.7V 3790 G04 60 0 1.90 1.96 -50 -25 70 2.00 1.95 1.97 ILOAD (mA) 70 VREF (V) 2.20 VREF (V) 2.04 4.00 25 50 75 100 125 150 TEMPERATURE (C) 3790 G03 6.00 5.00 0 3790 G02 INTVCC Load Regulation INTVCC (V) 60 10 4.80 -50 -25 40 70 62 60 58 56 54 54 52 52 50 0 10 20 40 30 VISP (V) 60 50 3790 G08 50 -50 -25 VISP = 60V VISP = 12V VISP = 0V 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G09 3790fa For more information www.linear.com/LT3790 5 LT3790 TYPICAL PERFORMANCE CHARACTERISTICS 72 V(ISP-ISN) Threshold vs VFB TA = 25C, unless otherwise noted. ISMON Voltage vs Temperature 1.24 VIN = 12V V(ISP-ISN) = 60mV 1.23 60 1.2 ISMON Voltage vs V(ISP-ISN) 1.0 36 24 0.8 1.21 VISMON (V) 48 VISMON (V) V(ISP-ISN) (mV) 1.22 1.20 1.19 0.6 0.4 1.18 12 0 1.17 1.19 1.18 1.20 1.21 VFB (V) 1.22 1.16 -50 -25 1.23 0 3790 G10 V(IVINP-IVINN) Threshold vs VIVINP 54 51.5 1.03 51.0 1.02 50.5 1.01 VIVINP = 60V 50 VIVINP = 3V 48 46 42 -50 -25 0 50.0 49.5 48.5 0.97 0 10 20 40 VIVINP (V) 50 50 40 0.500 1.23 0.475 1.22 0.450 VFB (V) 1.20 1.18 0 1.17 1.19 1.20 1.21 VFB (V) 1.22 1.23 3790 G16 1.16 -50 -25 RISING 0.425 0.400 FALLING 0.375 0.350 VIN = 60V VIN = 12V VIN = 4.7V 1.17 1.18 25 50 75 100 125 150 TEMPERATURE (C) 3790 G15 1.24 1.19 10 3790 G12 SHORT Threshold vs Temperature 1.21 20 0 3790 G14 FB VOLTAGE (V) 60 60 VIVINP = 12V V(IVINP-VINN) = 50mV 0.96 -50 -25 60 FB Regulation Voltage vs Temperature V(IVINP-IVINN) Threshold vs VFB V(IVINP-IVINSN) (mV) 30 50 0.99 0.98 3790 G13 30 20 30 40 V(ISP-ISN) (mV) 10 1.00 49.0 48.0 25 50 75 100 125 150 TEMPERATURE (C) VIVINMON (V) 1.04 V(IVINP-IVINN) (mV) 52.0 52 0 IVINMON Voltage vs Temperature 56 44 6 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G11 V(IVINP-IVINN) Threshold vs Temperature V(IVINP-IVINN) (mV) 0.2 1.17 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G17 0.325 0.300 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G18 3790fa For more information www.linear.com/LT3790 LT3790 TYPICAL PERFORMANCE CHARACTERISTICS 16 3.2 14 RISING FALLING 12 3.0 10 2.5 8 2.8 6 2.7 4 2.5 -50 -25 0 1.0 DISCHARGING 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 3790 G20 EN/UVLO THRESHOLD (V) 3 2 1 1.26 1.24 1.22 RISING 1.20 1.18 FALLING 1.16 1.14 1.10 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 TG1, TG2 MINIMUM OFF-TIME (ns) TG1, TG2 MINIMUM ON-TIME (ns) 70 60 50 40 30 20 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G26 3790 G22 RT = 59.0k 400 300 RT = 147k 200 0 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G25 V(BST1-SW1), V(BST2-SW2) UVLO vs Temperature 350 TG1 60 RT = 29.1k 500 TG1, TG2 Minimum Off-Time vs Temperature TG2 80 50 3790 G24 TG1, TG2 Minimum On-Time vs Temperature 90 40 600 25 50 75 100 125 150 TEMPERATURE (C) 3790 G23 100 30 VIN (V) 100 1.12 0 20 700 3.9 300 3.8 fSW = 200kHz 250 V(BST1-SW1), V(BST2-SW2) (V) EN/UVLO PIN CURRENT (A) 7 4 10 800 1.28 5 0 Oscillator Frequency vs Temperature 1.30 VEN/UVLO = 1V 0 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) EN/UVLO Threshold Voltage 6 TA = 150C TA = 25C TA = -50C 0.5 3790 G21 EN/UVLO Pin Current 8 2.0 1.5 2 2.6 3.5 CHARGING SWITCHING FREQUENCY (kHz) 2.9 4.0 IQ (mA) 3.1 3.0 Supply Current vs Input Voltage Soft-Start Current vs Temperature 3.3 ISS (A) OVLO THRESHOLD (V) OVLO Threshold vs Temperature TA = 25C, unless otherwise noted. fSW = 400kHz 200 fSW = 700kHz 150 100 50 0 -50 -25 RISING 3.7 3.6 3.5 3.4 FALLING 3.3 3.2 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G27 3.1 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G28 3790fa For more information www.linear.com/LT3790 7 LT3790 TYPICAL PERFORMANCE CHARACTERISTICS BG1, BG2 Driver On-Resistance vs Temperature TG1, TG2 Driver On-Resistance vs Temperature 4.0 14 3.5 3.0 2.5 2.0 PULL-DOWN 1.5 1.0 12 PWMOUT RESISTANCE () TG1, TG2 RESISTANCE () PULL-UP 3.5 PULL-UP 3.0 2.5 PULL-DOWN 2.0 1.5 1.0 0 -50 -25 0 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 6 0 V(SNSP-SNSN) (mV) BG2 1.0 BG1 0.8 0.6 40 60 DUTY CYCLE (%) 80 60 40 40 20 0 -20 -60 100 0.6 0.8 3790 G32 1.0 1.2 VC (V) 1.4 1.6 40 V(SNSP-SNSN) THRESHOLD (mV) V(SNSP-SNSN) (mV) 40 20 0 -20 -40 -60 1.2 VC (V) 1.4 1.6 0 -20 VC(MAX) -40 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G34 V(SNSP-SNSN) Boost Threshold vs Temperature 60 1.0 20 3790 G33 60 0.8 VC(MIN) -60 -50 -25 1.8 V(SNSP-SNSN) Boost Threshold vs VC -80 0.6 25 50 75 100 125 150 TEMPERATURE (C) 3790 G31 60 -40 0.2 20 0 V(SNSP-SNSN) Buck Threshold vs Temperature 0.4 0 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) V(SNSP-SNSN) THRESHOLD (mV) V(SNSP-SNSN) = 0V 1.2 PULL-DOWN 4 V(SNSP-SNSN) Buck Threshold vs VC 1.4 8 8 3790 G30 VC Voltage vs Duty Cycle 1.6 PULL-UP 2 3790 G29 0 10 0.5 0.5 VC (V) PWMOUT On-Resistance vs Temperature 4.0 4.5 BG1, BG2 RESISTANCE () TA = 25C, unless otherwise noted. 1.8 VC(MAX) 20 0 -20 -40 -60 -80 -50 -25 3790 G35 VC(MIN) 0 25 50 75 100 125 150 TEMPERATURE (C) 3790 G36 3790fa For more information www.linear.com/LT3790 LT3790 PIN FUNCTIONS CTRL (Pin 1): Output Current Sense Threshold Adjustment Pin. Regulating threshold V(ISP-ISN) is 1/20th of VCTRL. CTRL linear range is from 0V to 1.1V. For VCTRL > 1.3V, the current sense threshold is constant at the full-scale value of 60mV. For 1.1V < VCTRL < 1.3V, the dependence of the current sense threshold upon VCTRL transitions from a linear function to a constant value, reaching 98% of full scale by VCTRL = 1.2V. Connect CTRL to VREF for the 60mV default threshold. Force less than 50mV (typical) to stop switching. Do not leave this pin open. EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate 1.2V falling threshold with an externally programmable hysteresis is generated by the external resistor divider and a 3A pull-down current. Above the 1.2V (typical) threshold (but below 6V), EN/UVLO input bias current is sub-A. Below the falling threshold, a 3A pull-down current is enabled so the user can define the hysteresis with the external resistor selection. An undervoltage condition resets soft-start. Tie to 0.3V, or less, to disable the device and reduce VIN quiescent current below 1A. SS (Pin 2): Soft-start reduces the input power sources surge current by gradually increasing the controller's current limit. A minimum value of 22nF is recommended on this pin. A 100k resistor must be placed between SS and VREF for the LT3790. IVINP (Pin 10): Positive Input for the Input Current Limit and Monitor. Input bias current for this pin is typically 90A. PWM (Pin 3): A signal low turns off switches, idles switching and disconnects the VC pin from all external loads. The PWMOUT pin follows the PWM pin. PWM has an internal 90k pull-down resistor. If not used, connect to INTVCC. VIN (Pin 12): Main Input Supply. Bypass this pin to PGND with a capacitor. C/10 (Pin 4): C/10 Charge Termination Pin. An open-drain pull-down on C/10 asserts if V(ISP-ISN) is less than 5mV (typical). To function, the pin requires an external pull-up resistor. SHORT (Pin 5): Output Shorted Pin. An open-drain pulldown on SHORT asserts if FB is less than 400mV (typical) and V(ISP-ISN) is larger than 5mV (typical). To function, the pin requires an external pull-up resistor. VREF (Pin 6): Voltage Reference Output Pin, Typically 2V. This pin drives a resistor divider for the CTRL pin, either for output current adjustment or for temperature limit/ compensation of the output load. Can supply up to 200A of current. ISMON (Pin 7): Monitor pin that produces a voltage that is twenty times the voltage V(ISP-ISN). ISMON will equal 1.2V when V(ISP-ISN) = 60mV. For parallel applications, tie master LT3790 ISMON pin to slave LT3790 CTRL pin. IVINMON (Pin 8): Monitor pin that produces a voltage that is twenty times the voltage V(IVINP-IVINN). IVINMON will equal 1V when V(IVINP-IVINN) = 50mV. IVINN (Pin 11): Negative Input for the Input Current Limit and Monitor. The input bias current for this pin is typically 20A. INTVCC (Pin 13): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Bypass this pin to PGND with a minimum 4.7F ceramic capacitor. TG1 (Pin 14): Top Gate Drive. Drives the top N-channel MOSFET with a voltage equal to INTVCC superimposed on the switch node voltage SW1. BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin swings from a diode voltage below INTVCC up to a diode voltage below VIN + INTVCC. SW1 (Pin 16): Switch Node. SW1 pin swings from a diode voltage drop below ground up to VIN. PGND (Pins 17, 20): Power Ground. Connect these pins closely to the source of the bottom N-channel MOSFET. BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. BG2 (Pin 19): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. SW2 (Pin 21): Switch Node. SW2 pin swings from a diode voltage drop below ground up to VOUT. 3790fa For more information www.linear.com/LT3790 9 LT3790 PIN FUNCTIONS BST2 (Pin 22): Bootstrapped Driver Supply. The BST2 pin swings from a diode voltage below INTVCC up to a diode voltage below VOUT + INTVCC. NC (Pin 23): No Connect Pin. Leave this pin floating. TG2 (Pin 24): Top Gate Drive. Drives the top N-channel MOSFET with a voltage equal to INTVCC superimposed on the switch node voltage SW2. ISP (Pin 25): Connection Point for the Positive Terminal of the Output Current Feedback Resistor. ISN (Pin 26): Connection Point for the Negative Terminal of the Output Current Feedback Resistor. SNSP (Pin 27): The Positive Input to the Current Sense Comparator. The VC pin voltage and controlled offsets between the SNSP and SNSN pins, in conjunction with a resistor, set the current trip threshold. lows the inductor current to flow negative. When the pin voltage is less than 0.3V, the part runs in discontinuous conduction mode and does not allow the inductor current to flow backward. This pin is only meant to block inductor reverse current, and should only be pulled low when the output current is low. This pin must be either connected to INTVCC (pin 13) for continuous conduction mode across all loads, or it must be connected to the C/10 (pin 4) with a pull-up resistor to INTVCC for continuous conduction mode at heavy load and for discontinuous conduction mode at light load. CLKOUT (Pin 33): Clock Output Pin. A 180 out-of-phase clock is provided at the oscillator frequency to allow for paralleling two devices for extending output power capability. SNSN (Pin 28): The Negative Input to the Current Sense Comparator. SYNC (Pin 34): External Synchronization Input Pin. This pin is internally terminated to GND with a 90k resistor. The internal buck clock is synchronized to the rising edge of the SYNC signal while the internal boost clock is 180 phase shifted. TEST1 (Pin 29): This pin is used for testing purposes only and must be connected to SGND for the part to operate properly. RT (Pin 35): Frequency Set Pin. Place a resistor to GND to set the internal frequency. The range of oscillation is 200kHz to 700kHz. SGND (Pin 30, Exposed Pad Pin 39): Signal Ground. All small-signal components and compensation should connect to this ground, which should be connected to PGND at a single point. Solder the exposed pad directly to the ground plane. VC (Pin 36): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0.7V to 1.9V. PWMOUT (Pin 31): Buffered Version of PWM Signal for Driving Output Load Disconnect N-Channel MOSFET. The PWMOUT pin is driven from INTVCC. Use of a MOSFET with a gate cutoff voltage higher than 1V is recommended. CCM (Pin 32): Continuous Conduction Mode Pin. When the pin voltage is higher than 1.5V, the part runs in fixed frequency forced continuous conduction mode and al- 10 FB (Pin 37): Voltage Loop Feedback Pin. FB is intended for constant-voltage regulation. The internal transconductance amplifier with output VC will regulate FB to 1.2V (typical) through the DC/DC converter. OVLO (Pin 38): Overvoltage Input Pin. This pin is used for OVLO, if OVLO > 3V then SS is pulled low, the part stops switching and resets. Do not leave this pin open. 3790fa For more information www.linear.com/LT3790 LT3790 BLOCK DIAGRAM 26 - + A = 20 7 8 9 A1 A = 20 A = 20 12 ISMON_INT 33 INTVCC REGS SHDN_INT A = 24 TSD IVINMON_INT BST1 + A13 A3 EN/UVLO TG1 SW1 - - A4 SHDN_INT SHDN_INT SS_RESET SS LATCH PWM + BG1 PGND A15 BG2 SLOPE_COMP_BUCK SYNC CLKOUT SW2 + SHORT A16 TG2 BST2 - + SNSP + 0.4V - SNSN - FB A5 + C/10 A11 + 0.1V Q R S VREF 0.2V A8 A6 - ISMON_INT 17 19 + SS LATCH IVINMON_INT FB 21 24 22 27 28 37 1.2V A12 SS RESET - - + 14A - + + - ISMON_INT + 3V CTRL 1 A9 - 1.75V INTVCC PWM 18 BOOST LOGIC A10 3 16 INTVCC RT SLOPE_COMP_BOOST 4 14 INTVCC A14 A7 5 15 BUCK LOGIC CCM OSC 34 13 VREF IVINMON 1.2V 35 6 VIN IVINN ISMON 3A 32 11 IVINP + A2 10 ISN ISP - 25 A18 A17 PWMOUT SGND 30, 39 1.4A 31 - 38 VC SS 2 OVLO 36 3790 BD 3790fa For more information www.linear.com/LT3790 11 LT3790 OPERATION The LT3790 is a current mode controller that provides an output voltage above, equal to or below the input voltage. The LTC proprietary topology and control architecture uses a current sensing resistor in buck or boost operation. The sensed inductor current is controlled by the voltage on the VC pin, which is the output of the feedback amplifiers A11 and A12. The VC pin is controlled by three inputs, one input from the output current loop, one input from the input current loop, and the third input from the feedback loop. Whichever feedback input is higher takes precedence, forcing the converter into either a constant-current or a constant-voltage mode. The LT3790 is designed to transition cleanly between the two modes of operation. Current sense amplifier A1 senses the voltage between the IVINP and IVINN pins and provides a pre-gain to amplifier A11. When the voltage between IVINP and IVINN reaches 50mV, the output of A1 provides IVINMON_INT to the inverting input of A11 and the converter is in constant-current mode. If the current sense voltage exceeds 50mV, the output of A1 increases causing the output of A11 to decrease, thus reducing the amount of current delivered to the output. In this manner the current sense voltage is regulated to 50mV. The output current amplifier works similar to the input current amplifier but with a 60mV voltage instead of 50mV. The output current sense level is also adjustable by the CTRL pin. Forcing CTRL to less than 1.2V forces ISMON_INT to the same level as CTRL, thus providing current-level control. The output current amplifier provides rail-to-rail operation. Similarly if the FB pin goes above 1.2V the output of A11 decreases to reduce the current level and regulate the output (constant-voltage mode). The LT3790 provides monitoring pins IVINMON and ISMON that are proportional to the voltage across the input and output current amplifiers respectively. The main control loop is shut down by pulling the EN/ UVLO pin low. When the EN/UVLO pin is higher than 1.2V, an internal 14A current source charges soft-start capacitor CSS at the SS pin. The VC voltage is then clamped a diode voltage higher than the SS voltage while the CSS is slowly charged during start-up. This soft-start clamping prevents abrupt current from being drawn from the input power supply. 12 The top MOSFET drivers are biased from floating bootstrap capacitors C1 and C2, which are normally recharged through an external diode when the top MOSFET is turned off. A unique charge sharing technique eliminates top FET refresh switching cycle in buck or boost operation. Schottky diodes across the synchronous switch M4 and synchronous switch M2 are not required, but they do provide a lower drop during the dead time. The addition of the Schottky diode typically improves peak efficiency by 1% to 2% at 500kHz. Power Switch Control Figure 1 shows a simplified diagram of how the four power switches are connected to the inductor, VIN, VOUT and GND. Figure 2 shows the regions of operation for the LT3790 as a function of duty cycle D. The power switches are properly controlled so the transfer between regions is continuous. When VIN approaches VOUT, the buck-boost region is reached. VOUT VIN TG1 M1 L1 SW1 BG1 TG2 M4 M2 SW2 M3 BG2 RSENSE 3790 F01 Figure 1. Simplified Diagram of the Output Switches DMAX BOOST (BG2) DMIN BOOST DMAX BUCK (TG1) DMIN BUCK BOOST REGION BUCK-BOOST REGION BUCK REGION M1 ON, M2 OFF PWM M3, M4 SWITCHES 4-SWITCH PWM M4 ON, M3 OFF PWM M2, M1 SWITCHES 3790 F02 Figure 2. Operating Regions vs Duty Cycle 3790fa For more information www.linear.com/LT3790 LT3790 OPERATION Buck Region (VIN > VOUT) Switch M4 is always on and switch M3 is always off during this mode. At the start of every cycle, synchronous switch M2 is turned on first. Inductor current is sensed when synchronous switch M2 is turned on. After the sensed inductor current falls below the reference voltage, which is proportional to VC, synchronous switch M2 is turned off and switch M1 is turned on for the remainder of the cycle. Switches M1 and M2 will alternate, behaving like a typical synchronous buck regulator. The duty cycle of switch M1 increases until the maximum duty cycle of the converter in buck operation reaches DMAX(BUCK, TG1), given by: DMAX(BUCK,TG1) = 100% - D(BUCK-BOOST) where D(BUCK-BOOST) is the duty cycle of the buck-boost switch range: D(BUCK-BOOST) = 8% Figure 3 shows typical buck operation waveforms. If VIN approaches VOUT, the buck-boost region is reached. Buck-Boost Region (VIN ~ VOUT) When VIN is close to VOUT, the controller is in buck-boost operation. Figure 4 and Figure 5 show typical waveforms in this operation. Every cycle the controller turns on switches M2 and M4, then M1 and M4 are turned on until 180 later when switches M1 and M3 turn on, and then switches M1 and M4 are turned on for the remainder of the cycle. Boost Region (VIN < VOUT) Switch M1 is always on and synchronous switch M2 is always off in boost operation. Every cycle switch M3 is turned on first. Inductor current is sensed when synchronous switch M3 is turned on. After the sensed inductor current exceeds the reference voltage which is proportional to VC, switch M3 turns off and synchronous switch M4 is turned on for the remainder of the cycle. Switches M3 and M4 alternate, behaving like a typical synchronous boost regulator. The duty cycle of switch M3 decreases until the minimum duty cycle of the converter in boost operation reaches DMIN(BOOST,BG2), given by: DMIN(BOOST,BG2) = D(BUCK-BOOST) where D(BUCK-BOOST) is the duty cycle of the buck-boost switch range: D(BUCK-BOOST) = 8% Figure 6 shows typical boost operation waveforms. If VIN approaches VOUT, the buck-boost region is reached. Low Current Operation The LT3790 is recommended to run in forced continuous conduction mode at heavy load by pulling the CCM pin higher than 1.5V. In this mode the controller behaves as a continuous, PWM current mode synchronous switching regulator. In boost operation, switch M1 is always on, switch M3 and synchronous switch M4 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In buck operation, synchronous switch M4 is always on, switch M1 and synchronous switch M2 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In the forced continuous mode, the output can source or sink current. However, reverse inductor current from the output to the input is not desired for certain applications. For these applications, the CCM pin must be connected to C/10 (pin 4) with a pull-up resistor to INTVCC (see front page Typical Application). Therefore, the CCM pin will be pulled lower than 0.3V for discontinuous conduction mode by the C/10 pin when the output current is low. In this mode, switch M4 turns off when the inductor current flows negative. 3790fa For more information www.linear.com/LT3790 13 LT3790 OPERATION M2 + M4 M2 + M4 M1 + M4 M2 + M4 M1 + M4 M1 + M4 3790 F03 Figure 3. Buck Operation (VIN > VOUT) M1 + M4 M1 + M4 M2 + M4 M1+ M3 M1+ M3 M1 + M4 M1 + M4 M2 + M4 M1+ M3 M2 + M4 M1 + M4 M1 + M4 3790 F04 Figure 4. Buck-Boost Operation (VIN VOUT) M1 + M4 M2 + M4 M1 + M3 M1 + M4 M1 + M4 M2 + M4 M1 + M3 M1 + M4 M1 + M4 M2 + M4 M1 + M3 M1 + M4 3790 F05 Figure 5. Buck-Boost Operation (VIN VOUT) M1 + M3 M1 + M4 M1 + M3 M1 + M4 M1 + M3 M1 + M4 3790 F06 Figure 6. Boost Operation (VIN < VOUT) 14 3790fa For more information www.linear.com/LT3790 LT3790 APPLICATIONS INFORMATION The Typical Application on the front page is a basic LT3790 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs are selected. Finally, CIN and COUT are selected. This circuit can operate up to an input voltage of 60V. 200 180 IL/ISENSE(MAX) (%) 160 Programming The Switching Frequency The RT frequency adjust pin allows the user to program the switching frequency from 200kHz to 700kHz to optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. For an appropriate RT resistor value see Table 1. An external resistor from the RT pin to GND is required; do not leave this pin open. Table 1. Switching Frequency vs RT Value fOSC (kHz) RT (k) 200 147 300 84.5 400 59.0 500 45.3 600 35.7 700 29.4 Frequency Synchronization The LT3790 switching frequency can be synchronized to an external clock using the SYNC pin. Driving SYNC with a 50% duty cycle waveform is always a good choice, otherwise maintain the duty cycle between 10% and 90%. The falling edge of CLKOUT corresponds to the rising edge of SYNC thus allowing 2-phase paralleling converters. The rising edge of CLKOUT turns on switch M3 and the falling edge of CLKOUT turns on switch M2. Inductor Selection The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. The inductor 140 BOOST IL/ ISENSE(MAX) LIMIT 120 100 80 BUCK IL/ ISENSE(MAX) LIMIT 60 40 20 0 50 55 60 65 70 75 80 85 90 95 100 BG1, BG2 DUTY CYCLE (%) 3790 F07 Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle value has a direct effect on ripple current. The maximum inductor current ripple IL can be seen in Figure 7. This is the maximum ripple that will prevent subharmonic oscillation and also regulate with zero load. The ripple should be less than this to allow proper operation over all load currents. For a given ripple the inductance terms in continuous mode are as follows: LBUCK > ( ) VOUT * VIN(MAX ) - VOUT *100 f *IOUT(MAX ) * %Ripple * VIN(MAX ) LBOOST > ( ) VIN(MIN)2 * VOUT - VIN(MIN) *100 f *IOUT(MAX ) * %Ripple * VOUT 2 where: f is operating frequency % ripple is allowable inductor current ripple VIN(MIN) is minimum input voltage VIN(MAX) is maximum input voltage VOUT is output voltage IOUT(MAX) is maximum output load current For high efficiency, choose an inductor with low core loss. Also, the inductor should have low DC resistance to reduce the I2R losses, and must be able to handle the peak inductor current without saturating. To minimize radiated noise, use a shielded inductor. 3790fa For more information www.linear.com/LT3790 15 LT3790 APPLICATIONS INFORMATION RSENSE Selection and Maximum Output Current RSENSE is chosen based on the required output current. The current comparator threshold sets the peak of the inductor current in boost operation and the maximum inductor valley current in buck operation. In boost operation, the maximum average load current at VIN(MIN) is: 51mV I VIN(MIN) IOUT(MAX _ BOOST) = - L * R SENSE 2 VOUT where IL is peak-to-peak inductor ripple current. In buck operation, the maximum average load current is: 47.5mV I IOUT(MAX _ BUCK ) = + L R 2 SENSE 2 * 51mV* VIN(MIN) 2 *IOUT * VOUT + IL(BOOST) * VIN(MIN) The maximum current sensing RSENSE value for the buck operation is: 2 * 47.5mV RSENSE(MAX ) = 2 *IOUT - IL(BUCK ) The final RSENSE value should be lower than the calculated RSENSE(MAX) in both the boost and buck operation. A 20% to 30% margin is usually recommended. CIN and COUT Selection In boost operation, input current is continuous. In buck operation, input current is discontinuous. In buck operation, the selection of input capacitor, CIN, is driven by the need to filter the input square wave current. Use a low ESR capacitor sized to handle the maximum RMS current. For buck operation, the input RMS current is given by: IRMS = IOUT 2 *D+ 16 In boost operation, the discontinuous current shifts from the input to the output, so COUT must be capable of reducing the output voltage ripple. The effects of ESR (equivalent series resistance) and the bulk capacitance must be considered when choosing the right capacitor for a given output ripple voltage. The steady ripple due to charging and discharging the bulk capacitance is given by: VRIPPLE (BOOST _ CAP ) = The maximum current sensing RSENSE value for the boost operation is: RSENSE(MAX ) = The formula has a maximum at VIN = 2VOUT. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. ( IOUT * VOUT - VIN(MIN) ) COUT * VOUT * f IL VRIPPLE (BUCK _ CAP ) 8 * f *C OUT where COUT is the output filter capacitor. The steady ripple due to the voltage drop across the ESR is given by: VBOOST(ESR) = IOUT * ESR VBUCK(ESR) = IOUT * ESR Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Output capacitors are also used for stability for the LT3790. A good starting point for output capacitors is seen in the Typical Applications circuits. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and are recommended for applications less than 100W. Capacitors available with low ESR and high ripple current ratings, such as OS-CON and POSCAP may be needed for applications greater than 100W. IL 2 *D 12 3790fa For more information www.linear.com/LT3790 LT3790 APPLICATIONS INFORMATION Programming VIN UVLO and OVLO The falling UVLO value can be accurately set by the resistor divider R1 and R2. A small 3A pull-down current is active when the EN/UVLO is below the threshold. The purpose of this current is to allow the user to program the rising hysteresis. The following equations should be used to determine the resistor values: VIN(UVLO - ) = 1.2 * R1+R2 R2 The rising OVLO value can be accurately set by the resistor divider R3 and R4. The following equations should be used to determine the resistor values: R3+R4 R4 R3+R4 VIN(OVLO - ) = 2.925 * R4 VIN(OVLO + ) = 3 * LT3790 R1 R3 R2 R4 OVLO EN/UVLO 3790 F08 Figure 8. Resistor Connection to Set VIN UVLO and OVLO Thresholds Programming Output Current The output current is programmed by placing an appropriate value current sense resistor, ROUT, in series with the output load. The voltage drop across ROUT is (Kelvin) sensed by the ISP and ISN pins. The CTRL pin should be tied to a voltage higher than 1.2V to get the full-scale 60mV (typical) threshold across the sense resistor. The CTRL pin can also be used to adjust the output current, although relative accuracy decreases with the decreasing sense threshold. When the CTRL pin voltage is less than 1.1V, the output current is: VCTRL (V) V(ISP-ISN) (mV) 1.1 54.6 1.15 57 1.2 58.8 1.25 59.7 1.3 60 When VCTRL is higher than 1.3V, the output current is regulated to: IOUT = VIN VCTRL ROUT * 20 Table 2. V(ISP-ISN) Threshold vs CTRL R1+R2 R2 VIN(UVLO + ) = 3A *R1+1.215 * IOUT = When the CTRL pin voltage is between 1.1V and 1.3V the output current varies with VCTRL, but departs from the equation above by an increasing amount as VCTRL voltage increases. Ultimately, when VCTRL > 1.3V the output current no longer varies. The typical V(ISP-ISN) threshold vs VCTRL is listed in Table 2. 60mV ROUT The CTRL pin should not be left open (tie to VREF if not used). The CTRL pin can also be used in conjunction with a thermistor to provide overtemperature protection for the output load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high output load current, low switching frequency and/ or a smaller value output filter capacitor. Some level of ripple signal is acceptable: the compensation capacitor on the VC pin filters the signal so the average difference between ISP and ISN is regulated to the user-programmed value. Ripple voltage amplitude (peak-to-peak) in excess of 20mV should not cause mis-operation, but may lead to noticeable offset between the average value and the user-programmed value. ISMON The ISMON pin provides a linear indication of the current flowing through the output. The equation for VISMON is V(ISP-ISN) * 20. This pin is suitable for driving an ADC input, however, the output impedance of this pin is 12.5k so care must be taken not to load this pin. 3790fa For more information www.linear.com/LT3790 17 LT3790 APPLICATIONS INFORMATION Programming Input Current Limit Dimming Control The LT3790 has a standalone current sense amplifier. It can be used to limit the input current. The input current limit is calculated by the following equation: There are two methods to control the current source for dimming using the LT3790. One method uses the CTRL pin to adjust the current regulated in the output. A second method uses the PWM pin to modulate the current source between zero and full current to achieve a precisely programmed average current. To make PWM dimming more accurate, the switch demand current is stored on the VC node during the quiescent phase when PWM is low. This feature minimizes recovery time when the PWM signal goes high. To further improve the recovery time a disconnect switch may be used in the output current path to prevent the ISP node from discharging during the PWM signal low phase. The minimum PWM on- or off-time is affected by choice of operating frequency and external component selection. The best overall combination of PWM and analog dimming capabilities is available if the minimum PWM pulse is at least six switching cycles and the PWM pulse is synchronized to the SYNC signal. IIN = 50mV RIN For loop stability a lowpass RC filter is needed. For most applications, a 50 resistor and 470nF capacitor is sufficient. Table 3 RIN (m) 20 15 12 10 6 5 4 3 2 ILIMIT (A) 2.5 3.3 4.2 5.0 8.3 10.0 12.5 16.7 25 IVINMON The IVINMON pin provides a linear indication of the current flowing through the input. The equation for VIVINMON is V(IVINP-IVINN) * 20. This pin is suitable for driving an ADC input, however, the output impedance of this pin is 12.5k so care must be taken not to load this pin. Programming Output Voltage (Constant Voltage Regulation) For a voltage regulator, the output voltage can be set by selecting the values of R5 and R6 (see Figure 9) according to the following equation: R5+R6 VOUT = 1.2 * R6 VOUT R5 LT3790 SHORT Pin The LT3790 provides an open-drain status pin, SHORT, which pulls low when the FB pin is below 400mV and V(ISP-ISN) is above 5mV. The only time the FB pin will be below 400mV is during start-up or if the output is shorted. During start-up the LT3790 ignores the voltage on the FB pin until the soft-start capacitor reaches 1.75V. To prevent false tripping after startup, a large enough soft-start capacitor must be used to allow the output to get up to approximately 40% to 50% of the final value. C/10 Pin The LT3790 provides an open-drain status pin, C/10, which pulls low when the voltage across V(ISP-ISN) is less than 5mV. For battery charger applications with output current sense and limit, the C/10 provides a C/10 charge termination flag. FB R6 3790 F09 Figure 9. Resistor Connection for Constant Output Voltage Regulation 18 3790fa For more information www.linear.com/LT3790 LT3790 APPLICATIONS INFORMATION Soft-Start Soft-start reduces the input power sources' surge currents by gradually increasing the controller's current limit (proportional to an internally buffered clamped equivalent of VC). The soft-start interval is set by the soft-start capacitor selection according to the following equation tSS = 1.2V *C 14A SS A 100k resistor must be placed between SS and VREF for the LT3790. This 100k resistor also contributes the extra SS charge current. Make sure CSS is large enough when there is loading during start-up. Loop Compensation The LT3790 uses an internal transconductance error amplifier whose VC output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability. In order to select the power MOSFETs, the power dissipated by the device must be known. For switch M1, the maximum power dissipation happens in boost operation, when it remains on all the time. Its maximum power dissipation at maximum output current is given by: 2 I *V PM1(BOOST) = OUT OUT * T *RDS(ON) VIN where T is a normalization factor (unity at 25C) accounting for the significant variation in on-resistance with temperature, typically 0.4%/C as shown in Figure 10. For a maximum junction temperature of 125C, using a value of T = 1.5 is reasonable. Switch M2 operates in buck operation as the synchronous rectifier. Its power dissipation at maximum output current is given by: PM2(BUCK ) = VIN - VOUT *IOUT 2 * T *RDS(ON) VIN The inductor and output capacitor are chosen based on performance, size and cost. The compensation resistor and capacitor at VC are set to optimize control loop response and stability. For typical applications, a 22nF or higher compensation capacitor at VC is needed, and a series resistor should always be used to increase the slew rate on the VC pin to maintain tighter regulation of output current during fast transients on the input supply of the converter. Switch M3 operates in boost operation as the control switch. Its power dissipation at maximum current is given by: Power MOSFET Selections and Efficiency Considerations where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused by reverse-recovery current, is inversely proportional to the gate drive current and has an empirical value of 1.7. The LT3790 requires four external N-channel power MOSFETs, two for the top switches (switch M1 and M4, shown in Figure 1) and two for the bottom switches (switch M2 and M3 shown in Figure 1). Important parameters for the power MOSFETs are the breakdown voltage, VBR(DSS), threshold voltage, VGS(TH), on-resistance, RDS(ON), reverse transfer capacitance, CRSS, and maximum current, IDS(MAX). The drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LT3790 applications. If the input voltage is expected to drop below the 5V, then sub-logic threshold MOSFETs should be considered. PM3(BOOST) = ( VOUT - VIN ) * VOUT *I VIN 2 + k * VOUT 3 * OUT 2 * T *RDS(ON) IOUT *C *f VIN RSS For switch M4, the maximum power dissipation happens in boost operation, when its duty cycle is higher than 50%. Its maximum power dissipation at maximum output current is given by: 2 I OUT * VOUT * * T *RDS(ON) VIN For the same output voltage and current, switch M1 has the highest power dissipation and switch M2 has the lowest power dissipation unless a short occurs at the output. V PM4(BOOST) = IN VOUT 3790fa For more information www.linear.com/LT3790 19 LT3790 APPLICATIONS INFORMATION From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following formula: TJ = TA + P * RTH(JA) The RTH(JA) to be used in the equation normally includes the RTH(JC) for the device plus the thermal resistance from the case to the ambient temperature (RTH(JC)). This value of TJ can then be compared to the original, assumed value used in the iterative calculation process. T NORMALIZED ON-RESISTANCE () 2.0 1.5 1.0 where JA (in C/W) is the package thermal impedance. 50 100 0 JUNCTION TEMPERATURE (C) 150 3790 F10 Figure 10. Normalized RDS(ON) vs Temperature Optional Schottky Diode (D3, D4) Selection The Schottky diodes D3 and D4 shown in the Typical Applications section conduct during the dead time between the conduction of the power MOSFET switches. They are intended to prevent the body diode of synchronous switches M2 and M4 from turning on and storing charge during the dead time. In particular, D4 significantly reduces reverse-recovery current between switch M4 turn-off and switch M3 turn-on, which improves converter efficiency and reduces switch M3 voltage stress. In order for the diode to be effective, the inductance between it and the synchronous switch must be as small as possible, mandating that these components be placed adjacently. INTVCC Regulator An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LT3790. The 20 Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LT3790 to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC also needs to be taken into account for the power dissipation calculations. Power dissipation for the IC in this case is VIN * IINTVCC, and overall efficiency is lowered. The junction temperature can be estimated by using the equations given TJ = TA + (PD * JA) 0.5 0 -50 INTVCC pin regulator can supply a peak current of 67mA and must be bypassed to ground with a minimum of 4.7F ceramic capacitor or low ESR electrolytic capacitor. An additional 0.1F ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is necessary to supply the high transient current required by MOSFET gate drivers. For example, a typical application operating in continuous current operation might draw 24mA from a 24V supply: TJ = 70C + 24mA * 24V * 28C/W = 86C To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. Top Gate (TG) MOSFET Driver Supply (C1, D1, C2, D2) The external bootstrap capacitors C1 and C2 connected to the BST1 and BST2 pins supply the gate drive voltage for the topside MOSFET switches M1 and M4. When the top MOSFET switch M1 turns on, the switch node SW1 rises to VIN and the BST1 pin rises to approximately VIN + INTVCC. When the bottom MOSFET switch M2 turns on, the switch node SW1 drops low and the bootstrap capacitor C1 is charged through D1 from INTVCC. When the bottom MOSFET switch M3 turns on, the switch node SW2 drops low and the bootstrap capacitor C2, is charged through D2 from INTVCC. The bootstrap capacitors C1 and C2 need to store about 100 times the gate charge required by the top MOSFET switch M1 and M4. In most applications a 0.1F to 0.47F, X5R or X7R ceramic capacitor is adequate. 3790fa For more information www.linear.com/LT3790 LT3790 APPLICATIONS INFORMATION Efficiency Considerations PC Board Layout Checklist The power efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in circuits produce losses, four main sources account for most of the losses in LT3790 circuits: The basic PC board layout requires a dedicated ground plane layer. Also, for high current, a multilayer board provides heat sinking for power components. 1. DC I2R losses. These arise from the resistances of the MOSFETs, sensing resistor, inductor and PC board traces and cause the efficiency to drop at high output currents. n 2. Transition loss. This loss arises from the brief amount of time switch M1 or switch M3 spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: n n n n Transition Loss 2.7 * VIN2 * IOUT * CRSS * f where CRSS is the reverse-transfer capacitance. n 3. INTVCC current. This is the sum of the MOSFET driver and control currents. 4. CIN and COUT loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator in buck operation. The output capacitor has the difficult job of filtering the large RMS output current in boost operation. Both CIN and COUT are required to have low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. 5. Other losses. Schottky diode D3 and D4 are responsible for conduction losses during dead time and light load conduction periods. Inductor core loss occurs predominately at light loads. Switch M3 causes reverse recovery current loss in boost operation. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in the input current, then there is no change in efficiency. n n n n n The PGND ground plane layer should not have any traces and it should be as close as possible to the layer with power MOSFETs. Place CIN, switch M1, switch M2 and D1 in one compact area. Place COUT, switch M3, switch M4 and D2 in one compact area. Use immediate vias to connect the components (including the LT3790's SGND and PGND pins) to the ground plane. Use several large vias for each power component. Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. Connect the copper areas to any DC net (VIN or PGND). Separate the signal and power grounds. All small-signal components should return to the SGND pin at one point, which is then tied to the PGND pin close to the sources of switch M2 and switch M3. Place switch M2 and switch M3 as close to the controller as possible, keeping the PGND, BG and SW traces short. Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and TG2 nodes away from sensitive small-signal nodes. The path formed by switch M1, switch M2, D1 and the CIN capacitor should have short leads and PC trace lengths. The path formed by switch M3, switch M4, D2 and the COUT capacitor also should have short leads and PC trace lengths. The output capacitor (-) terminals should be connected as close as possible to the (-) terminals of the input capacitor. Connect the top driver bootstrap capacitor, C1, closely to the BST1 and SW1 pins. Connect the top driver bootstrap capacitor, C2, closely to the BST2 and SW2 pins. For more information www.linear.com/LT3790 3790fa 21 LT3790 APPLICATIONS INFORMATION n n n n Connect the input capacitors, CIN, and output capacitors, COUT, closely to the power MOSFETs. These capacitors carry the MOSFET AC current in boost and buck operation. Route SNSN and SNSP leads together with minimum PC trace spacing. Avoid sense lines pass through noisy areas, such as switch nodes. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. Connect the VC pin compensation network close to the IC, between VC and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output voltage ripple voltage from the compensation loop. Connect the INTVCC bypass capacitor, CVCC, close to the IC, between the INTVCC and the power ground pins. This capacitor carries the MOSFET drivers' current peaks. An additional 0.1F ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 22 Differences Between LT3790 and LT3791-1 The LT3790 is an improved version of the LT3791-1 and is recommended for use in new designs. Some external component values may change, but otherwise, the LT3790 is functionally equivalent to the LT3791-1. The differences between the two products are: 1. The LT3790 has a 60mV (typical) full-scale V(ISP-ISN) current sense voltage, compared to 100mV (typical) for the LT3791-1. This change allows lower power current sense resistors to be used for most applications. 2. The LT3790 CTRL pin linear range is from 0V to 1.1V, and has a turn-off threshold of 50mV(typical), compared to a 200mV to 1.1V linear range and 175mV (typical) turn-off threshold for the LT3791-1. These changes make it easier to parallel two or more LT3790 ICs for higher power levels. 3. The LT3790 C/10 pin pulls low when the V(ISP-ISN) voltage is less than 1/10 full scale, compared to the LT3791-1, where C/10 pulls low when both V(ISP-ISN) is less than 1/10 full scale and VFB is greater than 1.15V(typical). Since the C/10 pin is used to allow DCM mode for some applications, this change ensures that negative current does not occur at light loads for a broader range of applications. 3790fa For more information www.linear.com/LT3790 LT3790 TYPICAL APPLICATIONS 98% Efficient 60W (12V 5A) Voltage Regulator Runs Down to 3V VIN RIN 0.003 VIN 3V TO 55V STARTS UP ABOVE 5.5V R7 51 D5 D6 INTVCC VIN C3 1F BST2 IVINN IVINP TG1 M1 SWI EN/UVLO BG1 OVLO R4 57.6k INTVCC R9 100k R10 200k RFAULT 100k M2 D4 + VO M4 L1, 6.8H COUT2 100F 35V ROUT 0.009 VOUT 12V 5A COUT 10F 25V x3 R5 73.2k M3 D3 R6 8.06k SNSP RSENSE 0.004 LT3790 SNSN SHORT C/10 CCM IVINMON PGND BG2 SW2 TG2 ISP ISN FB ISMON CLKOUT PWMOUT VREF C8 0.1F C2 0.1F C1 0.1F BST1 R3 1M R2 576k CVCC 4.7F D1 D2 C7 470nF R1 866k CIN 4.7F 100V x4 VO PWM CTRL SS SYNC VC RT RC 5.1k CC 22nF CSS 33nF SGND R8 84.5k 300kHz 3790 TA02a D1, D2: NXP BAT46WJ D3: IRF 10BQ060 D4: IRF 10BQ040 D5, D6: DIODES INC. BAT46W L1: WURTH ELEKTRONIK WE-HCI 7443556680 M1, M2: RENASAS RJK0651DPB 60VDS M3, M4: VISHAY SiR424DP 40VDS COUT2: SUNCON 35HVT100M Efficiency vs Load Current Maximum Output Current vs VIN 100 6 MAXIMUM OUTPUT CURRENT (A) 95 EFFICIENCY (%) 90 85 80 75 70 VIN = 3V VIN = 6V VIN = 12V VIN = 28V VIN = 48V 65 60 55 50 0 1 2 3 LOAD CURRENT (A) 4 5 5 4 3 2 1 0 3 3790 TA02b 4 5 6 7 8 9 10 20 30 40 50 60 INPUT VOLTAGE (V) 3790 TA02c 3790fa For more information www.linear.com/LT3790 23 LT3790 TYPICAL APPLICATIONS 98% Efficient 240W (24V 10A) Parallel Voltage Regulators 0.003 VIN 12V TO 58V 499k 470nF 499k IVINP EN/UVLO 51 IVINN VIN INTVCC 27.4k BST1 INTVCC1 TG1 M1 SWI SHORT VREF 0.1F BG1 LT3790 PWM M2 0.1F L1 10H VOUT 1V/DIV 4.7F 50V x2 0.009 VOUT 24V 10A M4 51 M3 + COUT1 220F 35V x2 IL2 5A/DIV IL1 5A/DIV 0.47F SNSP CTRL VIN = 36V IOUT = 5A TO 10A SNSN PGND SS BG2 IVINMON CLKOUT ISMON SYNC VC SW2 TG2 ISP ISN FB SGND RT Startup Waveform 715k 13.7k VOUT 5V/DIV 38.3k VSS 1V/DIV 147k 200kHz 3.3k 33nF IL1 5A/DIV IL2 5A/DIV 0.003 VIN 499k IVINP EN/UVLO 1F IVINN VIN INTVCC 27.4k BST1 TG1 200k 0.1F SHORT VREF BG1 LT3790 M6 0.1F L2 10H 4.7F 50V x2 0.009 Mismatch Current vs Load Current M7 51 + 1000pF 0.4 0.3 0.22F 0.2 0.1 SNSN PGND CTRL IVINMON ISMON CLKOUT SYNC VC 2.2k 22nF RT SW2 TG2 ISP ISN FB SGND 147k 200kHz 0.0 -0.1 BG2 1nF 0.5 COUT2 220F 35V x2 0.004 SS 3790 TA03b VIN = 36V IOUT = 5A M8 SNSP 100k 33nF M5 SWI PWM 4.7F 10V D3 D4 0.1F C/10 C2 47F 80V 1ms/DIV BST2 INTVCC2 SHORT + INTVCC2 CCM OVLO 56.2k 4.7F 100V I (A) 499k 470nF 51 3790 TA03a 1ms/DIV 0.004 100k 33nF 4.7F 10V D1 D2 0.1F C/10 Transient Waveform C1 47F 80V BST2 200k SHORT + INTVCC1 CCM OVLO 56.2k 4.7F 100V 1F 715k -0.2 -0.3 140k -0.4 -0.5 38.3k D1-D4: NXP BAT46WJ L1, L2: COILCRAFT SER2915L-103KL 10H M1, M2, M5, M6: RENESAS RJK0651DPB 60Vds M3, M4, M7, M8: RENESAS RJK0451DPB 40Vds COUT1, COUT2: SUNCON 35HVT220M x2 C1, C2: NIPPON CHEMICON EMZA800ADA470MJAOG 0 2 6 4 LOAD CURRENT (A) 8 10 3790 TA03c 3790 TA03 24 3790fa For more information www.linear.com/LT3790 1F VIN 12V TO 58V 27.4k 56.2k For more information www.linear.com/LT3790 33nF 332k SS CTRL PWM SHORT VREF C/10 OVLO IVINP EN/UVLO OUT2 OUT3 PH MOD SET GND LTC6902 OUT1 VC DIV V+ 33nF 3.3k IVINMON CLKOUT ISMON SYNC1 SYNC 100k INTVCC1 0.1F SHORT 200k INTVCC1 499k 499k 51 1F RT BG2 SNSN PGND SNSP SW2 TG2 ISP ISN FB SGND 147k 200kHz LT3790 BG1 SWI TG1 BST1 BST2 CCM IVINN VIN INTVCC 470nF 0.003 M2 M1 M3 M4 + 0.004 L1 10H 0.1F 0.1F D1 D2 INTVCC1 4.7F 100V 4.7F 10V C1 47F 80V 0.47F 51 4.7F 50V x2 0.009 38.3k 13.7k 715k + SYNC3 SYNC2 COUT1 220F 35V x2 VOUT 24V 15A 0.1F INTVCC2 27.4k 499k 200k 1nF 33nF 100k SHORT 56.2k 499k VIN SLAVE x 2 22nF 1000pF 2.2k CTRL IVINMON ISMON CLKOUT SYNC VC SS PWM SHORT VREF C/10 OVLO IVINP EN/UVLO 51 1F 147k 200kHz SW2 TG2 ISP ISN FB SGND BG2 SNSN PGND SNSP M6 M5 M7 M8 + 0.004 L2 10H 0.1F 0.1F D3 D4 INTVCC2 4.7F 100V 4.7F 10V C2 47F 80V D1-D4: NXP BAT46WJ L1, L2: COILCRAFT SER2915L-103KL 10H M1, M2, M5, M6: RENESAS RJK0651DPB 60Vds M3, M4, M7, M8: RENESAS RJK0451DPB 40Vds COUT1, COUT2: SUNCON 35HVT220M x2 C1, C2: NIPPON CHEMICON EMZA800ADA470MJAOG RT LT3790 BG1 SWI TG1 BST1 BST2 CCM IVINN VIN INTVCC 470nF 0.003 0.22F 51 4.7F 50V x2 0.009 38.3k 140k 715k + 3790 TA04 COUT2 220F 35V x2 LT3790 TYPICAL APPLICATIONS 98% Efficient 360W (24V 15A) Parallel Voltage Regulators 3790fa 25 LT3790 PACKAGE DESCRIPTION Please refer to http://www.linear.com/product/LT3790#packaging for the most recent package drawings. FE Package 38-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1772 Rev C) Exposed Pad Variation AA 4.75 REF 38 9.60 - 9.80* (.378 - .386) 4.75 REF (.187) 20 6.60 0.10 4.50 REF 2.74 REF SEE NOTE 4 6.40 2.74 REF (.252) (.108) BSC 0.315 0.05 1.05 0.10 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 - 4.50* (.169 - .177) 0.50 - 0.75 (.020 - .030) 0.09 - 0.20 (.0035 - .0079) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN MILLIMETERS (INCHES) 3. DRAWING NOT TO SCALE 26 1 0.25 REF 19 1.20 (.047) MAX 0 - 8 0.50 (.0196) BSC 0.17 - 0.27 (.0067 - .0106) TYP 0.05 - 0.15 (.002 - .006) FE38 (AA) TSSOP REV C 0910 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3790fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. For more information www.linear.com/LT3790 LT3790 REVISION HISTORY REV DATE DESCRIPTION A 12/15 Clarified Typical Application schematic Clarified Electrical V(ISP-ISN) Threshold parameters Clarified ISP/ISN Bias Current parameters Clarified Block Diagram Clarified ILED to IOUT in Equations Clarified Loop Compensation paragraph PAGE NUMBER 1 3 3 11 17, 20 10 3790fa For more information www.linear.com/LT3790 27 LT3790 TYPICAL APPLICATION 2.4A Buck-Boost 36V SLA Battery Charger PVIN 9V TO 58V RIN 0.003 1F 51 VIN INTVCC D1 TG1 IVINP INTVCC 200k 30.9k BG1 LT3790 SHORT PWMOUT CHARGE CURRENT CONTROL RBAT 0.025 SNSN PGND IVINMON ISMON CTRL BG2 SW2 TG2 ISP PWM ISN FB C/10 SS CCM RT VREF SYNC VC D1, D2: BAT46WJ 22nF L1: COILCRAFT SER2915L-103K M1-M4: RENESAS RJK0651DPB M5: NXP NX7002AK CIN2: NIPPON CHEMI-CON EMZA630ADA101MJA0G COUT2: SUNCON 50HVT220M M3 RSENSE 0.004 SGND 100k L1 10H M2 COUT 4.7F 50V x2 M4 0.1F CIN2 100F 63V + COUT2 220F 50V SNSP CLKOUT 0.1F M1 SWI OVLO + 0.1F BST1 EN/UVLO 57.6k 4.7F D2 BST2 IVINN 470nF 499k CIN 4.7F 100V x2 2.2k + 1F 2.4A CHARGE 36V SLA BATTERY AGM TYPE 41V FLOAT 44V CHARGE AT 25C 51 1.00M INTVCC 20k 84.5k 300kHz 402k 30.1k M5 22nF 3790 TA05 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS 4.7V VIN 60V, 1.2V VOUT 60V, PWM Dimming, TSSOP-38 LT3791 60V 4-Switch Synchronous Buck-Boost LED Driver LT8705 80V VIN and VOUT Synchronous 4-Switch Buck-Boost 2.8V VIN 80V, 1.3V VOUT 80V, Regulates VOUT, IOUT, VIN, IIN, 5mm x DC/DC Controller 7mm QFN-38, Modified TSSOP Package for High Voltage LTC3789 High Efficiency Synchronous 4-Switch Buck-Boost Controller 4V VIN 38V, 0.8V VOUT 38V, 4mm x 5mm QFN-28, SSOP-28 LTC3780 High Efficiency Synchronous 4-Switch Buck-Boost Controller 4V VIN 36V, 0.8V VOUT 30V, 5mm x 5mm QFN-32, SSOP-24 LT3741/LT3741-1 High Power, Constant Current, Constant Voltage, Step-Down Controller 6V VIN 36V, 4mm x 4mm QFN-20, TSSOP-20 LT3763 6V VIN 60V, 4mm x 4mm QFN-20, TSSOP-20 60V High Current Step-Down LED Driver Controller LT3757/LT3757A Boost, Flyback, SEPIC and Inverting Controller 2.9V VIN 40V, Positive or Negative VOUT, 3mm x 3mm DFN-10, MSOP-10 LT3758 High Input Voltage, Boost, Flyback, SEPIC and Inverting Controller 5.5V VIN 100V, Positive or Negative VOUT, 3mm x 3mm DFN-10, MSOP-10 LT8490 High Voltage, High Current Buck-Boost Battery Charge 6V VIN 80V, 1.3V VBAT 80V, Automatic MPPT, 7mm x 11mm QFN-64 Controller with Maximum Power Point Tracking (MPPT) LT8710 Synchronous SEPIC/Inverting/Boost Controller with Output Current Control 28 Linear Technology Corporation 4.5V VIN 80V, Rail-to-Rail Output Current Monitor and Control, TSSOP-28E Package 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT3790 (408) 432-1900 FAX: (408) 434-0507 www.linear.com/LT3790 3790fa LT 1215 REV A * PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2014