REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
a
ADE7751
*
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700www.analog.com
Fax: 781/326-8703 © Analog Devices, Inc., 2002
Energy Metering IC
with On-Chip Fault Detection
FEATURES
High Accuracy, Surpasses 50 Hz/60 Hz IEC 687/1036
Less than 0.1% Error over a Dynamic Range of 500 to 1
Supplies Average Real Power on the Frequency
Outputs F1 and F2
High-Frequency Output CF Is Intended for Calibration
and Supplies Instantaneous Real Power
Continuous Monitoring of the Phase and Neutral
Current Allows Fault Detection in 2-Wire
Distribution Systems
ADE7751 Uses the Larger of the Two Currents (Phase
or Neutral) to Bill—Even During a Fault Condition
Two Logic Outputs (FAULT and REVP) Can Be Used to
Indicate a Potential Miswiring or Fault Condition
Direct Drive for Electromechanical Counters and
2-Phase Stepper Motors (F1 and F2)
A PGA in the Current Channel Allows the Use of Small
Values of Shunt and Burden Resistance
Proprietary ADCs and DSP Provide High Accuracy over
Large Variations in Environmental Conditions and Time
On-Chip Power Supply Monitoring
On-Chip Creep Protection (No Load Threshold)
On-Chip Reference 2.5 V ⴞ 8% (30 ppm/ⴗC Typical)
with External Overdrive Capability
Single 5 V Supply, Low Power (15 mW Typical)
Low-Cost CMOS Process
FUNCTIONAL BLOCK DIAGRAM
AC/DC
AV
DD
DV
DD
HPF
AGND DGND
PHASE
CORRECTION
4k⍀
...
110101
...
SIGNAL
PROCESSING
BLOCK
PGA
ⴛ1, ⴛ2, ⴛ8, ⴛ16
POWER
SUPPLY MONITOR
ADE7751
...
11011001
...
2.5V
REFERENCE
PGA
ⴛ1, ⴛ2, ⴛ8, ⴛ16
ADC
...
110101
...
B>A
A>B
A<>B
ADC
ADC
⌽
FAULT
MULTIPLIER
LPF
DIGITAL-TO-FREQUENCY
CONVERTER
G0 G1
V1A
V1N
V2P
V2N
V1B
A
B
CLKOUTCLKINREF
IN/OUT
RESET
F1 F2CFREVPSCF S0 S1
*US Patent 5,745,323; 5,760,617; 5,862,069; 5,872,469.
The only analog circuitry used in the ADE7751 is in the ADCs
and reference circuit. All other signal processing (e.g., multipli-
cation and filtering) is carried out in the digital domain. This
approach provides superior stability and accuracy over extremes
in environmental conditions and over time.
The ADE7751 incorporates a novel fault detection scheme that
warns of fault conditions and allows the ADE7751 to continue
accurate billing during a fault event. The ADE7751 does this
by continuously monitoring both the phase and neutral (return)
currents. A fault is indicated when these currents differ by more
than 12.5%. Billing is continued using the larger of the two currents.
The ADE7751 supplies average real power information on the
low-frequency outputs F1 and F2. These logic outputs may be
used to directly drive an electromechanical counter or interface
to an MCU. The CF logic output gives instantaneous real power
information. This output is intended to be used for calibration purposes.
The ADE7751 includes a power supply monitoring circuit on the
AV
DD
supply pin. The ADE7751 will remain in a reset condition
until the supply voltage on AV
DD
reaches 4 V. If the supply falls
below 4 V, the ADE7751 will also be reset and no pulses will be
issued on F1, F2, and CF.
Internal phase matching circuitry ensures that the voltage and
current channels are matched whether the HPF in Channel 1 is
on or off. The ADE7751 also has anticreep protection.
The ADE7751 is available in 24-lead DIP and SSOP packages.
GENERAL DESCRIPTION
The ADE7751 is a high-accuracy, fault-tolerant electrical energy
measurement IC that is intended for use with 2-wire distribution
systems. The part specifications surpass the accuracy require-
ments as quoted in the IEC1036 standard.
REV. 0
–2–
ADE7751–SPECIFICATIONS
1, 2
(AVDD = DVDD = 5 V ⴞ 5%, AGND = DGND = 0 V, On-Chip Reference,
CLKIN = 3.58 MHz, TMIN to TMAX = –40ⴗC to +85ⴗC.)
Parameter Value Unit Test Conditions/Comments
ACCURACY
3
Measurement Error
1
on Channels 1 and 2 One Channel with Full-Scale Signal (±660 mV)
Gain = 1 0.1 % Reading typ Over a Dynamic Range 500 to 1
Gain = 2 0.1 % Reading typ Over a Dynamic Range 500 to 1
Gain = 8 0.1 % Reading typ Over a Dynamic Range 500 to 1
Gain = 16 0.1 % Reading typ Over a Dynamic Range 500 to 1
Phase Error
1
between Channels Line Frequency = 45 Hz to 55 Hz
V1 Phase Lead 37°
(PF = 0.8 Capacitive) ±0.1 Degrees(°) max AC/DC = 0 and AC/DC = 1
V1 Phase Lag 60°
(PF = 0.5 Inductive) ±0.1 Degrees(°) max AC/DC = 0 and AC/DC = 1
AC Power Supply Rejection
1
AC/DC = 1, S0 = S1 = 1, G0 = G1 = 0
Output Frequency Variation (CF) 0.2 % Reading typ V1 = 100 mV rms, V2 = 100 mV rms @ 50 Hz
Ripple on AV
DD
of 200 mV rms @ 100 Hz
DC Power Supply Rejection
1
AC/DC = 1, S0 = S1 = 1, G0 = G1 = 0
Output Frequency Variation (CF) ±0.3 % Reading typ V1 = 100 mV rms, V2 = 100 mV rms,
AV
DD
= DV
DD
= 5 V ± 250 mV
FAULT DETECTION
1, 4
See Fault Detection Section
Fault Detection Threshold
Inactive i/p <> Active i/p 12.5 % typ (V1A or V1B Active)
Input Swap Threshold
Inactive i/p > Active i/p 14 % of Active typ (V1A or V1B Active)
Accuracy Fault Mode Operation
V1A Active, V1B = AGND 0.1 % Reading typ Over a Dynamic Range 500 to 1
V1B Active, V1A = AGND 0.1 % Reading typ Over a Dynamic Range 500 to 1
Fault Detection Delay 3 Second typ
Swap Delay 3 Second typ
ANALOG INPUTS See Analog Inputs Section
Maximum Signal Levels ±1V max V1A, V1B, V1N, V2N, and V2P to AGND
Input Impedance (DC) 390 kΩ min CLKIN = 3.58 MHz
Bandwidth 14 kHz typ CLKIN/256, CLKIN = 3.58 MHz
ADC Offset Error
1
±25 mV max See Terminology and Typical Performance
Characteristics
Gain Error
1
±10 % Ideal typ External 2.5 V Reference, Gain = 1,
V1 = V2 = 660 mV dc
Gain Error Match
1
±0.4 % Ideal typ External 2.5 V Reference
REFERENCE INPUT
REF
IN/OUT
Input Voltage Range 2.7 V max 2.5 V + 8%
2.3 V min 2.5 V – 8%
Input Impedance 3.2 kΩ min
Input Capacitance 10 pF max
ON-CHIP REFERENCE Nominal 2.5 V
Reference Error ±200 mV max
Temperature Coefficient ±30 ppm/°C typ
CLKIN Note All Specifications for CLKIN of 3.58 MHz
Input Clock Frequency 4 MHz max
1MHz min
LOGIC INPUTS
4
SCF, S0, S1, AC/DC,
RESET, G0, and G1
Input High Voltage, V
INH
2.4 V min DV
DD
= 5 V ± 5%
Input Low Voltage, V
INL
0.8 V max DV
DD
= 5 V ± 5%
Input Current, I
IN
±3µA max Typically 10 nA, V
IN
= 0 V to DV
DD
Input Capacitance, C
IN
10 pF max
REV. 0 –3–
ADE7751
Parameter Value Unit Test Conditions/Comments
LOGIC OUTPUTS
4
F1 and F2
Output High Voltage, V
OH
I
SOURCE
= 10 mA
4.5 V min DV
DD
= 5 V
Output Low Voltage, V
OL
I
SINK
= 10 mA
0.5 V max DV
DD
= 5 V
CF, FAULT, and REVP
Output High Voltage, V
OH
I
SOURCE
= 5 mA
4V minDV
DD
= 5 V
Output Low Voltage, V
OL
I
SINK
= 5 mA
0.5 V max DV
DD
= 5 V
POWER SUPPLY For Specified Performance
AV
DD
4.75 V min 5 V – 5%
5.25 V max 5 V + 5%
DV
DD
4.75 V min 5 V – 5%
5.25 V max 5 V + 5%
AI
DD
3mA max Typically 2 mA
DI
DD
2.5 mA max Typically 1.5 mA
NOTES
1
See Terminology section for explanation of specifications.
2
See plots in Typical Performance Characteristics graphs.
3
See Fault Detection section of data sheet for explanation of fault detection functionality.
4
Sample tested during initial release and after any redesign or process change that may affect this parameter.
Specifications subject to change without notice.
TIMING CHARACTERISTICS
1, 2
Parameter Value Unit Test Conditions/Comments
t
13
275 ms F1 and F2 Pulsewidth (Logic Low)
t
2
See Table III sec Output Pulse Period. See Transfer Function section.
t
3
1/2 t
2
sec Time Between F1 Falling Edge and F2 Falling Edge
t
43
90 ms CF Pulsewidth (Logic High)
t
5
See Table IV sec CF Pulse Period. See Transfer Function section.
t
6
CLKIN/4 sec Minimum Time Between F1 and F2 Pulse
NOTES
1
Sample tested during initial release and after any redesign or process change that may affect this parameter.
2
See Figure 1.
3
The pulsewidths of F1, F2, and CF are not fixed for higher output frequencies. See Frequency Outputs F1 and F2 section.
Specifications subject to change without notice.
.
t
2
.t
3
t
4
.t
5
.t
6
t
1
F1
F2
CF
Figure 1. Timing Diagram for Frequency Outputs
(AVDD = DVDD = 5 V ⴞ 5%, AGND = DGND = 0 V, On-Chip Reference, CLKIN = 3.58 MHz,
TMIN to TMAX = –40ⴗC to +85ⴗC.)
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ADE7751
–4–
ABSOLUTE MAXIMUM RATINGS*
(T
A
= 25°C, unless otherwise noted.)
AV
DD
to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
DV
DD
to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
DV
DD
to AV
DD
. . . . . . . . . . . . . . . . . . . . . . –0.3 V to +0.3 V
Analog Input Voltage to AGND
V1A, V1B, V1N, V2P, and V2N . . . . . . . . . . –6 V to +6 V
Reference Input Voltage to AGND . . –0.3 V to AV
DD
+ 0.3 V
Digital Input Voltage to DGND . . . . –0.3 V to DV
DD
+ 0.3 V
Digital Output Voltage to DGND . . . –0.3 V to DV
DD
+ 0.3 V
Operating Temperature Range
Industrial . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
24-Lead Plastic DIP, Power Dissipation . . . . . . . . . . 450 mW
θ
JA
Thermal Impedance . . . . . . . . . . . . . . . . . . . . 105°C/W
Lead Temperature, (Soldering 10 sec) . . . . . . . . . . . 260°C
24-Lead SSOP, Power Dissipation . . . . . . . . . . . . . . 450 mW
θ
JA
Thermal Impedance . . . . . . . . . . . . . . . . . . . . 112°C/W
Lead Temperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . 215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C
*Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute maximum rating condi-
tions for extended periods may affect device reliability.
ORDERING GUIDE
Package
Model Package Description Option
ADE7751AN Plastic DIP N-24
ADE7751ARS Shrink Small Outline Package RS-24
ADE7751ARSRL Shrink Small Outline Package RSRL-24
in Reel
EVAL-ADE7751EB ADE7751 Evaluation Board
ADE7751AAN-REF ADE7751 Reference Design
PCB (See AN-563)
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the ADE7751 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. 0
ADE7751
–5–
PIN CONFIGURATION
TOP VIEW
(Not to Scale)
24
23
22
21
20
19
18
17
16
15
14
13
1
2
3
4
5
6
7
8
9
10
11
12
ADE7751
SCF
AGND
REFIN/OUT
RESET
V2P
DVDD
AC/DC
AVDD
V1A
V2N
V1N
V1B
S1
S0
G1
G0
CLKIN
F1
F2
CF
DGND
CLKOUT
FAULT
REVP
PIN FUNCTION DESCRIPTIONS
Pin No. Mnemonic Description
1DV
DD
Digital Power Supply. This pin provides the supply voltage for the digital circuitry in the ADE7751.
The supply voltage should be maintained at 5 V ± 5% for specified operation. This pin should be
decoupled with a 10 µF capacitor in parallel with a ceramic 100 nF capacitor.
2AC/DC High-Pass Filter Select. This logic input is used to enable the HPF in Channel 1 (the current
channel). A Logic 1 on this pin enables the HPF. The associated phase response of this filter has
been internally compensated over a frequency range of 45 Hz to 1 kHz. The HPF filter should be
enabled in energy metering applications.
3AV
DD
Analog Power Supply. This pin provides the supply voltage for the analog circuitry in the ADE7751.
The supply should be maintained at 5 V ± 5% for specified operation. Every effort should be made
to minimize power supply ripple and noise at this pin by the use of proper decoupling. This pin
should be decoupled to AGND with a 10 µF capacitor in parallel with a ceramic 100 nF capacitor.
4, 5 V1A, V1B Analog Inputs for Channel 1 (Current Channel). These inputs are fully differential voltage inputs
with a maximum signal level of ±660 mV with respect to pin V1N for specified operation. The
maximum signal level at these pins is ±1 V with respect to AGND. Both inputs have internal ESD
protection circuitry and an overvoltage of ±6 V can also be sustained on these inputs without risk of
permanent damage.
6V1N Negative Input Pin for Differential Voltage Inputs V1A and V1B. The maximum signal level at this
pin is ±1 V with respect to AGND. The input has internal ESD protection circuitry and an overvoltage
of ±6 V can also be sustained without risk of permanent damage. This input should be directly con-
nected to the burden resistor and held at a fixed potential, i.e., AGND. See Analog Input section.
7, 8 V2N, V2P Negative and Positive Inputs for Channel 2 (Voltage Channel). These inputs provide a fully differ-
ential input pair. The maximum differential input voltage is ±660 mV for specified operation. The
maximum signal level at these pins is ±1 V with respect to AGND. Both inputs have internal ESD
protection circuitry and an overvoltage of ±6 V can also be sustained on these inputs without risk of
permanent damage.
9RESET Reset Pin for the ADE7751. A logic low on this pin will hold the ADCs and digital circuitry in a
reset condition. Bringing this pin logic low will clear the ADE7751 internal registers.
10 REF
IN/OUT
Provides Access to the On-Chip Voltage Reference. The on-chip reference has a nominal value of
2.5 V ± 8% and a typical temperature coefficient of 30 ppm/°C. An external reference source may also
be connected at this pin. In either case, this pin should be decoupled to AGND with a 1 µF ceramic
capacitor and 100 nF ceramic capacitor.
11 AGND Provides the Ground Reference for the Analog Circuitry in the ADE7751, i.e., ADCs and Refer-
ence. This pin should be tied to the analog ground plane of the PCB. The analog ground plane is
the ground reference for all analog circuitry, e.g., antialiasing filters, current and voltage trans-
ducers, and more. For good noise suppression, the analog ground plane should only be connected to
the digital ground plane at one point. A star ground configuration will help to keep noisy digital
return currents away from the analog circuits.
12 SCF Select Calibration Frequency. This logic input is used to select the frequency on the calibration
output CF. Table IV shows how the calibration frequencies are selected.
REV. 0
ADE7751
–6–
Pin No. Mnemonic Description
13, 14 S1, S0 These logic inputs are used to select one of four possible frequencies for the digital-to-frequency
conversion. This offers the designer greater flexibility when designing the energy meter. See Select-
ing a Frequency for an Energy Meter Application section.
15, 16 G1, G0 These logic inputs are used to select one of four possible gains for the analog inputs V1A and V1B.
The possible gains are 1, 2, 8, and 16. See Analog Inputs section.
17 CLKIN An external clock can be provided at this logic input. Alternatively, a parallel resonant AT crystal can
be connected across CLKIN and CLKOUT to provide a clock source for the ADE7751. The clock
frequency for specified operation is 3.579545 MHz. Crystal load capacitors of between 22 pF
and 33 pF (ceramic) should be used with the gate oscillator circuit.
18 CLKOUT A crystal can be connected across this pin and CLKIN as described above to provide a clock source
for the ADE7751. The CLKOUT pin can drive one CMOS load when an external clock is supplied
at CLKIN or by the gate oscillator circuit.
19 FAULT This logic output will go active high when a fault condition occurs. A fault is defined as a condition
under which the signals on V1A and V1B differ by more than 12.5%. The logic output will be reset
to zero when a fault condition is no longer detected. See Fault Detection section.
20 REVP This logic output will go logic high when negative power is detected, i.e., when the phase angle
between the voltage and current signals is greater that 90°. This output is not latched and will be
reset when positive power is once again detected. The output will go high or low at the same time as
a pulse is issued on CF.
21 DGND This provides the ground reference for the digital circuitry in the ADE7751, i.e., multiplier, filter,
and digital-to-frequency converter. This pin should be tied to the analog ground plane of the PCB.
The digital ground plane is the ground reference for all digital circuitry, e.g., counters (mechanical
and digital), MCUs, and indicator LEDs. For good noise suppression, the analog ground plane
should only be connected to the digital ground plane at one point, e.g., a star ground.
22 CF Calibration Frequency Logic Output. The CF logic output gives instantaneous real power informa-
tion. This output is intended to be used for calibration purposes. Also see SCF pin description.
23, 24 F2, F1 Low-Frequency Logic Outputs. F1 and F2 supply average real power information. The logic
outputs can be used to directly drive electromechanical counters and 2-phase stepper motors. See
Transfer Function section.
PIN FUNCTION DESCRIPTIONS (continued)
TERMINOLOGY
ADC Offset Error
This refers to the dc offset associated with the analog inputs to
the ADCs. It means that with the analog inputs connected to
AGND the ADCs still see an analog input signal of 1 mV to
10 mV. However, when the HPF is switched on, the offset is
removed from the current channel and the power calculation is
not affected by this offset.
Gain Error
The gain error of the ADE7751 is defined as the difference between
the measured output frequency (minus the offset) and the ideal
output frequency. It is measured with a gain of 1 in Channel
V1A. The difference is expressed as a percentage of the ideal
frequency. The ideal frequency is obtained from the transfer
function—see Transfer Function section.
Gain Error Match
The gain error match is defined as the gain error (minus the
offset) obtained when switching between a gain of 1 and a gain
of 2, 8, or 16. It is expressed as a percentage of the output
frequency obtained under a gain of 1. This gives the gain
error observed when the gain selection is changed from
1 to 2, 8, or 16.
Measurement Error
The error associated with the energy measurement made by the
ADE7751 is defined by the following formula:
Percentage Error =
Energy Registered by the ADE7751– True Energy
True Energy ×100%
Phase Error Between Channels
The HPF (high-pass filter) in Channel 1 has a phase lead
response. To offset this phase response and equalize the phase
response between channels, a phase correction network is also
placed in Channel 1. The phase correction network matches the
phase to within ±0.1° over a range of 45 Hz to 65 Hz and ±0.2°
over a range 40 Hz to 1 kHz (see Figures 10 and 11).
Power Supply Rejection
This quantifies the ADE7751 measurement error as a percent-
age of reading when the power supplies are varied.
For the ac PSR measurement, a reading at nominal supplies
(5 V) is taken. A 200 mV rms/100 Hz signal is then introduced
onto the supplies and a second reading is obtained under the
same input signal levels. Any error introduced is expressed as a
percentage of the reading—see Measurement Error definition.
For the dc PSR measurement, a reading at nominal supplies
(5 V) is taken. The supplies are then varied ±5% and a second
reading is obtained with the same input signal levels. Any error
introduced is again expressed as a percentage of the reading.
REV. 0
ADE7751
–7–
Typical Performance Characteristics–
AMPS
–0.50
0.01
ERROR – %
0.10 1.00 10.0 100
–0.40
–0.30
–0.20
–0.10
0.00
0.10
0.20
0.30
0.40
0.50
PF = 1
GAIN = 1
ON-CHIP REFERENCE
–40ⴗC
+25ⴗC
+85ⴗC
TPC 1. Error as a % of Reading (Gain = 1)
AMPS
0.25
0.00
0.01 1000.1
ERROR – %
110
0.20
0.15
0.10
0.05
–0.05
PF = 1
GAIN = 2
ON-CHIP REFERENCE
+85ⴗC
–40ⴗC
+25ⴗC
TPC 2. Error as a % of Reading (Gain = 2)
AMPS
0.20
–0.10
0.01 0.1
ERROR – %
110
0.15
0.10
0.05
0.00
–0.15
PF = 1
GAIN = 8
ON-CHIP REFERENCE
–0.05
100
+85ⴗC
+25ⴗC
–40ⴗC
TPC 3. Error as a % of Reading (Gain = 8)
AMPS
0.60
–0.60
0.01 0.1
ERROR – %
110
0.40
0.20
0.00
–0.20
–0.80
PF = 1
GAIN = 16
ON-CHIP REFERENCE
–0.40
100
+85ⴗC PF = 1
+25ⴗC PF = 1
–40ⴗC PF = 1
TPC 4. Error as a % of Reading (Gain = 16)
AMPS
0.40
–0.20
0.01 0.1
ERROR – %
110
0.30
0.20
0.10
0.00
–0.30
PF = 0.5
GAIN = 1
ON-CHIP REFERENCE
–0.10
100
–40ⴗC PF = 0.5
+25ⴗC PF = 0.5
+25ⴗC PF = 1
+85ⴗC PF = 0.5
TPC 5. Error as a % of Reading (PF = 0.5, Gain = 1)
AMPS
0.30
–0.20
0.01 1000.1
ERROR – %
110
0.20
0.10
0.00
–0.10
–0.30
PF = 0.5
GAIN = 2
ON-CHIP REFERENCE
+85ⴗC PF = 0.5
+25ⴗC PF = 1
–40ⴗC PF = 0.5
+25ⴗC PF = 0.5
TPC 6. Error as a % of Reading (Gain = 2)
REV. 0
ADE7751
–8–
AMPS
0.20
0.01 1000.1
ERROR – %
110
0.10
0.00
–0.10
–0.20
–0.30
–0.40
–0.50
–0.60
PF = 0.5
GAIN = 8
ON-CHIP REFERENCE
+25ⴗC PF=0.5
+25ⴗC PF=1
+85ⴗC PF=0.5
–40ⴗC PF=0.5
TPC 7. Error as a % of Reading (PF = 0.5, Gain = 8)
AMPS
0.60
–0.60
0.01 0.1
ERROR – %
110
0.40
0.20
–0.20
–1.00
PF = 0.5
GAIN = 16
ON-CHIP REFERENCE
–0.40
100
–0.80
0.00
+25ⴗC PF = 1
+85ⴗC PF = 0.5
–40ⴗC PF = 0.5
+25ⴗC PF = 0.5
TPC 8. Error as a % of Reading (Gain = 16)
AMPS
0.20
–0.05
0.01 1000.1
ERROR – %
110
0.15
0.10
0.05
0.00
–0.10
–0.15
–0.20
–0.25
–0.30
PF = 1
GAIN = 2
EXTERNAL REFERENCE +85ⴗC
+25ⴗC
–40ⴗC
TPC 9. Error as a % of Reading over Temperature with
an External Reference (Gain = 2)
AMPS
0.30
–0.30
0.01 0.1
ERROR – %
110
0.20
0.10
0.00
–0.10
–0.40
PF = 1
GAIN = 8
EXTERNAL REFERENCE
–0.20
100
+85ⴗC
–40ⴗC
+25ⴗC
TPC 10. Error as a % of Reading over Temperature with
an External Reference (Gain = 8)
AMPS
0.60
–0.60
0.01 0.1
ERROR – %
110
0.40
0.20
0.00
–0.20
–0.80
PF = 1
GAIN = 16
EXTERNAL REFERENCE
–0.40
100
+85ⴗC PF = 0.5
–40ⴗC PF = 1
+25ⴗC PF = 1
TPC 11. Error as a % of Reading over Temperature with
an External Reference (Gain = 16)
10
ⴚ15 15ⴚ10 ⴚ50
16
14
12
8
6
4
2
0
510
DISTRIBUTION
CHARACTERISTICS
NUMBER OF PTS: 138
MINIMUM: –11.1367
MAXIMUM: +10.1775
MEAN: –1.44576
STD DEV: 4.6670
GAIN = 1
TEMP = 25ⴗC
TPC 12. Channel 1 Offset Distribution (Gain = 1)
REV. 0
ADE7751
–9–
15
ⴚ15 15ⴚ10 ⴚ50
21
18
12
9
6
3
0
510
DISTRIBUTION
CHARACTERISTICS
NUMBER OF PTS: 138
MINIMUM: –7.01774
MAXIMUM: +6.65068
MEAN: –0.421358
STD DEV: 2.974
GAIN = 2
TEMP = 25ⴗC
TPC 13. Channel 1 Offset Distribution (Gain = 2)
25
ⴚ15 15ⴚ10 ⴚ50
35
30
20
15
10
5
0
510
DISTRIBUTION
CHARACTERISTICS
NUMBER OF PTS: 138
MINIMUM: –5.36107
MAXIMUM: +4.30413
MEAN: 0.346894
STD DEV: 1.86651
GAIN = 8
TEMP = 25ⴗC
TPC 14. Channel 1 Offset Distribution (Gain = 8)
25
–15 15–10 –5 0
35
30
20
15
10
5
0
510
DISTRIBUTION
CHARACTERISTICS
NUMBER OF PTS: 138
MINIMUM: +4.37379
MAXIMUM: –5.08496
MEAN: 0.47494
STD DEV: 1.71819
GAIN = 16
TEMP = 25ⴗ C
TPC 15. Channel 1 Offset Distribution (Gain = 16)
AVDD AC/DC DVDD
V1A
V1B
33nF
1k⍀
V1N
REF
IN/OUT
100nF
33nF
931⍀
V2N
33nF
33nF V2P
100nF
930k⍀
220V
10F
1k⍀
931⍀
10F
100nF 10F
V
DD
RESET AGND DGND
F1
F2
CF
REVP
FAULT
CLKOUT
CLKIN
G0
G1
S0
S1
SCF
Y1
3.58MHz
100nF 100nF
100nF
22pF
22pF
40A TO
40mA
GAIN
SELECT
U3
PS2501-1
K9
K10
U1
ADE7751
10k⍀
V
DD
V
DD
GAIN
1
2
8
16
18.2⍀
8.2⍀
2.2⍀
0.68⍀
RB
33nF
1k⍀
RB
RB
TPC 16. Test Circuit for Performance Curves
REV. 0
ADE7751
–10–
THEORY OF OPERATION
The two ADCs digitize the voltage and current signals from the
current and voltage transducers. These ADCs are 16-bit second
order sigma-delta converters with an oversampling rate of 900 kHz.
This analog input structure greatly simplifies transducer interfacing
by providing a wide dynamic range for direct connection to the
transducer and also by simplifying the antialiasing filter design.
A programmable gain stage in the current channel further
facilitates easy transducer interfacing. A high-pass filter in the
current channel removes any dc component from the current
signal. This eliminates any inaccuracies in the real power calcu-
lation due to offsets in the voltage or current signals—see
HPF and Offset Effects section.
The real power calculation is derived from the instantaneous
power signal. The instantaneous power signal is generated by a
direct multiplication of the current and voltage signals. In order
to extract the real power component (i.e., the dc component), the
instantaneous power signal is low-pass filtered. Figure 2 illustrates
the instantaneous real power signal and shows how the real power
information can be extracted by low-pass filtering the instantaneous
power signal. This scheme correctly calculates real power for
nonsinusoidal current and voltage waveforms at all power factors.
All signal processing is carried out in the digital domain for
superior stability over temperature and time.
LPF
DIGITAL-TO-
FREQUENCY
F1
F2
CH1
MULTIPLIER
PGA
CH2 ADC
VⴛI
2
VⴛI
VⴛI
2
p(t) = i(t)ⴛv(t)
WHERE:
v(t) = Vⴛcos(t)
i(t) = Iⴛcos(t)
p(t) = VⴛI
2
{
1+cos(2t)}
⌺
⌺
ADC
TIME
HPF
DIGITAL-TO-
FREQUENCY
CF
INSTANTANEOUS REAL
POWER SIGNAL
INSTANTANEOUS
POWER SIGNAL – p(t)
Figure 2. Signal Processing Block Diagram
The low-frequency output of the ADE7751 is generated by
accumulating this real power information. This low frequency
inherently means a long accumulation time between output
pulses. The output frequency is therefore proportional to the
average real power. This average real power information can in
turn be accumulated (e.g., by a counter) to generate real-energy
information. Because of its high output frequency, and hence
shorter integration time, the CF output is proportional to the
instantaneous real power. This is useful for system calibration
purposes that would take place under steady load conditions.
Power Factor Considerations
The method used to extract the real power information from the
instantaneous power signal (i.e., by low-pass filtering) is still
valid even when the voltage and current signals are not in phase.
Figure 3 displays the unity power factor condition and a DPF
(displacement power factor) = 0.5, i.e., current signal lagging
the voltage by 60°. If we assume the voltage and current waveforms
are sinusoidal, the real power component of the instantaneous
power signal (i.e., the dc term) is given by:
VI×
×°
()
260cos
(1)
This is the correct real power calculation.
INSTANTANEOUS
REAL POWER SIGNAL
INSTANTANEOUS
POWER SIGNAL
VⴛI
2ⴛcos(60ⴗ)
VⴛI
2
INSTANTANEOUS
POWER SIGNAL
INSTANTANEOUS
REAL POWER SIGNAL
60ⴗ
CURRENT
VOLTAGE
CURRENT
VOLTAGE
0V
0V
Figure 3. DC Component of Instantaneous Power Signal
Conveys Real Power Information PF < 1
Nonsinusoidal Voltage and Current
The real power calculation method also holds true for nonsinu-
soidal current and voltage waveforms. All voltage and current
waveforms in practical applications will have some harmonic
content. Using the Fourier Transform, instantaneous voltage
and current waveforms can be expressed in terms of their
harmonic content.
vt V V h t
Oh h
h
() sin( )=+×
∑×+
≠
∞
20ωα
(2)
where:
v(t)= The instantaneous voltage
V
O
=The average value
V
h
=The rms value of voltage harmonic h
and
␣
h
=The phase angle of the voltage harmonic
it I I
h
ht
Oh h
() sin( )=+×
≠×+
∞
∑
20ωβ
(3)
where:
i(t)=The instantaneous current
I
O
=The dc component
I
h
=The rms value of current harmonic h
and

h
=The phase angle of the current harmonic
REV. 0
ADE7751
–11–
Using Equations 2 and 3, the real power P can be expressed in
terms of its fundamental real power (P
1
) and harmonic real
power (P
H
).
P = P1 + PH
where:
PVI
111 1
111
=×
=
cos( )
–
φ
φαβ
(4)
and (5)
As shown in Equation 5 above, a harmonic real power compo-
nent is generated for every harmonic, provided that harmonic is
present in both the voltage and current waveforms. The power
factor calculation has been shown previously to be accurate in
the case of a pure sinusoid, therefore the harmonic real power
must also correctly account for the power factor since it is made
up of a series of pure sinusoids.
Note that the input bandwidth of the analog inputs is 14 kHz
with a master clock frequency of 3.5795 MHz.
ANALOG INPUTS
Channel V2 (Voltage Channel)
The output of the line voltage transducer is connected to the
ADE7751 at this analog input. Channel V2 is a fully differen-
tial voltage input. The maximum peak differential signal on
Channel 2 is ±660 mV. Figure 4 illustrates the maximum
signal levels that can be connected to the ADE7751 Channel 2.
DIFFERENTIAL INPUT
ⴞ600mV MAX PEAK
+600mV
AGND
VCM
V2
V2P
VCM
–600mV
COMMON MODE
ⴞ100mV MAX
V2N
V2
Figure 4. Maximum Signal Levels, Channel 2
Channel 2 must be driven from a common-mode voltage, i.e.,
the differential voltage signal on the input must be referenced to
a common mode (usually AGND). The analog inputs of the
ADE7751 can be driven with common-mode voltages of up to
100 mV with respect to AGND. However, best results are
achieved using a common mode equal to AGND.
Channel V1 (Current Channel)
The voltage outputs from the current transducers are connected
to the ADE7751 here. Channel V1 has two voltage inputs, namely
V1A and V1B. These inputs are fully differential with respect to
V1N. However, at any one time, only one is selected to perform
the power calculation—see Fault Detection section.
The analog inputs V1A, V1B, and V1N have the same maximum
signal level restrictions as V2P and V2N. However, Channel 1
has a programmable gain amplifier (PGA) with user-selectable
gains of 1, 2, 8, or 16—see Table I. These gains facilitate easy
transducer interfacing.
Figure 5 illustrates the maximum signal levels on V1A, V1B,
and V1N. The maximum differential voltage is ±660 mV divided
by the gain selection. Again, the differential voltage signal on the
inputs must be referenced to a common mode, e.g., AGND. The
maximum common-mode signal is ±100 mV as shown in Figure 5.
V1A, V1B DIFFERENTIAL INPUT A
ⴞ660mV/GAIN MAX PEAK
AGND
V
CM
V1A
V
CM
–660mV
GAIN
COMMON MODE
ⴞ100mV MAX
V1N
+660mV
GAIN
V1 V1B
DIFFERENTIAL INPUT B
ⴞ660mV/GAIN MAX PEAK
V1
Figure 5. Maximum Signal Levels, Channel 1
Table I.
Maximum
G1 G0 Gain Differential Signal
00 1±660 mV
01 2±330 mV
10 8±82 mV
11 16±41 mV
Typical Connection Diagrams
Figure 6 shows a typical connection diagram for Channel V1.
Here the analog inputs are being used to monitor both the
phase and neutral currents. Because of the large potential
difference between the phase and neutral, two CTs (current
transformers) must be used to provide the isolation. Notice
both CTs are referenced to AGND (analog ground), hence
the common-mode voltage is 0 V. The CT turns ratio and
burden resistor (Rb) are selected to give a peak differential
voltage of ±660 mV/gain.
V1B
ⴞ660mV
GAIN
CT
IN
V1A
AGND
Rb
Rf
CT
NEUTRALPHASE
IP V1N
Cf
Cf
Rb ⴞ660mV
GAIN
Rf
Figure 6. Typical Connection for Channel 1
PVI
Hhh h
hhh
h
=∑××
=
≠
∞
1cos( )
–
φ
φαβ
REV. 0
ADE7751
–12–
Figure 7 shows two typical connections for Channel V2. The first
option uses a PT (potential transformer) to provide complete isola-
tion from the mains voltage. In the second option, the ADE7751
is biased around the neutral wire and a resistor divider is used to
provide a voltage signal that is proportional to the line voltage.
Adjusting the ratio of Ra and Rb is also a convenient way of
carrying out a gain calibration on the meter.
ⴞ660mV
Ra
Rb
VR
V2P
AGND
Rf
Rf
CT
NEUTRALPHASE
V2N
Cf
Cf
ⴞ660mV
V2P
Rf
NEUTRALPHASE
V2N
Cf
Cf
NOTE
Ra Rf;
Rb + VR = R
f
Figure 7. Typical Connections for Channel 2
POWER SUPPLY MONITOR
The ADE7751 contains an on-chip power supply monitor. The
analog supply (AV
DD
) is continuously monitored by the ADE7751.
If the supply is less than 4 V ± 5%, the ADE7751 will be reset.
This is useful to ensure correct device start up at power-up and
power-down. The power supply monitor has built-in hysteresis
and filtering. This gives a high degree of immunity to false
triggering due to noisy supplies.
As can be seen in Figure 8, the trigger level is nominally set at 4 V.
The tolerance on this trigger level is about ±5%. The power supply
and decoupling for the part should be such that the ripple at AV
DD
does not exceed 5 V ± 5% as specified for normal operation.
AV
DD
5V
4V
0V
INTERNAL
RESET RESET
TIME
ACTIVE RESET
Figure 8. On-Chip Power Supply Monitor
HPF and Offset Effects
Figure 9 shows the effect of offsets on the real power calculation. As
shown in Figure 9, an offset on Channel 1 and Channel 2 will
contribute a dc component after multiplication. Since this dc
component is extracted by the LPF and used to generate the real
power information, the offsets will have contributed a constant
error to the real power calculation. This problem is easily avoided by
enabling the HPF (i.e., pin AC/DC is set logic high) in Channel 1.
By removing the offset from at least one channel, no error component
can be generated at dc by the multiplication. Error terms at cos(ωt)
are removed by the LPF and the digital-to-frequency conversion—
see Digital-to-Frequency Conversion section.
VtV I tI
VIVIVI t
VI t VI t
OS OS
OS OS OS
OS
cos( ) cos( )
cos( )
cos( ) cos( )
ωω
ω
ωω
+
()
×× +
()
=
×+×+××
+× × + ××
2
22
V
OS
ⴛ I
OS
I
OS
ⴛ V
V
OS
ⴛ I
DC COMPONENT (INCLUDING ERROR TERM)
IS EXTRACTED BY THE LPF FOR REAL
POWER CALCULATION
2
FREQUENCY – RAD/S
2
V ⴛ I
0
Figure 9. Effect of Channel Offsets on the
Real Power Calculation
The HPF in Channel 1 has an associated phase response that is
compensated for on-chip. The phase compensation is activated
when the HPF is enabled and is disabled when the HPF is not
activated. Figures 10 and 11 show the phase error between chan-
nels with the compensation network activated. The ADE7751
is phase compensated up to 1 kHz as shown. This will ensure
correct active harmonic power calculation even at low-power factors.
FREQUENCY – Hz
0100
PHASE – Degrees
–0.05
–0.10
0
0.05
0.10
0.15
0.20
0.25
0.30
200 300 400 500 600 700 800 900 1000
Figure 10. Phase Error Between Channels (0 Hz to 1 kHz)
REV. 0 –13–
ADE7751
FREQUENCY – Hz
40
PHASE – Degrees
–0.05
–0.10
0
0.05
0.10
0.15
0.20
0.25
0.30
45 50 55 60 65 70
Figure 11. Phase Error Between Channels (40 Hz to 70 Hz)
DIGITAL-TO-FREQUENCY CONVERSION
As previously described, the digital output of the low-pass filter
after multiplication contains the real power information. However,
since this LPF is not an ideal “brick wall” filter implementation,
the output signal also contains attenuated components at
the line frequency and its harmonics, i.e., cos(hωt) where
h = 1, 2, 3, . . . and so on.
The magnitude response of the filter is given by:
Hf fHz
()
=+
()
1
189/.
(6)
For a line frequency of 50 Hz, this would give an attenuation of
the 2 ω (100 Hz) component of approximately –22 dB. The
dominating harmonic will be at twice the line frequency, i.e.,
cos(2ωt), due to the instantaneous power signal.
Figure 12 shows the instantaneous real power signal output of LPF,
which still contains a significant amount of instantaneous power
information, i.e., cos(2ωt). This signal is then passed to the
digital-to-frequency converter where it is integrated (accumulated)
over time in order to produce an output frequency. This
accumulation of the signal will suppress or average out any
non-dc components in the instantaneous real power signal. The
average value of a sinusoidal signal is zero. Hence, the frequency
generated by the ADE7751 is proportional to the average real
power. Figure 12 shows the digital-to-frequency conversion
for steady load conditions, i.e., constant voltage and current.
As shown in the diagram, the frequency output CF varies over
time, even under steady load conditions. This frequency variation
is primarily due to the cos(2ωt) component in the instantaneous
real power signal. The output frequency on CF can be up to 128
times higher than the frequency on F1 and F2. This higher
output frequency is generated by accumulating the instantaneous
real power signal over a much shorter time while converting it to
a frequency. This shorter accumulation period means less aver-
aging of the cos(2ωt) component. As a consequence, some of
this instantaneous power signal passes through the digital-to-
frequency conversion. This will not be a problem in the application.
Where CF is used for calibration purposes, the frequency should
be averaged by the frequency counter. This will remove any ripple.
If CF is being used to measure energy, e.g., in a microprocessor-
based application, the CF output should also be averaged to
calculate power. However, if an energy measurement is being
made by counting pulses, no averaging is required. Because the
outputs F1 and F2 operate at a much lower frequency, a lot
more averaging of the instantaneous real power signal is carried
out. The result is a greatly attenuated sinusoidal content and a
virtually ripple-free frequency output.
2
V ⴛ I
2
FREQUENCY – RAD/S
LPF
DIGITAL-TO-
FREQUENCY
F1
F2
⌺
⌺
DIGITAL-TO-
FREQUENCY
CF
INSTANTANEOUS REAL POWER SIGNAL
(FREQUENCY DOMAIN)
MULTIPLIER
TIME
FREQUENCY
F1
FREQUENCY
CF
TIME
V
I
0
LPF TO EXTRACT
REAL POWER
(DC TERM)
cos(2t)
ATTENUATED BY LPF
Figure 12. Real Power-to-Frequency Conversion
FAULT DETECTION
The ADE7751 incorporates a novel fault detection scheme that
warns of fault conditions and allows the ADE7751 to continue
accurate billing during a fault event. The fault detection function is
designed to work over a line frequency of 45 Hz to 55 Hz. The
ADE7751 does this by continuously monitoring both the phase
and neutral (return) currents. A fault is indicated when these
currents differ by more than 12.5%. However, even during a
fault, the output pulse rate on F1 and F2 is generated using the
larger of the two currents. Because the ADE7751 looks for a
difference between the signals on V1A and V1B, it is important
that both current transducers are closely matched.
On power-up the output pulse rate of the ADE7751 is proportional
to the product of the signals on Channel V1A and Channel 2. If
there is a difference of greater than 12.5% between V1A and
V1B on power-up, the fault indicator (FAULT) will go active
after about one second. In addition, if V1B is greater than V1A
the ADE7751 will select V1B as the input. The fault detection is
automatically disabled when the voltage signal on Channel 1 is less
than 0.5% of the full-scale input range. This will eliminate false
detection of a fault due to noise at light loads.
Fault with Active Input Greater than Inactive Input
If V1A is the active current input (i.e., is being used for billing),
and the signal on V1B (inactive input) falls by more than 12.5%
of V1A, the fault indicator will go active. Both analog inputs are
filtered and averaged to prevent false triggering of this logic
output. As a consequence of the filtering, there is a time delay of
approximately one second on the logic output FAULT after the
fault event. The FAULT logic output is independent of any activ-
ity on outputs F1 or F2. Figure 13 illustrates one condition under
which FAULT becomes active. Since V1A is the active input and it
is still greater than V1B, billing is maintained on VIA, i.e., no swap
to the V1B input will occur. V1A remains the active input.
REV. 0
ADE7751
–14–
V1B < 87.5% OF V1A
0V
V1A
V1B
V1A
V1A
V1N
AGND
V1B
V1B
FILTER
AND
COMPARE
TO
MULTIPLIER
FAULT
A
B
Figure 13. Fault Conditions for Inactive Input
Less than Active Input
Fault with V1B Greater than V1A
Figure 14 illustrates another fault condition. If V1A is the active
input (i.e., is being used for billing) and the voltage signal on
V1B (inactive input) becomes greater than 114% of V1A, the
FAULT indicator goes active, and there is also a swap over to
the V1B input. The analog input V1B has now become the
active input. Again there is a time delay of about 1.2 seconds
associated with this swap. V1A will not swap back to being the
active channel until V1A becomes greater than 114% of V1B.
However, the FAULT indicator will become inactive as soon as
V1A is within 12.5% of V1B. This threshold eliminates poten-
tial chatter between V1A and V1B.
V1A < 87.5% OF V1B
OR
V1B > 114% OF V1A
0V
V1A
V1B
V1A
V1A
V1N
AGND
V1B
V1B
FILTER
AND
COMPARE
TO
MULTIPLIER
FAULT
A
B
Figure 14. Fault Conditions for Inactive Input
Greater than Active Input
Calibration Concerns
Typically, when a meter is being calibrated, the voltage and current
circuits are separated as shown in Figure 15. This means that
current will only pass through the phase or neutral circuit. Figure 15
shows current being passed through the phase circuit. This is the
preferred option since the ADE7751 starts billing on the input
V1A on power-up. The phase circuit CT is connected to V1A in
the diagram. Since there is no current in the neutral circuit, the
FAULT indicator will come on under these conditions. However,
this does not affect the accuracy of the calibration and can be
used as a means to test the functionality of the fault detection.
If the neutral circuit is chosen for the current circuit in the
arrangement shown in Figure 15, it may have implications for
the calibration accuracy. The ADE7751 will power up with the
V1A input active as normal. However, since there is no current
in the phase circuit, the signal on V1A is zero. This will cause a
FAULT to be flagged and the active input to be swapped to V1B
(Neutral). The meter may be calibrated in this mode, but the
phase and neutral CTs may differ slightly. Since under no-fault
conditions all billing is carried out using the phase CT, the meter
should be calibrated using the phase circuit. Of course, both
phase and neutral circuits may be calibrated.
Ib
V
240Vrms
NOTE
Ra Rf;
Rb + VR = Rf
Rb
Rb
V1A
0V
V1B
CT
V1A
AGND
Rf
CT
NEUTRAL
PHASE
V1N
Cf
Cf
Rf
Rb
VR
V2P
RfV2N
Cf
Cf
TEST
CURRENT Ib
Ra
Figure 15. Fault Conditions for Inactive
Input Greater than Active Input
TRANSFER FUNCTION
Frequency Outputs F1 and F2
The ADE7751 calculates the product of two voltage signals (on
Channel 1 and Channel 2) and then low-pass filters this product
to extract real power information. This real power information
is then converted to a frequency. The frequency information is
output on F1 and F2 in the form of active low pulses. The pulse
rate at these outputs is relatively low, e.g., 0.34 Hz maximum for
ac signals with S0 = S1 = 0 (see Table III). This means that the
frequency at these outputs is generated from real power informa-
tion accumulated over a relatively long period of time. The result is
an output frequency that is proportional to the average real
power. The averaging of the real power signal is implicit to the
digital-to-frequency conversion. The output frequency or pulse
rate is related to the input voltage signals by the following equation.
Freq VV Gain F
V
REF
=×× × ×574 1 2
14
2
.
–
(7)
where,
Freq =Output frequency on F1 and F2 (Hz)
V1=Differential rms voltage signal on Channel 1 (Volts)
V2=Differential rms voltage signal on Channel 2 (Volts)
Gain =1, 2, 8, or 16, depending on the PGA gain selection
made using logic inputs G0 and G1
V
REF
=The reference voltage (2.5 V ± 8%) (Volts)
F
1–4
=One of four possible frequencies selected by using the
logic inputs S0 and S1 (see Table II)
Table II.
S1 S0 F
1–4
(Hz) XTAL/CLKIN*
00 1.7 3.579 MHz/2
21
01 3.4 3.579 MHz/2
20
10 6.8 3.579 MHz/2
19
11 13.6 3.579 MHz/2
18
*F
1–4
are a binary fraction of the master clock and will thus vary if the specified
CLKIN frequency is altered.
REV. 0
ADE7751
–15–
Table IV.
F
1–4
CF Max for AC Signals
SCF S1 S0 (Hz) (Hz)
1001.7 128 × F1, F2 = 43.52
0001.7 64 × F1, F2 = 21.76
1013.4 64 × F1, F2 = 43.52
0013.4 32 × F1, F2 = 21.76
1106.8 32 × F1, F2 = 43.52
0106.8 16 × F1, F2 = 21.76
11113.6 16 × F1, F2 = 43.52
01113.6 8 × F1, F2 = 21.76
SELECTING A FREQUENCY FOR AN ENERGY METER
APPLICATION
As shown in Table II, the user can select one of four frequencies.
This frequency selection determines the maximum frequency
on F1 and F2. These outputs are intended to be used to drive
the energy register (electromechanical or other). Since only four
different output frequencies can be selected, the available
frequency selection has been optimized for a meter constant of
100 imp/kWhr with a maximum current of between 10 A and
120 A. Table V shows the output frequency for several maximum
currents (I
MAX
) with a line voltage of 220 V. In all cases, the
meter constant is 100 imp/kWhr.
Table V.
I
MAX
F1 and F2 (Hz)
12.5 A 0.076
25 A 0.153
40 A 0.244
60 A 0.367
80 A 0.489
120 A 0.733
The F
1–4
frequencies allow complete coverage of this range of
output frequencies on F1 and F2. When designing an energy
meter, the nominal design voltage on Channel 2 (voltage) should
be set to half scale to allow for calibration of the meter constant.
The current channel should also be no more than half scale when the
meter sees maximum load. This will allow overcurrent signals and
signals with high crest factors to be accommodated. Table VI
shows the output frequency on F1 and F2 when both analog
inputs are half scale. The frequencies listed in Table VI align very
well with those listed in Table V for maximum load.
Table VI.
Frequency on F1 and F2 –
CH1 and CH2
S1 S0 F
1–4
Half-Scale AC Inputs
00 1.7 0.085 Hz
01 3.4 0.17 Hz
10 6.8 0.34 Hz
11 13.6 0.68 Hz
Example 1
If full-scale differential dc voltages of +660 mV and –660 mV are
applied to V1 and V2 respectively (660 mV is the maximum
differential voltage that can be connected to Channel 1 and
Channel 2), the expected output frequency is calculated as follows.
Gain = 1, G0 = G1 = 0
F
1–4
=1.7 Hz, S0 = S1 = 0
V1 = +660 mV dc = 0.66 V (rms of dc = dc)
V2 = –660 mV dc = 0.66 V (rms of dc = |dc|)
V
REF
=2.5 V (nominal reference value)
Note: If the on-chip reference is used, actual output frequencies
may vary from device to device due to reference tolerance of ±8%.
Freq Hz Hz=×××× =
574 066 066 1 17
25 068
2
... .
..
(8)
Example 2
In this example, if ac voltages of ±660 mV peak are applied to
V1 and V2, the expected output frequency is calculated as follows.
Gain = 1, G0 = G1 = 0
F
1–4
=1.7 Hz, S0 = S1 = 0
V1 = rms of 660 mV peak ac = 0.66/√2 V
V2 = rms of 660 mV peak ac = 0.66/√2 V
V
REF
=2.5 V (nominal reference value)
Note: If the on-chip reference is used, actual output frequencies
may vary from device to device due to reference tolerance of ±8%.
Freq Hz Hz=××××
×× =
574 066 066 1 17
2225
034
2
... .
.
.
(9)
As shown in these two example calculations, the maximum
output frequency for ac inputs is always half of that for dc
input signals. Table III shows a complete listing of all maxi-
mum output frequencies.
Table III.
Max Frequency Max Frequency
S1 S0 for DC Inputs (Hz) for AC Inputs (Hz)
00 0.68 0.34
01 1.36 0.68
10 2.72 1.36
11 5.44 2.72
Frequency Output CF
The pulse output CF (calibration frequency) is intended for use
during calibration. The output pulse rate on CF can be up to 128
times the pulse rate on F1 and F2. The lower the F
1–4
frequency
selected the higher the CF scaling. Table IV shows how the two
frequencies are related depending on the states of the logic inputs
S0, S1, and SCF. Because of its relatively high-pulse rate, the
frequency at this logic output is proportional to the instantaneous
real power. As is the case with F1 and F2, the frequency is derived
from the output of the low-pass filter after multiplication. However,
because the output frequency is high, this real power information
is accumulated over a much shorter time. Hence, less averaging
is carried out in the digital-to-frequency conversion. With much
less averaging of the real power signal, the CF output is much
more responsive to power fluctuations (see Figure 2).
REV. 0
–16–
C02953–0–5/02(0)
PRINTED IN U.S.A.
ADE7751
When selecting a suitable F
1–4
frequency for a meter design, the
frequency output at I
MAX
(maximum load) with a meter constant
of 100 imp/kWhr should be compared with Column 4 of Table VI.
The frequency that is closest in Table VI will determine the best
choice of frequency (F
1–4
). For example, if a meter with a maxi-
mum current of 25 A is being designed, the output frequency on F1
and F2, with a meter constant of 100 imp/kWhr, is 0.153 Hz at 25 A
and 220 V (from Table V). Looking at Table VI, the closest
frequency to 0.153 Hz in column four is 0.17 Hz. Therefore F
2
(3.4 Hz—see Table II) is selected for this design.
Frequency Outputs
Figure 1 shows a timing diagram for the various frequency
outputs. The outputs F1 and F2 are the low-frequency outputs
that can be used to directly drive a stepper motor or electrome-
chanical impulse counter. The F1 and F2 outputs provide two
alternating low-going pulses. The pulsewidth (t
1
) is set at 275 ms
and the time between the falling edges of F1 and F2 (t
3
) is
approximately half the period of F1 (t
2
).
If, however, the period of F1 and F2 falls below 550 ms (1.81 Hz),
the pulsewidth of F1 and F2 is set to half of their period. The maxi-
mum output frequencies for F1 and F2 are shown in Table III.
The high-frequency CF output is intended to be used for
communications and calibration purposes. CF produces a
90 ms wide active high pulse (t
4
) at a frequency that is propor-
tional to active power. The CF output frequencies are given in
Table IV. As in the case of F1 and F2, if the period of CF (t
5
)
falls below 180 ms, the CF pulsewidth is set to half the period.
For example, if the CF frequency is 20 Hz, the CF pulse-
width is 25 ms.
NO LOAD THRESHOLD
The ADE7751 also includes a “no load threshold” and “start-
up current” feature that will eliminate any creep effects in the
meter. The ADE7751 is designed to issue a minimum output
frequency. Any load generating a frequency lower than this
minimum frequency will not cause a pulse to be issued on F1,
F2, or CF. The minimum output frequency is given as 0.0014%
of the full-scale output frequency for each of the F
1–4
frequency
selections (see Table II). For example, for an energy meter with
a meter constant of 100 imp/kWhr on F1, F2 using F
2
(3.4 Hz),
the maximum output frequency at F1 or F2 would be 0.0014%
of 3.4 Hz or 4.76 × 10
–5
Hz. This would be 3.05 × 10
–3
Hz at
CF (64 × F1 Hz). In this example, the no load threshold would be
equivalent to 1.7 W of load or a start-up current of 8 mA at 220 V.
Compare this value to the IEC1036 specification, which states
that the meter must start up with a load equal to or less than
0.4% Ib. For a 5 A(Ib) meter, 0.4% of Ib is equivalent to 20 mA.
24-Shrink Small Outline Package
(RS-24)
24
1
13
12
0.328 (8.33)
0.318 (8.08)
0.311 (7.9)
0.301 (7.64)
0.212 (5.38)
0.205 (5.207)
PIN 1
SEATING
PLANE
0.07 (1.78)
0.066 (1.67)
0.008 (0.203)
0.002 (0.050)
0.0256
(0.65)
BSC
0.078 (1.98)
0.068 (1.73)
0.015 (0.38)
0.010 (0.25) 0.009 (0.229)
0.005 (0.127)
0.037 (0.94)
0.022 (0.559)
8°
0°
24-Lead Plastic DIP
(N-24)
24
112
13
PIN 1
1.275 (32.30)
1.125 (28.60)
0.280 (7.11)
0.240 (6.10)
0.195 (4.95)
0.115 (2.93)
0.015 (0.381)
0.008 (0.204)
0.325 (8.25)
0.300 (7.62)
SEATING
PLANE
0.060 (1.52)
0.015 (0.38)
0.210
(5.33)
MAX
0.022 (0.558)
0.014 (0.356)
0.200 (5.05)
0.125 (3.18)
0.150
(3.81)
MIN
0.100
(2.54)
BSC
0.070 (1.77)
0.045 (1.15)
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
