LM4780 Overture™ Audio
Power Amplifier Series
January 22, 2010
Stereo 60W, Mono 120W Audio Power Amplifier with Mute
General Description
The LM4780 is an stereo audio amplifier capable of typically
delivering 60W per channel of continuous average output
power into an 8Ω load with less than 0.5% THD+N from 20Hz
- 20kHz.
The LM4780 is fully protected utilizing National's Self Peak
Instantaneous Temperature (°Ke) (SPiKeTM) protection cir-
cuitry. SPiKe provides a dynamically optimized Safe Operat-
ing Area (SOA). SPiKe protection completely safeguards the
LM4780's outputs against over-voltage, under-voltage, over-
loads, shorts to the supplies or GND, thermal runaway and
instantaneous temperature peaks. The advanced protection
features of the LM4780 places it in a class above discrete and
hybrid amplifiers.
Each amplifier of the LM4780 has an independent smooth
transition fade-in/out mute.
The LM4780 can easily be configured for bridge or parallel
operation for 120W mono solutions.
Key Specifications
■ Output Power/Channel
with 0.5% THD+N, 1kHz into 8Ω60W (typ)
■THD+N at 2 x 30W into 8Ω (20Hz - 20kHz) 0.03% (typ)
■THD+N at 2 x 30W into 6Ω (20Hz - 20kHz) 0.05% (typ)
■THD+N at 2 x 30W into 4Ω (20Hz - 20kHz) 0.07% (typ)
■Mute Attenuation 110dB (typ)
■PSRR 85dB (min)
■Slew Rate 19V/µs (typ)
Features
■SPiKe Protection
■Low external component count
■Quiet fade-in/out mute mode
■Wide supply range: 20V - 84V
■Signal-to-Noise Ratio ≥ 97dB (ref. to PO = 1W)
Applications
■Audio amplifier for component stereo
■Audio amplifier for compact stereo
■Audio amplifier for self-powered speakers
■Audio amplifier for high-end and HD TVs
Typical Application
20058622
FIGURE 1. Typical Audio Amplifier Application Circuit
Overture® is a registered trademark of National Semiconductor.
© 2010 National Semiconductor Corporation 200586 www.national.com
LM4780 Overture® Audio Power Amplifier Series
Stereo 60W, Mono 120W Audio Power Amplifier with Mute
Absolute Maximum Ratings (Note 1, Note
2)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |V+| + |V-|
(No Signal) 94V
Supply Voltage |V+| + |V-|
(Input Signal) 84V
Common Mode Input Voltage (V+ or V-) and
|V+| + |V-| ≤ 80V
Differential Input Voltage (Note 13) 60V
Output Current Internally Limited
Power Dissipation (Note 3) 125W
ESD Susceptability (Note 4) 3.0kV
ESD Susceptability (Note 5) 200V
Junction Temperature (TJMAX) (Note 9)150°C
Soldering Information
TA Package (10 seconds) 260°C
Storage Temperature -40°C to +150°C
Thermal Resistance
θJA 30°C/W
θJC 0.8°C/W
Operating Ratings (Note 1, Note 2)
Temperature Range
TMIN ≤ TA ≤ TMAX −20°C ≤ TA ≤ +85°C
Supply Voltage |V+| + |V-|20V ≤ VTOTAL ≤ 84V
Note: Operation is guaranteed up to 84V; however, distortion
may be introduced from SPiKe protection circuitry if proper
thermal considerations are not taken into account. Refer to
the Thermal Considerations section for more information.
Electrical Characteristics (Note 1, Note 2)
The following specifications apply for V+ = +35V, V- = −35V, IMUTE = -1mA and RL = 8Ω unless otherwise specified. Limits apply
for TA = 25°C.
Symbol Parameter Conditions
LM4780
Units
(Limits)
Typical Limit
(Note 6) (Note 7,
Note 8)
|V+| + |V-|Power Supply Voltage
(Note 10)GND − V- ≥ 9V 18 20
84
V (min)
V (max)
AMMute Attenuation IMUTE = 0mA 110 80 dB (min)
POOutput Power (RMS)
THD+N = 0.5% (max)
f = 1kHz; f = 20kHz
|V+| = |V-| = 25V, RL = 4Ω
|V+| = |V-| = 30V, RL = 6Ω
|V+| = |V-| = 35V, RL = 8Ω
55
55
60
50
50
50
W (min)
W (min)
W (min)
THD+N Total Harmonic Distortion +
Noise
PO = 30W, f = 20Hz - 20kHz
AV = 26dB
|V+| = |V-| = 25V, RL = 4Ω
|V+| = |V-| = 30V, RL = 6Ω
|V+| = |V-| = 35V, RL = 8Ω
0.07
0.05
0.03
%
%
%
Xtalk Channel Separation (Note 11) PO = 10W, f = 1kHz 70 dB
PO = 10W, f = 10kHz 72 dB
SR Slew Rate VIN = 2.0VP-P, tRISE = 2ns 19 8 V/μs (min)
IDD
Total Quiescent Power VCM = 0V, 110 170 mA (max)
Supply Current VO = 0V, IO = 0A
VOS Input Offset Voltage VCM = 0V, IO = 0mA 1 10 mV (max)
IBInput Bias Current VCM = 0V, IO = 0mA 0.2 1 μA (max)
IOOutput Current Limit |V+| = |V-| = 20V, tON = 10ms 11.5 7 A (min)
VOD
Output Dropout Voltage
(Note 12)
|V+ - VO|, V+ = 28V, IO = +100mA
|V- - VO|, V- = -28V, IO = -100mA
1.6
2.5
2.0
3.0
V (max)
V (max)
PSRR Power Supply Rejection Ratio
(Note 15)
V+ = 40V to 20V, V- = -40V,
VCM = 0V, IO = 0mA 120 85 dB (min)
V+ = 40V, V- = -40V to -20V,
VCM = 0V, IO = 0mA 105 85 dB (min)
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LM4780
Symbol Parameter Conditions
LM4780
Units
(Limits)
Typical Limit
(Note 6) (Note 7,
Note 8)
CMRR Common Mode Rejection Ratio
(Note 15)
V+ = 60V to 20V, V- = -20V to -60V,
VCM = 20V to -20V, IO = 0mA 110 85 dB (min)
AVOL Open Loop Voltage Gain RL = 2kΩ, ΔVO = 40V 115 90 dB (min)
GBWP Gain Bandwidth Product fIN = 100kHz, VIN = 50mVRMS 8 2 MHz (min)
eIN Input Noise IHF-A-Weighting Filter,
RIN = 600Ω (Input Referred) 2.0 10 μV (max)
SNR Signal-to-Noise Ratio
PO = 1WRMS; A-Weighted Filter
fIN = 1kHz, RS = 25Ω 97 dB
PO = 50WRMS; A-Weighted Filter
fIN = 1kHz, RS = 25Ω 114 dB
IMD Intermodulation Distortion 60Hz, 7kHz, 4:1 (SMPTE)
60Hz, 7kHz, 1:1 (SMPTE)
0.004
0.009
%
%
Note 1: All voltages are measured with respect to the ground pins, unless otherwise specified.
Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions
which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters
where no limit is given; however, the typical value is a good indication of device performance.
Note 3: The maximum power dissipation must be de-rated at elevated temperatures and is dictated by TJMAX, θJC, and the ambient temperature TA. The maximum
allowable power dissipation is PDMAX = (TJMAX -TA) / θJC or the number given in the Absolute Maximum Ratings, whichever is lower. For the LM4780, TJMAX =
150°C and the typical θJC is 0.8°C/W. Refer to the Thermal Considerations section for more information.
Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.
Note 5: Machine Model: a 220pF - 240pF discharged through all pins.
Note 6: Typical specifications are measured at 25°C and represent the parametric norm.
Note 7: Tested limits are guaranteed to National's AOQL (Average Outgoing Quality Level).
Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
Note 9: The maximum operating junction temperature is 150°C. However, the instantaneous Safe Operating Area temperature is 250°C.
Note 10: V- must have at least - 9V at its pin with reference to GND in order for the under-voltage protection circuitry to be disabled. In addition, the voltage
differential between V+ and V- must be greater than 14V.
Note 11: Cross talk performance was measured using the demo board shown in the datasheet. PCB layout will affect cross talk. It is recommended that input
and output traces be separated by as much distance as possible. Return ground traces from outputs should be independent back to a single ground point and
use as wide of traces as possible.
Note 12: The output dropout voltage is defined as the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the
Typical Performance Characteristics section.
Note 13: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages V+ = 40V and V- = - 40V.
Note 14: The TA27A is a non-isolated package. The package's metal back, and any heat sink to which it is mounted are connected to the V- potential when using
only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink will be electrically isolated
from V-.
Note 15: DC electrical test.
Note 16: CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).
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LM4780
Bridged Amplifier Application Circuit
20058614
FIGURE 2. Bridged Amplifier Application Circuit
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LM4780
Parallel Amplifier Application Circuit
20058613
FIGURE 3. Parallel Amplifier Application Circuit
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LM4780
Single Supply Application Circuit
20058606
*Optional components dependent upon specific design requirements.
FIGURE 4. Single Supply Amplifier Application Circuit
Auxiliary Amplifier Application Circuit
200586d5
FIGURE 5. Special Audio Amplifier Application Circuit
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LM4780
External Components Description
(Figures 1-5)
Components Functional Description
1RB
Prevents current from entering the amplifier's non-inverting input. This current may pass through to the load
during system power down, because of the amplifier's low input impedance when the undervoltage circuitry is
off. This phenomenon occurs when the V+ and V- supply voltages are below 1.5V.
2RiInverting input resistance. Along with Rf, sets AC gain.
3RfFeedback resistance. Along with Ri, sets AC gain.
4Rf2
(Note 17)
Feedback resistance. Works with Cf and Rf creating a lowpass filter that lowers AC gain at high frequencies.
The -3dB point of the pole occurs when: (Rf - Ri)/2 = Rf // [1/(2πfcCf) + Rf2] for the Non-Inverting configuration
shown in Figure 5.
5Cf
(Note 17)Compensation capacitor. Works with Rf and Rf2 to reduce AC gain at higher frequencies.
6CC
(Note 17)
Compensation capacitor. Reduces the gain at higher frequencies to avoid quasi-saturation oscillations of the
output transistor. Also suppresses external electromagnetic switching noise created from fluorescent lamps.
7Ci
(Note 17)
Feedback capacitor which ensures unity gain at DC. Along with Ri also creates a highpass filter at fc = 1/
(2πRiCi).
8CS
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper
placement and selection of bypass capacitors.
9RV
(Note 17)Acts as a volume control by setting the input voltage level.
10 RIN
(Note 17)
Sets the amplifier's input terminals DC bias point when CIN is present in the circuit. Also works with CIN to create
a highpass filter at fC = 1/(2πRINCIN). If the value of RIN is too large, oscillations may be observed on the outputs
when the inputs are floating. Recommended values are 10kΩ to 47kΩ. Refer to Figure 5.
11 CIN
(Note 17)Input capacitor. Prevents the input signal's DC offsets from being passed onto the amplifier's inputs.
12 RSN
(Note 17)Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
13 CSN
(Note 17)
Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The
pole is set at fC = 1/(2πRSNCSN). Refer to Figure 5.
14 L (Note 17) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the
Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and pass audio
signals to the load. Refer to Figure 5.
15 R (Note 17)
16 RAProvides DC voltage biasing for the transistor Q1 in single supply operation.
17 CAProvides bias filtering for single supply operation.
18 RINP
(Note 17)
Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to the Clicks and
Pops application section for a more detailed explanation of the function of RINP.
19 RBI
Provides input bias current for single supply operation. Refer to the Clicks and Pops application section for a
more detailed explanation of the function of RBI.
20 RE
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the half-
supply point along with CA.
21 RM
Mute resistance set up to allow 0.5mA to be drawn from each MUTE pin to turn the muting function off.
→ RM is calculated using: RM ≤ (|VEE| − 2.6V)/l where l ≥ 0.5mA. Refer to the Mute Attenuation vs Mute Current
curves in the Typical Performance Characteristics section.
22 CMMute capacitance set up to create a large time constant for turn-on and turn-off muting.
23 S1Mute switch. When open or switched to GND, the amplifier will be in mute mode.
24 ROUT Reduces current flow between outputs that are caused by Gain or DC offset differences between the amplifiers.
Note 17: Optional components dependent upon specific design requirements.
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LM4780
Optional External Component
Interaction
The optional external components have specific desired func-
tions. Their values are chosen to reduce the bandwidth and
eliminate unwanted high frequency oscillation. They may,
however, cause certain undesirable effects when they inter-
act. Interaction may occur when the components produce
reactions that are nearly equal to one another. One example
is the coupling capacitor, CC, and the compensation capaci-
tor, Cf. These two components are low impedances at certain
frequencies. They may couple signals from the input to the
output. Please take careful note of basic amplifier component
functionality when selecting the value of these components
and their placement on a printed circuit board (PCB).
The optional external components shown in Figure 4 and
Figure 5, and described above, are applicable in both single
and split supply voltage configurations.
Typical Performance Characteristics
THD+N vs Frequency
± 25V, POUT = 1W & 30W/Channel
RL= 4Ω, 80kHz BW
200586e3
THD+N vs Frequency
± 30V, POUT = 1W & 30W/Channel
RL= 6Ω, 80kHz BW
200586e4
THD+N vs Frequency
± 35V, POUT = 1W & 30W/Channel
RL= 8Ω, 80kHz BW
200586e5
THD+N vs Output Power/Channel
± 25V, RL= 4Ω, 80kHz BW
200586e8
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LM4780
THD+N vs Output Power/Channel
± 30V, RL= 6Ω, 80kHz BW
200586e9
THD+N vs Output Power/Channel
± 35V, RL= 8Ω, 80kHz BW
200586f0
Output Power/Channel
vs Supply Voltage
f = 1kHz, RL = 4Ω, 80kHz BW
20058625
Output Power/Channel
vs Supply Voltage
f = 1kHz, RL = 6Ω, 80kHzBW
20058626
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LM4780
Output Power/Channel
vs Supply Voltage
f = 1kHz, RL = 8Ω, 80kHz BW
20058623
Total Power Dissipation
vs Output Power/Channel
1% THD (max), RL = 4Ω, 80kHz BW
200586a6
Total Power Dissipation
vs Output Power/Channel
1% THD (max), RL = 6Ω, 80kHz BW
200586a7
Total Power Dissipation
vs Output Power/Channel
1% THD (max), RL = 8Ω, 80kHz BW
200586a8
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LM4780
Crosstalk vs Frequency
± 25V, POUT = 10W
RL = 4Ω, 80kHz BW
200586c5
Crosstalk vs Frequency
± 35V, POUT = 10W
RL = 8Ω, 80kHz BW
200586a5
Mute Attenuation
vs Mute Pin Current
POUT = 10W/Channel
200586c6
Supply Current
vs Supply Voltage
200586b4
Large Signal Response
200586c7
Power Supply
Rejection Ratio
200586c8
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LM4780
Common Mode
Rejection Ratio
200586c9
Open Loop
Frequency Response
200586d0
Clipping Voltage
vs Supply Voltage
200586d1
Clipping Voltage
vs Supply Voltage
200586d2
THD+N vs Frequency
± 25V, POUT = 1W & 50W
Bridge Mode (Note 18), RL = 8Ω, 80kHz BW
200586e6
THD+N vs Frequency
± 35V, POUT = 1W & 50W
Parallel Mode (Note 19), RL = 4Ω, 80kHz BW
200586e7
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LM4780
THD+N vs Output Power
± 25V, Bridge Mode (Note 18)
RL = 8Ω, 80kHz BW
200586f1
THD+N vs Output Power
± 35V, Parallel Mode (Note 19)
RL = 4Ω, 80kHz BW
200586f2
Output Power vs
Supply Voltage, Bridge Mode (Note 18)
f = 1kHz, RL = 8Ω, 80kHz BW
20058627
Output Power vs
Supply Voltage, Parallel Mode (Note 19)
f = 1kHz, RL = 4Ω, 80kHz BW
20058624
Safe Area
200586d3
SPiKe
Protection Response
200586d4
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LM4780
Frequency Response of Demo Board
POUT = 10W/Channel = 0dB
RIN = 47kΩ, RL = 8Ω, No Filters
200586e2
Note 18: Bridge mode graphs were taken using the demo board and inverting the signal to the channel B input.
Note 19: Parallel mode graphs were taken using the demo board and connecting each output through a 0.1Ω/3W resistor to the load.
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LM4780
Application Information
MUTE MODE
The muting function allows the user to mute the amplifier. This
can be accomplished as shown in the Typical Application Cir-
cuit. The resistor RM is chosen with reference to the negative
supply voltage and is used in conjunction with a switch. The
switch, when opened or switched to GND, cuts off the current
flow from the MUTE pins to −VEE, thus placing the LM4780
into mute mode. Refer to the Mute Attenuation vs Mute Cur-
rent curves in the Typical Performance Characteristics
section for values of attenuation per current out of each MUTE
pin. The resistance RM is calculated by the following equation:
RM ≤ (|−VEE| − 2.6V) / IMUTE
Where IMUTE ≥ 0.5mA for each MUTE pin.
The MUTE pins can be tied together so that only one resistor
is required for the mute function. The mute resistor value must
be chosen so that a minimum of 1mA is pulled through the
resistor RM. This ensures that each amplifier is fully opera-
tional. Taking into account supply line fluctuations, it is a good
idea to pull out 1mA per MUTE pin or 2mA total if both pins
are tied together.
A turn-on MUTE or soft start circuit may also be used during
power up. A simple circuit like the one shown below may be
used.
200586a3
The RC combination of CM and RM1 may cause the voltage at
point A to change more slowly than the -VEE supply voltage.
Until the voltage at point A is low enough to have 0.5mA of
current per MUTE pin flow through RM2, the IC will be in mute
mode. The series combination of RM1 and RM2 needs to sat-
isfy the mute equation above for all operating voltages or mute
mode may be activated during normal operation. For a longer
turn-on mute time, a larger time constant, τ = RC = RM1CM
(sec), is needed. For the values show above and with the
MUTE pins tied together, the LM4780 will enter play mode
when the voltage at point A is -17.6V. The voltage at point A
is found with Equation (1) below.
VA(t) = (Vf - VO)e-t/τ (Volts) (1)
where:
t = time (sec)
τ = RC (sec)
Vo = Voltage on C at t = 0 (Volts)
Vf = Final voltage, -VEE in this circuit (Volts)
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry
allows the power supplies and their corresponding capacitors
to come up close to their full values before turning on the
LM4780. Since the supplies have essentially settled to their
final value, no DC output spikes occur. At power down, the
outputs of the LM4780 are forced to ground before the power
supply voltages fully decay preventing transients on the out-
put.
OVER-VOLTAGE PROTECTION
The LM4780 contains over-voltage protection circuitry that
limits the output current while also providing voltage clamp-
ing. The clamp does not, however, use internal clamping
diodes. The clamping effect is quite the same because the
output transistors are designed to work alternately by sinking
large current spikes.
SPiKe PROTECTION
The LM4780 is protected from instantaneous peak-tempera-
ture stressing of the power transistor array. The Safe Oper-
ating graph in the Typical Performance Characteristics
section shows the area of device operation where SPiKe
Protection Circuitry is not enabled. The SPiKe Protection Re-
sponse waveform graph shows the waveform distortion when
SPiKe is enabled. Please refer to AN-898 for more detailed
information.
THERMAL PROTECTION
The LM4780 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die exceeds 150°C, the LM4780 shuts
down. It starts operating again when the die temperature
drops to about 145°C, but if the temperature again begins to
rise, shutdown will occur again above 150°C. Therefore, the
device is allowed to heat up to a relatively high temperature
if the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion be-
tween the thermal shutdown temperature limits of 150°C and
145°C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the heat
sink used, the heat sink should be chosen so that thermal
shutdown is not activated during normal operation. Using the
best heat sink possible within the cost and space constraints
of the system will improve the long-term reliability of any pow-
er semiconductor device, as discussed in the Determining
the Correct Heat Sink section.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understanding
if optimum power output is to be obtained. An incorrect max-
imum power dissipation calculation may result in inadequate
heat sinking causing thermal shutdown and thus limiting the
output power.
Equation 2 shows the theoretical maximum power dissipation
point of each amplifier in a single-ended configuration where
VCC is the total supply voltage.
PDMAX = (VCC)2 / 2π2RL(2)
Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calculated.
The package dissipation is twice the number which results
from Equation 2 since there are two amplifiers in each
LM4780. Refer to the graphs of Power Dissipation versus
Output Power in the Typical Performance Characteristics
section which show the actual full range of power dissipation
not just the maximum theoretical point that results from Equa-
tion 2.
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LM4780
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such that
the thermal protection circuitry is not activated under normal
circumstances.
The thermal resistance from the die to the outside air, θJA
(junction to ambient), is a combination of three thermal resis-
tances, θJC (junction to case), θCS (case to sink), and θSA (sink
to ambient). The thermal resistance, θJC (junction to case), of
the LM4780T is 0.8°C/W. Using Thermalloy Thermacote ther-
mal compound, the thermal resistance, θCS (case to sink), is
about 0.2°C/W. Since convection heat flow (power dissipa-
tion) is analogous to current flow, thermal resistance is anal-
ogous to electrical resistance, and temperature drops are
analogous to voltage drops, the power dissipation out of the
LM4780 is equal to the following:
PDMAX = (TJMAX−TAMB) / θJA (3)
where TJMAX = 150°C, TAMB is the system ambient tempera-
ture and θJA = θJC + θCS + θSA.
20058652
Once the maximum package power dissipation has been cal-
culated using Equation 2, the maximum thermal resistance,
θSA, (heat sink to ambient) in °C/W for a heat sink can be
calculated. This calculation is made using Equation 4 which
is derived by solving for θSA in Equation 3.
θSA = [(TJMAX−TAMB)−PDMAX(θJC +θCS)] / PDMAX (4)
Again it must be noted that the value of θSA is dependent upon
the system designer's amplifier requirements. If the ambient
temperature that the audio amplifier is to be working under is
higher than 25°C, then the thermal resistance for the heat
sink, given all other things are equal, will need to be smaller.
SUPPLY BYPASSING
The LM4780 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM4780 should have its supply leads bypassed with low-in-
ductance capacitors having short leads that are located close
to the package terminals. Inadequate power supply bypass-
ing will manifest itself by a low frequency oscillation known as
“motorboating” or by high frequency instabilities. These in-
stabilities can be eliminated through multiple bypassing uti-
lizing a large tantalum or electrolytic capacitor (10μF or larger)
which is used to absorb low frequency variations and a small
ceramic capacitor (0.1μF) to prevent any high frequency feed-
back through the power supply lines.
If adequate bypassing is not provided, the current in the sup-
ply leads which is a rectified component of the load current
may be fed back into internal circuitry. This signal causes dis-
tortion at high frequencies requiring that the supplies be by-
passed at the package terminals with an electrolytic capacitor
of 470μF or more.
BRIDGED AMPLIFIER APPLICATION
The LM4780 has two operational amplifiers internally, allow-
ing for a few different amplifier configurations. One of these
configurations is referred to as “bridged mode” and involves
driving the load differentially through the LM4780's outputs.
This configuration is shown in Figure 2. Bridged mode oper-
ation is different from the classical single-ended amplifier
configuration where one side of its load is connected to
ground.
A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Theoretically, four times the output power is possible
as compared to a single-ended amplifier under the same con-
ditions. This increase in attainable output power assumes that
the amplifier is not current limited or clipped.
A direct consequence of the increased power delivered to the
load by a bridge amplifier is an increase in internal power dis-
sipation. For each operational amplifier in a bridge configu-
ration, the internal power dissipation will increase by a factor
of two over the single ended dissipation. Thus, for an audio
power amplifier such as the LM4780, which has two opera-
tional amplifiers in one package, the package dissipation will
increase by a factor of four. To calculate the LM4780's max-
imum power dissipation point for a bridged load, multiply
Equation 2 by a factor of four.
This value of PDMAX can be used to calculate the correct size
heat sink for a bridged amplifier application. Since the internal
dissipation for a given power supply and load is increased by
using bridged-mode, the heatsink's θSA will have to decrease
accordingly as shown by Equation 4. Refer to the section,
Determining the Correct Heat Sink, for a more detailed
discussion of proper heat sinking for a given application.
PARALLEL AMPLIFIER APPLICATION
Parallel configuration is normally used when higher output
current is needed for driving lower impedance loads (i.e. 4Ω
or lower) to obtain higher output power levels. As shown in
Figure 3 , the parallel amplifier configuration consist of de-
signing the amplifiers in the IC to have identical gain, con-
necting the inputs in parallel and then connecting the outputs
in parallel through a small external output resistor. Any num-
ber of amplifiers can be connected in parallel to obtain the
needed output current or to divide the power dissipation
across multiple IC packages. Ideally, each amplifier shares
the output current equally. Due to slight differences in gain the
current sharing will not be equal among all channels. If current
is not shared equally among all channels then the power dis-
sipation will also not be equal among all channels. It is rec-
ommended that 0.1% tolerance resistors be used to set the
gain (Ri and Rf) for a minimal amount of difference in current
sharing.
When operating two or more amplifiers in parallel mode the
impedance seen by each amplifier is equal to the total load
impedance multiplied by the number of amplifiers driving the
load in parallel as shown by Equation 5 below:
RL(parallel) = RL(total) * Number of amplifiers (5)
Once the impedance seen by each amplifier in the parallel
configuration is known then Equation (2) can be used with this
calculated impedance to find the amount of power dissipation
for each amplifier. Total power dissipation (PDMAX) within an
IC package is found by adding up the power dissipation for
each amplifier in the IC package. Using the calculated
17 www.national.com
LM4780
PDMAX the correct heat sink size can be determined. Refer to
the section, Determining the Correct Heat Sink, for more
information and detailed discussion of proper heat sinking.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM4780 is a split supply ampli-
fier. But as shown in Figure 4, the LM4780 can also be used
in a single power supply configuration. This involves using
some external components to create a half-supply bias which
is used as the reference for the inputs and outputs. Thus, the
signal will swing around half-supply much like it swings
around ground in a split-supply application. Along with proper
circuit biasing, a few other considerations must be accounted
for to take advantage of all of the LM4780 functions, like the
mute function.
CLICKS AND POPS
In the typical application of the LM4780 as a split-supply audio
power amplifier, the IC exhibits excellent “click” and “pop”
performance when utilizing the mute mode. In addition, the
device employs Under-Voltage Protection, which eliminates
unwanted power-up and power-down transients. The basis
for these functions are a stable and constant half-supply po-
tential. In a split-supply application, ground is the stable half-
supply potential. But in a single-supply application, the half-
supply needs to charge up at the same rate as the supply rail,
VCC. This makes the task of attaining a clickless and popless
turn-on more challenging. Any uneven charging of the ampli-
fier inputs will result in output clicks and pops due to the
differential input topology of the LM4780.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the same.
Such a signal will be common-mode in nature, and will be
rejected by the LM4780. In Figure 4, the resistor RINP serves
to keep the inputs at the same potential by limiting the voltage
difference possible between the two nodes. This should sig-
nificantly reduce any type of turn-on pop, due to an uneven
charging of the amplifier inputs. This charging is based on a
specific application loading and thus, the system designer
may need to adjust these values for optimal performance.
As shown in Figure 4, the resistors labeled RBI help bias up
the LM4780 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of RBI, namely 10kΩ and 200kΩ. These re-
sistors bring up the inputs at the same rate resulting in a
popless turn-on. Adjusting these resistors values slightly may
reduce pops resulting from power supplies that ramp ex-
tremely quick or exhibit overshoot during system turn-on.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components is required to meet
the design targets of an application. The choice of external
component values that will affect gain and low frequency re-
sponse are discussed below.
The gain of each amplifier is set by resistors Rf and Ri for the
non-inverting configuration shown in Figure 1. The gain is
found by Equation 6 below:
AV = 1 + Rf / Ri (V/V) (6)
For best noise performance, lower values of resistors are
used. A value of 1kΩ is commonly used for Ri and then setting
the value of Rf for the desired gain. For the LM4780 the gain
should be set no lower than 10V/V and no higher than 50V/
V. Gain settings below 10V/V may experience instability and
using the LM4780 for gains higher than 50V/V will see an in-
crease in noise and THD.
The combination of Ri with Ci (see Figure 1) creates a high
pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
Equation 7 shown below:
fi = 1 / (2πRiCi) (Hz) (7)
If an input coupling capacitor is used to block DC from the
inputs as shown in Figure 5, there will be another high pass
filter created with the combination of CIN and RIN. When using
a input coupling capacitor RIN is needed to set the DC bias
point on the amplifier's input terminal. The resulting -3dB fre-
quency response due to the combination of CIN and RIN can
be found from Equation 8 shown below:
fIN = 1 / (2πRINCIN) (Hz) (8)
With large values of RIN oscillations may be observed on the
outputs when the inputs are left floating. Decreasing the value
of RIN or not letting the inputs float will remove the oscillations.
If the value of RIN is decreased then the value of CIN will need
to increase in order to maintain the same -3dB frequency re-
sponse.
HIGH PERFORMANCE CONSIDERATIONS
Using low cost electrolytic capacitors in the signal path such
as CIN and Ci (see Figures 1 - 5) will result in very good per-
formance. However, electrolytic capacitors are less linear
than other premium capacitors. Higher THD+N performance
may be obtained by using high quality polypropylene capac-
itors in the signal path. A more cost effective solution may be
the use of smaller value premium capacitors in parallel with
the larger electrolytic capacitors. This will maintain signal
quality in the upper audio band where any degradation is most
noticeable while also coupling in the signals in the lower audio
band for good bass response.
Distortion is introduced as the audio signal approaches the
lower -3dB point, determined as discussed in the section
above. By using larger values of capacitors such that the -3dB
point is well outside of the audio band will reduce this distor-
tion and improve THD+N performance.
Increasing the value of the large supply bypass capacitors will
improve burst power output. The larger the supply bypass
capacitors the higher the output pulse current without supply
droop increasing the peak output power. This will also in-
crease the headroom of the amplifier and reduce THD.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated
circuit designs to discrete amplifier designs. Discrete transis-
tor amps often “run out of gain” at high frequencies and
therefore have small bandwidths to noise as indicated below.
www.national.com 18
LM4780
20058699
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic dis-
tortion and improve frequency response. It is this additional
bandwidth that can lead to erroneous signal-to-noise mea-
surements if not considered during the measurement pro-
cess. In the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10in
bandwidth, (200kHz to 2MHz) can result in a 10dB theoretical
difference in the signal-to-noise ratio (white noise is propor-
tional to the square root of the bandwidth in a system).
In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
“weighting” filter (Note 16). A “weighting” filter alters the fre-
quency response in order to compensate for the average
human ear's sensitivity to the frequency spectra. The weight-
ing filters at the same time provide the bandwidth limiting as
discussed in the previous paragraph.
In addition to noise filtering, differing meter types give different
noise readings. Meter responses include:
1. RMS reading,
2. average responding,
3. peak reading, and
4. quasi peak reading.
Although theoretical noise analysis is derived using true RMS
based calculations, most actual measurements are taken with
ARM (Average Responding Meter) test equipment.
Typical signal-to-noise figures are listed for an A-weighted fil-
ter which is commonly used in the measurement of noise. The
shape of all weighting filters is similar, with the peak of the
curve usually occurring in the 3kHz–7kHz region.
LEAD INDUCTANCE
Power op amps are sensitive to inductance in the output
leads, particularly with heavy capacitive loading. Feedback to
the input should be taken directly from the output terminal,
minimizing common inductance with the load.
Lead inductance can also cause voltage surges on the sup-
plies. With long leads to the power supply, energy is stored in
the lead inductance when the output is shorted. This energy
can be dumped back into the supply bypass capacitors when
the short is removed. The magnitude of this transient is re-
duced by increasing the size of the bypass capacitor near the
IC. With at least a 20μF local bypass, these voltage surges
are important only if the lead length exceeds a couple feet
(>1μH lead inductance). Twisting together the supply and
ground leads minimizes the effect.
PHYSICAL IC MOUNTING CONSIDERATIONS
Mounting of the package to a heat sink must be done such
that there is sufficient pressure from the mounting screws to
insure good contact with the heat sink for efficient heat flow.
Over tightening the mounting screws will cause the package
to warp reducing contact area with the heat sink. Less contact
with the heat sink will increase the thermal resistance from
the package case to the heat sink (θCS) resulting in higher
operating die temperatures and possible unwanted thermal
shut down activation. Extreme over tightening of the mounting
screws will cause severe physical stress resulting in cracked
die and catastrophic IC failure. The recommended mounting
screw size is M3 with a maximum torque of 50 N-cm. Addi-
tionally, it is best to use washers under the screws to distribute
the force over a wider area or a screw with a wide flat head.
To further distribute the mounting force a solid mounting bar
in front of the package and secured in place with the two
mounting screws may be used. Other mounting options in-
clude a spring clip. If the package is secured with pressure on
the front of the package the maximum pressure on the molded
plastic should not exceed 150N/mm2.
Additionally, if the mounting screws are used to force the
package into correct alignment with the heat sink, package
stress will be increased. This increase in package stress will
result in reduced contact area with the heat sink increasing
die operating temperature and possible catastrophic IC fail-
ure.
LAYOUT, GROUND LOOPS AND STABILITY
The LM4780 is designed to be stable when operated at a
closed-loop gain of 10 or greater, but as with any other high-
current amplifier, the LM4780 can be made to oscillate under
certain conditions. These oscillations usually involve printed
circuit board layout or output/input coupling issues.
When designing a layout, it is important to return the load
ground, the output compensation ground, and the low level
(feedback and input) grounds to the circuit board common
ground point through separate paths. Otherwise, large cur-
rents flowing along a ground conductor will generate voltages
on the conductor which can effectively act as signals at the
input, resulting in high frequency oscillation or excessive dis-
tortion. It is advisable to keep the output compensation com-
ponents and the 0.1μF supply decoupling capacitors as close
as possible to the LM4780 to reduce the effects of PCB trace
resistance and inductance. For the same reason, the ground
return paths should be as short as possible.
In general, with fast, high-current circuitry, all sorts of prob-
lems can arise from improper grounding which again can be
avoided by returning all grounds separately to a common
point. Without isolating the ground signals and returning the
grounds to a common point, ground loops may occur.
“Ground Loop” is the term used to describe situations occur-
ring in ground systems where a difference in potential exists
between two ground points. Ideally a ground is a ground, but
unfortunately, in order for this to be true, ground conductors
with zero resistance are necessary. Since real world ground
leads possess finite resistance, currents running through
them will cause finite voltage drops to exist. If two ground re-
turn lines tie into the same path at different points there will
be a voltage drop between them. The first figure below shows
a common ground example where the positive input ground
and the load ground are returned to the supply ground point
via the same wire. The addition of the finite wire resistance,
R2, results in a voltage difference between the two points as
shown below.
19 www.national.com
LM4780
20058698
The load current IL will be much larger than input bias current
II, thus V1 will follow the output voltage directly, i.e. in phase.
Therefore the voltage appearing at the non-inverting input is
effectively positive feedback and the circuit may oscillate. If
there was only one device to worry about then the values of
R1 and R2 would probably be small enough to be ignored;
however, several devices normally comprise a total system.
Any ground return of a separate device, whose output is in
phase, can feedback in a similar manner and cause instabil-
ities. Out of phase ground loops also are troublesome, caus-
ing unexpected gain and phase errors.
The solution to most ground loop problems is to always use
a single-point ground system, although this is sometimes im-
practical. The third figure above is an example of a single-
point ground system.
The single-point ground concept should be applied rigorously
to all components and all circuits when possible. Violations of
single-point grounding are most common among printed cir-
cuit board designs, since the circuit is surrounded by large
ground areas which invite the temptation to run a device to
the closest ground spot. As a final rule, make all ground re-
turns low resistance and low inductance by using large wire
and wide traces.
Occasionally, current in the output leads (which function as
antennas) can be coupled through the air to the amplifier in-
put, resulting in high-frequency oscillation. This normally hap-
pens when the source impedance is high or the input leads
are long. The problem can be eliminated by placing a small
capacitor, CC, (on the order of 50pF to 500pF) across the
LM4780 input terminals. Refer to the External Components
Description section relating to component interaction with
Cf.
REACTIVE LOADING
It is hard for most power amplifiers to drive highly capacitive
loads very effectively and normally results in oscillations or
ringing on the square wave response. If the output of the
LM4780 is connected directly to a capacitor with no series
resistance, the square wave response will exhibit ringing if the
capacitance is greater than about 0.2μF. If highly capacitive
loads are expected due to long speaker cables, a method
commonly employed to protect amplifiers from low
impedances at high frequencies is to couple to the load
through a 10Ω resistor in parallel with a 0.7μH inductor. The
inductor-resistor combination as shown in the Figure 5 iso-
lates the feedback amplifier from the load by providing high
output impedance at high frequencies thus allowing the 10Ω
resistor to decouple the capacitive load and reduce the Q of
the series resonant circuit. The LR combination also provides
low output impedance at low frequencies thus shorting out the
10Ω resistor and allowing the amplifier to drive the series RC
load (large capacitive load due to long speaker cables) di-
rectly.
INVERTING AMPLIFIER APPLICATION
The inverting amplifier configuration may be used instead of
the more common non-inverting amplifier configuration
shown in Figure 1. The inverting amplifier can have better
THD+N performance and eliminates the need for a large ca-
pacitor (Ci) reducing cost and space requirements. The val-
ues show in Figure 6 are only one example of an amplifier
with a gain of 20V/V (Gain = -Rf/Ri). For different resistor val-
ues, the value of RB should be eqaul to the parallel combina-
tion of Rf and Ri.
www.national.com 20
LM4780
20058621
FIGURE 6. Inverting Amplifier Application Circuit
21 www.national.com
LM4780
200586f3
FIGURE 7. Reference PCB Schematic
www.national.com 22
LM4780
LM4780 REFERENCE BOARD ARTWORK
200586d9
Composite Layer
200586d8
Silk Layer
200586d7
Top Layer
200586e0
Bottom Layer
23 www.national.com
LM4780
Bill Of Materials For Reference Pcb
Symbol Value Tolerance Type/Description Comment
RIN1, RIN2 15kΩ5% 1/4 Watt
RB1, RB2 1kΩ1% 1/4 Watt
RF1, RF2 20kΩ1% 1/4 Watt
Ri1, Ri2 1kΩ1% 1/4 Watt
RSN1, RSN2,2.7Ω 5% 1/4 Watt
RG2.7Ω 5% 1/4 Watt
RM10kΩ5% 1/4 Watt
CIN1, CIN2 1µF 10% Metallized Polyester Film
Ci1, Ci2, 68µF 20% Electrolytic Radial / 50V
CSN1, CSN2 0.1µF 20% Monolithic Ceramic
CN1, CN2 15pF 20% Monolithic Ceramic
CS1, CS2, CS3 0.1µF 20% Monolithic Ceramic
CS4, CS5, CS6 10µF 20% Electrolytic Radial / 50V
CS7, CS8 1,000µF 20% Electrolytic Radial / 50V
S1 SPDT (on-on) Switch
J1, J2
Non-Switched PC Mount RCA
Jack
J4, J7, J8 PCB Banana Jack - BLACK
J3, J5, J6, J9 PCB Banana Jack - RED
U1
27 lead TO-220 Power Socket
with push lever release or LM4780
IC
www.national.com 24
LM4780
Physical Dimensions inches (millimeters) unless otherwise noted
Non-Isolated TO-220 27-Lead Package
Order Number LM4780TA
NS Package Number TA27A
25 www.national.com
LM4780
Notes
LM4780 Overture® Audio Power Amplifier Series
Stereo 60W, Mono 120W Audio Power Amplifier with Mute
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