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LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

LM1771 Low-Voltage Synchronous Buck Controller With Precision Enable and No

External Compensation

1 Features 3 Description

The LM1771 device is an efficient synchronous buck

1• Input Voltage Range of 2.8 V to 5.5 V switching controller with a precision enable requiring

• 0.8-V Reference Voltage no external compensation. The constant ON-time

• Precision Enable control scheme provides a simple design free of

compensation components, allowing minimal

• No Compensation Required component count and board space. The precision

• Constant Frequency Across Input Range enable pin allows flexibility in sequencing multiple

• Low Quiescent Current of 400 µA rails and setting UVLO. The LM1771 also

incorporates a unique input feedforward to maintain a

• Internal Soft-Start constant frequency independent of the input voltage.

• Short Circuit Protection The LM1771 is optimized for a low-voltage input

• 6-Pin WSON Package and 8-Pin VSSOP Package range of 2.8 V to 5.5 V and can provide an adjustable

output as low as 0.8 V. Driving an external high-side

2 Applications PFET and low-side NFET, it can provide efficiencies

as high as 95%.

• Simple-to-Design, High-Efficiency, Step-Down

Switching Regulators Three versions of the LM1771 are available

depending on the switching frequency desired for the

• FPGAs, DSPs, and ASIC Power Supplies application. Nominal switching frequencies are in the

• Set-Top Boxes range of 100 kHz to 1000 kHz.

• Cable Modems

• Printers Device Information(1)

PART NUMBER PACKAGE BODY SIZE (NOM)

• Digital Video Recorders WSON (6) 3.00 mm × 3.00 mm

• Servers LM1771 VSSOP (8) 3.00 mm × 3.00 mm

• Graphic Cards (1) For all available packages, see the orderable addendum at

the end of the data sheet.

Typical Application Circuit

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

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Table of Contents

1 Features.................................................................. 18 Application and Implementation ........................ 13

8.1 Application Information............................................ 13

2 Applications ........................................................... 18.2 Typical Applications ................................................ 13

3 Description............................................................. 19 Power Supply Recommendations...................... 20

4 Revision History..................................................... 210 Layout................................................................... 21

5 Pin Configuration and Functions......................... 310.1 Layout Guidelines ................................................. 21

6 Specifications......................................................... 410.2 Layout Examples................................................... 21

6.1 Absolute Maximum Ratings ..................................... 410.3 Thermal Considerations........................................ 22

6.2 ESD Ratings.............................................................. 411 Device and Documentation Support................. 23

6.3 Recommended Operating Conditions....................... 411.1 Device Support .................................................... 23

6.4 Thermal Information.................................................. 411.2 Documentation Support ........................................ 23

6.5 Electrical Characteristics........................................... 511.3 Community Resources.......................................... 23

6.6 Typical Characteristics.............................................. 611.4 Trademarks........................................................... 23

7 Detailed Description.............................................. 911.5 Electrostatic Discharge Caution............................ 23

7.1 Overview................................................................... 911.6 Glossary................................................................ 23

7.2 Functional Block Diagram......................................... 912 Mechanical, Packaging, and Orderable

7.3 Feature Description................................................. 10 Information........................................................... 23

7.4 Device Functional Modes........................................ 12

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision C (April 2013) to Revision D Page

• Added ESD Ratings table, Feature Description section, Device Functional Modes,Application and Implementation

section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and

Mechanical, Packaging, and Orderable Information section. ................................................................................................ 1

Changes from Revision B (April 2013) to Revision C Page

• Changed layout of National Data Sheet to TI format ........................................................................................................... 22

Product Folder Links: LM1771

GND

VIN

LGHG

GND

4 5

DAP

GND

HG LG

VIN

LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

5 Pin Configuration and Functions

NGG Package

6-Pin WSON

Top View

DGK Package

8-Pin VSSOP

Top View

Pin Functions

PIN TYPE(1) DESCRIPTION

NAME WSON VSSOP

Die Attach Pad is internally connected to GND, but it cannot be used as the

DAP — — GND primary GND connection

EN 6 8 I Enable Pin

FB 1 1 A Feedback Pin

GND 2 2, 3 GND Ground

HG 3 4 AO PFET Gate Drive

LG 4 5 AO NFET Gate Drive

VIN 5 6, 7 PWR Input Supply

(1) I = Input, A = Analog, GND = Ground, AO = Analog output, PWR = Power

Product Folder Links: LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

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6 Specifications

6.1 Absolute Maximum Ratings

See (1) (2)

MIN MAX UNIT

VIN –0.3 6 V

EN, FB, HG, LG –0.3 VIN V

Junction temperature 150 °C

Lead temperature Soldering, 10 sec 260 °C

Storage temperature, Tstg –65 150 °C

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

6.2 ESD Ratings VALUE UNIT

V(ESD) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1) ±2000 V

(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

6.3 Recommended Operating Conditions MIN MAX UNIT

VIN to GND 2.8 5.5 V

Junction temperature, TJ−40 125 °C

6.4 Thermal Information LM1771

THERMAL METRIC(1) NGG (WSON) DGK (VSSOP) UNIT

6 PINS 8 PINS

RθJA Junction-to-ambient thermal resistance 52.8 169.2 °C/W

RθJC(top) Junction-to-case (top) thermal resistance 51.4 59.7 °C/W

RθJB Junction-to-board thermal resistance 27.2 89.3 °C/W

ψJT Junction-to-top characterization parameter 0.7 7.0 °C/W

ψJB Junction-to-board characterization parameter 27.3 87.9 °C/W

RθJC(bot) Junction-to-case (bottom) thermal resistance 7.4 N/A °C/W

(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application

report, SPRA953.

Product Folder Links: LM1771

LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

6.5 Electrical Characteristics

Specifications are for TJ= 25°C. All maximum and minimum limits apply over the full junction temperature range (−40°C to

+125°C), unless otherwise specified. Minimum and maximum limits are ensured through test, design or statistical correlation.

Typical values represent the most likely parametric norm at TJ= 25°C and are provided for reference purposes only. Unless

otherwise specified, VIN = 3.3 V.

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

VFB Feedback pin voltage 0.782 0.8 0.818 V

IQQuiescent current VFB = 0.9 V 400 700 µA

LM1771S - (500 ns) 0.4 0.5 0.6

TON Switch ON-time LM1771T - (1000 ns) 0.8 1 1.2 µs

LM1771U - (2000 ns) 1.6 2 2.4

LM1771S - (500 ns) 150 250

TOFF_MIN Minimum OFF-time LM1771T - (1000 ns) 135 225 ns

LM1771U - (2000 ns) 120 220

TDGate drive dead-time 70 ns

VIH_EN EN pin rising threshold 1.15 1.2 1.25 V

VEN_HYS EN pin hysteresis 50 200 mV

IFB Feedback pin bias current VFB = 0.9 V 50 nA

VUVLO Undervoltage lockout VIN Rising Edge 2.65 2.8 V

VUVLO_HYS Undervoltage lockout hysteresis 50 mV

VSC_TH Feedback pin short circuit latch 0.42 0.55 0.65 V

threshold

RDS(ON) 1 HG FET driver pullup ON-resistance IHG = 20 mA 4 Ω

RDS(ON) 2 HG FET driver pulldown ON-resistance IHG = 20 mA 6 Ω

RDS(ON) 3 LG FET driver pullup ON-resistance ILG = 20 mA 4 Ω

RDS(ON) 4 LG FET driver pulldown ON-resistance ILG = 20 mA 6 Ω

Product Folder Links: LM1771

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6.6 Typical Characteristics

All curves taken at VIN = 3.3 V with configuration in typical application circuit shown in Typical Applications. TJ= 25°C, unless

otherwise specified.

Figure 1. TON vs VIN (LM1771S) Figure 2. TON vs VIN (LM1771T)

Figure 3. TON vs VIN (LM1771U) Figure 4. TON vs Temperature (LM1771S)

Figure 5. TON vs Temperature (LM1771T) Figure 6. TON vs Temperature (LM1771U)

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Typical Characteristics (continued)

All curves taken at VIN = 3.3 V with configuration in typical application circuit shown in Typical Applications. TJ= 25°C, unless

otherwise specified.

Figure 7. TOFF vs Temperature (LM1771S) Figure 8. TOFF vs Temperature (LM1771T)

Figure 9. TOFF vs Temperature (LM1771U) Figure 10. Feedback Voltage vs Temperature

Figure 11. VEN Threshold vs Temperature Figure 12. Short-Circuit Threshold vs Temperature

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Typical Characteristics (continued)

All curves taken at VIN = 3.3 V with configuration in typical application circuit shown in Typical Applications. TJ= 25°C, unless

otherwise specified.

Figure 13. Quiescent Current vs Temperature Figure 14. Dead-Time vs Temperature

VIN = 5 V VOUT = 1.8 V FSW = 545 kHz VIN = 5 V VOUT = 2.5 V FSW = 379 kHz

Figure 15. Efficiency vs IOUT (LM1771T) Figure 16. Efficiency vs IOUT (LM1771U)

VIN = 5 V VOUT = 3.3 V FSW = 500 kHz VIN = 5 V VOUT = 1.2 V FSW = 727 kHz

Figure 17. Efficiency vs IOUT (LM1771U) Figure 18. Efficiency vs IOUT (LM1771S)

Product Folder Links: LM1771

VIN

ON TIMER

High-Side

Driver

0.8V

Vin

LM1771

REGULATION

COMPARATOR

OFF TIMER

GND

Level Shift

and

Shoot

Through

Protection Low-Side

Driver

SHORT

CIRCUIT

PROTECTION

0.55 V

/Soft-Start

UVLO

UVLO SD

1.2 V

ENABLE

COMPARATOR

LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

7 Detailed Description

7.1 Overview

The LM1771 synchronous buck controller has a control scheme that is referred to as adaptive ON-time control.

This topology relies on a fixed switch ON-time to regulate the output voltage. This ON-time is internally set by

EEPROM and is available with three different set-points to allow for different frequency options. The LM1771

automatically adjusts the ON-time during operation inversely with the input voltage (VIN) to maintain a constant

frequency. Therefore the switching frequency during continuous conduction mode is independent of the inductor

and capacitor size unlike hysteretic switchers.

At the beginning of the cycle, the LM1771 turns on the high-side PFET for a fixed duration. This ON-time is

predetermined (internally set by EEPROM and adjusted by VIN) and the switch does not turn off until the timer

has completed its period. The PFET then turns off for a minimum predetermined time period. This minimum TOFF

of 150 ns is internally set and cannot be adjusted. This is to prevent false triggering from occurring on the

comparator due to noise from the SW node transition. After the minimum TOFF period has expired, the PFET

remains off until the comparator trip-point has been reached. Upon passing this trip-point (set at 0.8 V at the

feedback pin), the PFET turns back on and the process repeats, thus regulating the output.

The NFET control is complementary to the PFET control with the exception of a short dead-time to prevent

shoot-through from occurring.

7.2 Functional Block Diagram

Product Folder Links: LM1771

VOUT

fSW = D

VOUT

F = VIN x TON

TP = fSW

VOUT

D = VIN

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

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7.3 Feature Description

7.3.1 Timing Opinion

Three versions of the LM1771 are available each with a predetermined TON set internally by EEPROM. This TON

setting determines the switching frequency for the application. Derivation and calculation of the dependence of

the switching frequency on VIN and TON is shown in Equation 1 through Equation 6.

In a PWM buck switcher, Equation 1,Equation 2, and Equation 3 can be manipulated to obtain the switching

frequency. Equation 1 shows the standard duty-cycle equation given by the volts-seconds balance on the

inductor with Equation 2 and Equation 3 defining standard relationships:

(1)

TON = D x TP(2)

(3)

Using these equations and solving for duty-cycle for Equation 4:

D = fSW x TON (4)

Frequency can now be expressed in Equation 5:

(5)

Or simply written as Equation 6:

where

•α= VIN x TON (6)

To maintain a set frequency in an application, αis always held constant by varying TON inversely with VIN. The

three versions of the LM1771 are identified by the ON-times at a VIN of 3.3 V for consistency. For clarification,

see Table 1.

Table 1. Timing for All Variants

Product ID TON at 3.3 V α(V µs)

LM1771S 0.5 µs 1.65

LM1771T 1.0 µs 3.3

LM1771U 2.0 µs 6.6

The variation of TON versus VIN can also be expressed graphically. These graphs can be found in the Typical

Characteristics of the data sheet.

With αbeing a constant regardless of the version of the LM1771 used, Equation 6 shows that the only

dependent variable remaining is VOUT. Because VOUT is a constant in any application, the frequency also remains

constant. The switching frequency at which the application runs depends upon the VOUT desired and the LM1771

version chosen. For any VOUT, three frequency options (LM1771 versions) can be selected. This can be seen

Table 2. The recommended frequency range of operation is 100 kHz to 1000 kHz.

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Table 2. Frequency vs VOUT for Variants

TIMING OPTIONS (1)

VOUT 500 ns 1000 ns 2000 ns

0.8 485 242 121

1 606 303 152

1.2 727 364 182

1.5 909 455 227

1.8 1091 545 273

2.5 1515 758 379

3.3 2000 1000 500

(1) Switching Frequency (kHz) of LM1771 based on output voltage and

timing option.

7.3.2 Short-Circuit Protection

The LM1771 has an internal short-circuit comparator that constantly monitors the feedback node (except during

soft-start). If the feedback voltage drops below 0.55 V (equivalent to the output voltage dropping below 68% of

nominal), the comparator trips causing the part to latch off. The LM1771 does not resume switching until the

input voltage is taken below the UVLO threshold and then brought back into its normal operating range, or the

part is disabled then re-enabled through the enable pin. The purpose of this function is to prevent a severe short

circuit from causing damage to the application. Due to the fast transient response of the LM1771 a severe short

on the output causing the feedback to drop would only occur if the load applied had an effective resistance that

approaches the PMOS RDS(ON).

7.3.3 Precision Enable

The LM1771 features a precision enable circuit. If the voltage on the EN pin is 1.2 V or greater, the part is

enabled and switching occurs. If the enable voltage falls below 1.2 V, the part is placed into a shutdown state

and the drivers is tri-stated. This allows the LM1771 to be easily sequenced using a resistive divider from the

output of another regulator, or the working input voltage range of the LM1771 to be set using a resistive divider

on VIN. There is no internal pullup connected to the EN pin, so an external signal is required to initiate switching.

NOTE

When power is first applied to the LM1771, there is a slight delay before the enable

comparator is functional. During this delay, typically on the order of 400 µs, the part is

disabled regardless of the voltage on the EN pin. The falling enable threshold features 50

mV of hysteresis

7.3.4 Soft-Start

To limit in-rush current and allow for a controlled start-up the LM1771 incorporates an internal soft-start scheme.

Every time the enable voltage rises above 1.2 V while VIN is greater than the UVLO threshold, the LM1771 goes

through an adaptive soft-start that limits the ON-time and expands the minimum OFF-time. In addition the part

only activates the PMOS allowing a discontinuous mode of operation enabling a prebiased start-up. The time

spent in soft-start depends on the load applied to the output, but is usually close to a set time that is dependent

on the timing option. The approximate soft-start time can be seen in Table 3 for each timing option.

Table 3. Soft-Start Timing for All Variants

PRODUCT ID TIMING TSS

LM1771S 0.5 µs 1 ms

LM1771T 1.0 µs 1.2 ms

LM1771U 2.0 µs 1.8 ms

Product Folder Links: LM1771

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Therefore, if the input supply is extremely slow rising such that at the end of soft-start the input voltage is still

near the UVLO threshold, a timing option must be chosen to ensure that maximum duty-cycle permits the output

to meet the minimum condition. TI recommends using the 2000-ns option (LM1771U) in conditions where the

output voltage is 2.5 V or greater to avoid false latch-offs when there is concern regarding the input supply slew

rate.

NOTE

As soon as soft-start terminates the short-circuit protection is enabled. This means that if

the output voltage does not reach at least 68% of its final value the part latches off.

In some situations, the internal soft-start routine can create a slight overshoot on the output voltage. If this must

be avoided, the use of a feedforward capacitor as detailed in the Feedforward Capacitor section.

7.3.5 Jitter

The LM1771 uses an adaptive ON-time control scheme that relies on the output voltage ripple to provide a

consistent switching frequency. Under certain conditions, excessive noise can couple onto the feedback pin

causing the switch node to appear to have a slight amount of jitter. This is not indicative of an unstable design.

The output voltage still regulates to the exact same value. Careful component selection and layout must minimize

any external influence.

In addition to any external noise that can add to the jitter seen on the switch node, the LM1771 always has a

slight amount of switch jitter. This is because the LM1771 makes a small alteration in the reference voltage every

128 cycles to improve its accuracy and long-term performance. This has the effect of causing a change in the

switching frequency at that instant. When viewed on an oscilloscope this can be seen as a jitter in the switch

node. The change in feedback voltage or output voltage, however, is almost indistinguishable.

7.4 Device Functional Modes

The LM1771 has essentially one normal operational mode: in normal operation, the part operates in full

synchronous mode to regulate the FB input to 0.8 V (typical) after soft-start period is over. The EN pin allows the

user to shut down the part. When the part is enabled, the IC enters soft-start for 1 ms to 1.8 ms depending on

the variant of the IC. See Soft-Start for more detail on the soft-start pattern.

Product Folder Links: LM1771

VOUT

D = VIN

VIN

LM1771 VOUT

GND

COUT

CIN

RFB1

RSNS

EN EN

RFB2

CFF

LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers must

validate and test their design implementation to confirm system functionality.

8.1 Application Information

Design Guide details the design procedure to follow for the typical application. Because of the nature of constant

ON-time controller, a certain degree of iteration might be necessary on the sizing of some key components to

achieve optimal response, such as the inductor L1.

8.2 Typical Applications

8.2.1 LM1771 Typical Application

Figure 19. Example Circuit Schematic

8.2.1.1 Design Requirements

The main parameters needed to successfully design an application based on the LM1171 are typical of buck

DC–DC converters. The input and output voltage must be known as well as the worst-case input voltage

operating conditions. The maximum output current helps the designer size the inductor appropriately. In addition,

the designer needs to be aware of the requirement on the output capacitor to achieve stable operation. See

Output Capacitor for details on the output capacitor requirements.

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Design Guide

As with any DC–DC converter, numerous trade-offs are possible to optimize the design for efficiency, size, or

performance. These are taken into account and highlighted throughout this discussion.

Equation 7 calculates for any buck converter is duty-cycle. Ignoring conduction losses associated with the FETs

and parasitic resistances it can be approximated by:

(7)

A more accurate calculation for duty-cycle can be used that takes into account the voltage drops across the

FETs. Equation 8 determines the slight load dependency on switch frequency if needed. Otherwise the simplified

equation works well for component calculation.

Product Folder Links: LM1771

L = (VIN ± VOUT) x D

0.3 x fSW x IOUT

'IL = (VIN ± VOUT) x D

L x fSW

VOUT + VDS_NMOS

D = VIN + VDS_NMOS + VDS_PMOS

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

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Typical Applications (continued)

(8)

8.2.1.2.1.1 Frequency Selection

The LM1771 is available with three preset timing options that select the ON-time and hence determine the

switching frequency of the application. Increasing the switching frequency has the effect of reducing the inductor

size needed for the application while requiring a slight trade-off in efficiency. Table 4 shows the same frequency

table as shown earlier (Table 2) with the exception that the recommended timing option for each VOUT is

highlighted. TI does not recommend using a high switching frequency with VOUT equal to or greater than 2.5 V

due to the maximum duty-cycle limitations of the device coupled with the internal start-up.

Table 4. Recommended Switching Frequency vs VOUT

TIMING OPTIONS (1)

VOUT 500 ns 1000 ns 2000 ns

0.8 485 242 -

1 606 303 -

1.2 727 364 -

1.5 909 455 2237

1.8 - 545 273

2.5 - - 379

3.3 - - 500

(1) Recommended switching frequency (kHz) based on output voltage

and timing option.

8.2.1.2.1.2 Inductor Selection

The inductor selection is an iterative process likely requiring several passes before settling on a final value. The

reason for this is because it influences the amount of ripple seen at the output, a critical component to ensure

general stability of an adaptive ON-time circuit. For the first pass at inductor selection the value can be obtained

by targeting a maximum peak-to-peak ripple current equal to 30% of the maximum load current. The inductor

current ripple (ΔIL) can be calculated by Equation 9:

(9)

Therefore, L can be initially set by applying the 30% rule in Equation 10:

(10)

The other features of the inductor that can be selected besides inductance value are saturation current and core

material. Because the LM1771 does not have a current limit, TI recommends having a saturation current higher

than the maximum output current to handle any ripple or momentary overcurrent events. The core material also

influences the saturation characteristics as ferrite materials have a hard saturation curve and take care that they

never saturate during normal use. A shielded inductor or low-profile unshielded inductor is recommended to

reduce EMI. This also helps prevent any spurious noise from picking up on the feedback node resulting in

unexpected tripping of the feedback comparator.

8.2.1.2.1.3 Output Capacitor

One of the most important components to select with the LM1771 is the output capacitor. This is because its size

and ESR have a direct effect on the stability of the loop. A constant ON-time control scheme works by sensing

the output voltage ripple and switching the FETs appropriately. The output voltage ripple on a buck converter can

be approximated by stating that the AC inductor ripple flows entirely into the output capacitor and is created by

the ESR of the capacitor. This can be expressed in Equation 11:

ΔVOUT =ΔILx RESR (11)

Product Folder Links: LM1771

VIN

LM1771 VOUT

GND

COUT

CIN

RFB1

RFB2

CFF

L1RSNS

EN EN

ICIN_RMS = IOUT

D1 - D + 'IL2

12 x IOUT2

H = VFB

VOUT

0.8V

VOUT

LM1771

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SNVS446D –JUNE 2006–REVISED JANUARY 2016

To ensure stability, two constraints need to be met. The first is that there is sufficient ESR to create enough

voltage ripple at the feedback pin. TI recommends having at least 10 mV of ripple seen at the feedback pin. This

can be calculated by multiplying the output voltage ripple by the gain seen through the feedback resistors. This

gain, H, can be calculated in Equation 12:

(12)

If the output voltage is fairly high, causing significant attenuation through the feedback resistors, a feedforward

capacitor can be used. This is actually recommended for most circuits as it improves performance. See the

Feedforward Capacitor section for more details.

The second criteria is to ensure that there is sufficient ripple at the output that is in-phase with the switch. The

problem exists that there is actually ripple caused by the capacitor charging and discharging, not only the ESR

ripple. Because these are effectively out of phase, problems can exist. To avoid this issue, TI recommends that

the ratio of the two ripples (β) is always greater than 5. To calculate the minimum ESR value needed,

Equation 13 can be used:

(13)

In general, the best capacitors to use are chemistries that have a known and consistent ESR across the entire

operating temperature range. Tantalum capacitors or similar chemistries such as Niobium Oxide perform well

along with certain families of Aluminum Electrolytics. Small value POSCAPs and SP CAPs also work as they

have sufficient ESR. When used in conjunction with a low-value inductor it is possible to have an extremely

stable design. The only capacitors that require modification to the circuit are ceramic capacitors. Ceramic

capacitors cause problems meeting both criteria because they have low ESR and low capacitance. Therefore, if

they are to be used, an external ESR resistor (RSNS) must be added. This can be seen in Figure 20.

Figure 20. LM1771 With an External ESR Resistor (RSNS)

This circuit uses an additional resistor in series with the inductor to add ripple at the output. It is placed in this

location and used in combination with the feedforward capacitor (CFF) to provide ripple to the feedback pin,

without adding ripple or a DC offset to the output. The benefit of using a ceramic capacitor is still obtained with

this technique. Because the addition of the resistor results in power loss, this circuit implementation is only

recommended for low currents (2 A and below). The power loss and rating of the resistor must be taken into

account when selecting this component.

This circuit implementation using the feedforward capacitor begins to experience limitations when the output

voltage is small. Previously the circuit relied on the CFF for all the ripple at the feedback node by assuming that

the resistor divider was negligible. As VOUT decreases this can not be assumed. The resistor divider contributes a

larger amount of ripple which is problematic as it is also out of phase. Therefore the resistor location must be

changed to be in series with the output capacitor. This can be viewed as adding an effective ESR to the output

capacitor. This can be seen in Figure 21.

Product Folder Links: LM1771

ICIN_RMS = IOUT x D(1 - D)

ICIN_RMS = IOUT

D1 - D + 'IL2

12 x IOUT2

VIN

LM1771 VOUT

GND

COUT

CIN

RFB1

RSNS

EN EN

RFB2

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

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Figure 21. LM1771 With ESR Resistor (RSNS) in Series With Output Capacitor

8.2.1.2.1.4 Feedforward Capacitor

The feedforward capacitor is used across the top feedback resistor to provide a lower impedance path for the

high-frequency ripple without degrading the DC accuracy. Typically the value for this capacitor must be small

enough to prevent load transient errors because of the discharging time, but large enough to prevent attenuation

of the ripple voltage. In general a small ceramic capacitor in the range of 1 nF to 10 nF is sufficient.

If CFF is used then it can be assumed that the ripple voltage seen at the feedback pin is the same as the ripple

voltage at the output. The attenuation factor H no longer needs to be used. However, in these conditions, TI

recommends having a minimum of 20-mV ripple at the feedback pin. The use of a CFF capacitor is recommended

as it improves the regulation and stability of the design. However, its benefit is diminished as VOUT starts

approaching VREF , therefore it is not needed in this situation.

8.2.1.2.1.5 Input Capacitor

The dominating factor that usually sets the size of an input capacitor is the current handling ability. This is usually

determined by the package size and ESR of the capacitor. If these two criteria are met then there usually must

be enough capacitance to prevent impedance interactions with the source. In general, TI recommends using a

ceramic capacitor for the input as they provide a low impedance and small footprint.

NOTE

Use a good dielectric for the ceramic capacitor such as X5R or X7R. These provide better

over temperature performance and also minimize the DC voltage derating that occurs on

Y5V capacitors.

To calculate the input capacitor RMS current, Equation 14 can be used:

(14)

which can be approximated by, Equation 15

(15)

8.2.1.2.2 MOSFET Selection

The two FETs used in the LM1771 requires attention to selection of parameters to ensure optimal performance of

the power supply. The high-side FET must be a PFET and the low-side an NFET. These can be integrated in

one package or as two separate packages.

Product Folder Links: LM1771

RFB1 = RFB2 VFB

VOUT - 1

LM1771

www.ti.com

SNVS446D –JUNE 2006–REVISED JANUARY 2016

8.2.1.2.2.1 VDS Voltage Rating

The first selection criteria is to select FETs that have sufficient VDS voltage ratings to handle the maximum

voltage seen at the input plus any transient spikes that can occur from parasitic ringing. In general most FETs

available for this application have ratings from 8 V to 20 V. If a larger voltage rating is used then the performance

is most likely be degraded because of higher gate capacitance.

8.2.1.2.2.2 RDSON

The RDS(ON) specification is important as it determines several attributes of the FET and the overall power supply.

The first is that it sets the maximum current of the FET for a given package. A lower RDS(ON) permits a higher

allowable current and reduce conduction losses, however, it increases the gate capacitance and the switching

losses.

8.2.1.2.2.3 Gate Drive

The next step is to ensure that the FETs are capable of switching at the low Vin supplies used by the LM1771.

The FET must have the Rdson specified at either 1.8 V or 2.5 V to ensure that it can switch effectively as soon

as the LM1771 starts up.

8.2.1.2.2.4 Gate Charge

Because the LM1771 uses a fixed dead-time scheme to prevent cross conduction, the FET transitions must

occur in this time. The rise and fall time of the FETs gate can be influenced by several factors including the gate

capacitance. Therefore the total gate charge of both FETs must be limited to less than 20 nC at 4.5 V VGS. The

lower the number the faster the FETs must switch and the better the efficiency.

8.2.1.2.2.5 Rise and Fall Times

A better indication of the actual switching times of the FETs can be found in their Electrical Characteristics table.

The rise and fall time must be specified and selected to be at a minimum. This helps improve efficiency and

ensuring that shoot-through does not occur.

8.2.1.2.2.6 Gate Charge Ratio

Another consideration in selecting the FETs is to pay attention to the Qgd / Qgs ratio. The reason for this is that

proper selection can prevent spurious turnon. If we look at the NFET for example, when the FET is turning off,

the gate signal pulls to ground. Conversely the PFET is turning on, causing the SW node to rise towards VIN. The

gate-to-drain capacitance of the NFET couples the SW node to the gate and causes it to rise. If this voltage is

excessive, then it could weakly turn on the low-side FET causing an efficiency loss. However, this coupling is

mitigated by having a large gate to source capacitance of the FET, which helps to hold the gate voltage down.

Ideally, a very low Qgd / Qgs would be ideal, but in practice it is common to find the number around 1. As a

general rule, the lower the ratio, the better.

If the above selection criteria have been met it is useful to generate a figure of merit to allow comparison

between the FETs. One such method is to multiply the RDS(ON) of the FET by the total gate charge. This allows

an easy comparison of the different FETs available. Once again, the lower the product, the better.

8.2.1.2.2.7 Feedback Resistors

The feedback resistors are used to scale the output voltage to the internal reference value such that the loop can

be regulated. The feedback resistors must not be made arbitrarily large as this creates a high impedance node at

the feedback pin that is more susceptible to noise. A combined value of 50 kΩfor the two resistors is adequate.

To calculate the resistor values use Equation 16. Typically, the low-side resistor is initially set to a predetermined

value such as 10 kΩ.

where

• VFB is the internal reference voltage that can be found in the Electrical Characteristics table or approximated

by 0.8 V (16)

Product Folder Links: LM1771

VOUT

VOUT_ACTUAL

VOUT_SET

OUT

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

www.ti.com

The output voltage value can be set in a precise manner by taking into account the fact that the reference

voltage is regulating the bottom of the output ripple as opposed to the average value. This relationship is shown

in Figure 22.

Figure 22. Output Voltage Waveform

The average output voltage (VOUT_ACTUAL) is higher than the output voltage (VOUT_SET) that was calculated by the

earlier equation by exactly half the output voltage ripple. The output voltage that is targeted for regulation may

then be appended according to the voltage ripple. This can be seen in Equation 17:

VOUT_ACTUAL= VOUT_SET + ½ΔVOUT = VOUT_SET + ½ΔILx RESR (17)

8.2.1.2.3 Efficiency Calculations

One of the most important parameters to calculate during the design stage is the expected efficiency of the

system. This can help determine optimal FET selection and can be used to calculate expected temperature rise

of the individual components.

8.2.1.2.3.1 Quiescent Current

The quiescent current consumed by the LM1771 is one of the major sources of loss within the controller.

However, from a system standpoint this is usually less than 0.5% of the overall efficiency. Therefore, it could

easily be omitted but is shown for completeness:

PIQ = VIN x IQ(18)

8.2.1.2.3.2 Conduction Loss

There are three losses associated with the external FETs. From the DC standpoint there is the I-squared R loss,

caused by the ON-resistance of the FET. This can be modeled for the PMOS by:

PP_COND = D x RDSON_PMOS x IOUT2(19)

and the NMOS by:

PN_COND = (1 - D) x RDSON_NMOS x IOUT2(20)

8.2.1.2.3.3 Switching Loss

The next loss is the switching loss that is caused by the need to charge and discharge the gate capacitance of

the FETs every cycle. This can be approximated by:

PP_SWITCH = VIN x Qg_PMOS x fSW (21)

for the PMOS, and the same approach can be adapted for the NMOS:

PN_SWITCH = VIN x Qg_NMOS x fSW (22)

8.2.1.2.3.4 Transitional Loss

The last FET power loss is the transitional loss. This is caused by switching the PMOS while it is conducting

current. This approach only models the PMOS transition, the NMOS loss is considered negligible because it has

minimal drain to source voltage when it switches due to the conduction of the body diode. Therefore the

transitional loss of the PMOS can be modeled by:

PP_TRANSITIONAL = 0.5 x VIN x IOUT x fSW x (tr+ tf) (23)

Product Folder Links: LM1771

VIN

LM1771 VOUT

GND

COUT

CIN

RFB1

RSNS

EN EN

RFB2

CFF

K = POUT

POUT + PLOSSES

LM1771

www.ti.com

SNVS446D –JUNE 2006–REVISED JANUARY 2016

trand tfare the rise and fall times of the FET and can be found in their corresponding datasheet. Typically these

numbers are simulated using a 6-Ωdrive, which corresponds well to the LM1771. Given this, no adjustment is

needed.

8.2.1.2.3.5 DCR Loss

The last source of power loss in the system that needs to be calculated is the loss associated with the inductor

resistance (DCR) which can be calculated by Equation 24:

PDCR = RDCR x IOUT2(24)

8.2.1.2.3.6 Efficiency

The efficiency, η, can then be calculated by summing all the power losses and then using Equation 25:

(25)

8.2.1.3 Application Curve

Figure 23. Typical Efficiency 5 VIN to 2.5 VOUT

8.2.2 Example Application 5 VIN to 1.8 VOUT

Figure 24 and Table 5 show an application with conversion from 5-V input to 1.8-V output with a switching

frequency of 1.1 MHz and a 2-A maximum output current.

Figure 24. Example Circuit Schematic, 5 VIN to 1.8 VOUT

Product Folder Links: LM1771

VIN

LM1771 VOUT

GND

COUT

CIN

RFB1

RSNS

EN EN

RFB2

CFF

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

www.ti.com

Table 5. Bill of Materials (5-V to 1.8-V Conversion, fSW = 1090 kHz, IOUT = 2 A)

DESIGNATOR DESCRIPTION PART NUMBER QUANTITY VENDOR

U1LM1771, 500 ns LM1771S 1 Texas Instruments

Q1PMOS Si3867DV 1 Siliconix

Q2NMOS Si3460DV 1 Siliconix

CIN 22-μF Capacitor, 0805 GRM21BR60J226ME39 1 Murata

COUT 100-μF Capacitor, 6.3 V, 100 mΩTPSY107M006R0100 1 AVX

RFB1 12.4-kΩResistor, 0603 CRCW06031242F 1 Vishay

RFB2 10-kΩResistor, 0603 CRCW06031002F 1 Vishay

CFF 1-nF Capacitor, 0603 VJ0603102KXXA 1 Vishay

L 3.3-μH Inductor MSS7341-332NLB 1 Coilcraft

8.2.3 Example Application 5 VIN to 3.3 VOUT

Figure 25 and Table 6 show an application with conversion from 5-V input to 3.3-V output with a switching

frequency of 500 kHz and a 5-A maximum output current.

Figure 25. Example Circuit Schematic, 5 VIN to 3.3 VOUT

Table 6. Bill of Materials (5-V to 3.3-V Conversion, fSW = 500 kHz, IOUT = 5 A)

DESIGNATOR DESCRIPTION PART NUMBER QUANTITY VENDOR

U1LM1771, 200 ns LM1771U 1 Texas Instruments

Q1PMOS Si9433BDY 1 Siliconix

Q2NMOS Si4894DY 1 Siliconix

CIN 100-μF Capacitor, 1812 GRM43SR60J107ME20B 1 Murata

COUT 150-μF Capacitor, 6.3 V, 70 mΩNOSD157M006R0070 1 AVX

RFB1 29.4-kΩResistor, 0805 CRCW08052942F 1 Vishay

RFB2 10-kΩResistor, 0805 CRCW08051002F 1 Vishay

CFF 1-nF Capacitor, 0805 VJ0805102KXXA 1 Vishay

L 2.2-μH Inductor DO3316P-222 1 Coilcraft

9 Power Supply Recommendations

The power line feeding the LM1771 must have low impedance. The input capacitor of the system must be placed

as close to VIN as possible. If the power supply is very noisy, an additional bulk capacitor might be necessary in

the system to ensure that clean power is delivered to the IC.

Product Folder Links: LM1771

CIN and COUT GND are close together

Keep path of FET’s gate short

Limit surface of switch node

to reduce capacitance

Feedback components are

close to the FB pin

LM1771

www.ti.com

SNVS446D –JUNE 2006–REVISED JANUARY 2016

10 Layout

10.1 Layout Guidelines

The LM1771, like all switching regulators, requires careful attention to layout to ensure optimal performance. The

following steps must be taken to aid in the layout. For more information refer to Application Note AN-1299

(SNVA074).

1. Ensure that the ground connections of the input capacitor, output capacitor and NMOS are as close as

possible. Ideally these must all be grounded together in close proximity on the component side of the board.

2. Keep the switch node small to minimize EMI without degrading thermal cooling of the FETs.

3. Locate the feedback resistors close to the IC and keep the feedback trace as short as possible. Do not run

any feedback traces near the switch node.

4. Keep the gate traces short and keep them away from the switch node as much as possible.

5. If a small bypass capacitor is used on VIN (0.1 µF) place it as close to the pin, with the ground connection as

close to the chip ground, as possible.

10.2 Layout Examples

Figure 26. LM1771 Layout Example (Top)

Product Folder Links: LM1771

LM1771

SNVS446D –JUNE 2006–REVISED JANUARY 2016

www.ti.com

Layout Examples (continued)

Figure 27. LM1771 Layout Example (Bottom)

10.3 Thermal Considerations

By breaking down the individual power loss in each component it makes it easy to determine the temperature

rise of each component. Generally the expected temperature rise of the LM1771 is extremely low as it is not in

the power path. Therefore the only two items of concern are the PMOS and the NMOS. The power loss in the

PMOS is the sum of the conduction loss and transitional loss, while the NMOS only has conduction loss. It is

assumed that any loss associated with the body diode conduction during the dead-time is negligible.

For completeness of design it is important to watch out for the temperature rise of the inductor. Assuming the

inductor is kept out of saturation the predominant loss is the DC copper resistance. At higher frequencies,

depending on the core material, the core loss could approach or exceed the DCR losses. Consult with the

inductor manufacturer for appropriate temp curves based on current.

Product Folder Links: LM1771

LM1771

www.ti.com

SNVS446D –JUNE 2006–REVISED JANUARY 2016

11 Device and Documentation Support

11.1 Device Support

11.1.1 Third-Party Products Disclaimer

TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT

CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES

OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER

ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.

11.2 Documentation Support

11.2.1 Related Documentation

For related documentation, see the following:

User's Guide, AN-1299 LM5041 Evaluation Board, SNVA074

11.3 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

11.4 Trademarks

E2E is a trademark of Texas Instruments.

All other trademarks are the property of their respective owners.

11.5 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

11.6 Glossary

SLYZ022 —TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

Product Folder Links: LM1771

PACKAGE OPTION ADDENDUM

www.ti.com 8-Oct-2015

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead/Ball Finish

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM1771SMM/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM SNRB

LM1771SSD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 1771S

LM1771TMM/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM SNSB

LM1771TSD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 1771T

LM1771UMM/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM SNTB

LM1771USD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 1771U

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability

information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that

lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between

the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight

in homogeneous material)

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

PACKAGE OPTION ADDENDUM

www.ti.com 8-Oct-2015

Addendum-Page 2

(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish

value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM1771SMM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM1771SSD/NOPB WSON NGG 6 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1

LM1771TMM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM1771TSD/NOPB WSON NGG 6 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1

LM1771UMM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM1771USD/NOPB WSON NGG 6 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 24-Sep-2015

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM1771SMM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0

LM1771SSD/NOPB WSON NGG 6 1000 210.0 185.0 35.0

LM1771TMM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0

LM1771TSD/NOPB WSON NGG 6 1000 210.0 185.0 35.0

LM1771UMM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0

LM1771USD/NOPB WSON NGG 6 1000 210.0 185.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 24-Sep-2015

Pack Materials-Page 2

MECHANICAL DATA

NGG0006A

www.ti.com

SDE06A (Rev A)

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