EVALUATION KIT AVAILABLE MAX15004A/B-MAX15005A/B General Description The MAX15004A/B/MAX15005A/B high-performance, current-mode PWM controllers operate at an automotive input voltage range from 4.5V to 40V (load dump). The input voltage can go lower than 4.5V after startup if IN is bootstrapped to a boosted output voltage. The controllers integrate all the building blocks necessary for implementing fixed-frequency isolated/nonisolated power supplies. The general-purpose boost, flyback, forward, and SEPIC converters can be designed with ease around the MAX15004/MAX15005. The current-mode control architecture offers excellent linetransient response and cycle-by-cycle current limit while simplifying the frequency compensation. Programmable slope compensation simplifies the design further. A fast 60ns current-limit response time, low 300mV current-limit threshold makes the controllers suitable for high-efficiency, high-frequency DC-DC converters. The devices include an internal error amplifier and 1% accurate reference to facilitate the primary-side regulated, single-ended flyback converter or nonisolated converters. An external resistor and capacitor network programs the switching frequency from 15kHz to 500kHz (1MHz for the MAX15005A/B). The MAX15004A/B/MAX15005A/B provide a SYNC input for synchronization to an external clock. The maximum FET-driver duty cycle for the MAX15004A/B is 50%. The maximum duty cycle can be set on the MAX15005A/B by selecting the right combination of RT and CT. The input undervoltage lockout (ON/OFF) programs the input-supply startup voltage and can be used to shutdown the converter to reduce the total shutdown current down to 10A. Protection features include cycle-by-cycle and hiccup current limit, output overvoltage protection, and thermal shutdown. 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Benefits and Features Wide Supply Voltage Range Meets Automotive Power-Supply Operating Requirement Including "Cold Crank" Conditions * 4.5V to 40V Operating Input Voltage Range (Can Operate at Lower Voltage After Startup if Input is Bootstrapped to a Boosted Output) Control Architecture Offers Excellent Performance While Simplifying the Design * Current-Mode Control * 300mV, 5% Accurate Current-Limit Threshold Voltage * Programmable Slope Compensation * 50% (MAX15004) or Adjustable (MAX15005) Maximum Duty Cycle Accurate, Adjustable Switching Frequency and Synchronization Avoids Interference with Sensitive Radio Bands * Switching Frequency Adjustable from 15kHz to 500kHz (1MHz for the MAX15005A/B) * RC Programmable 4% Accurate Switching Frequency * External Frequency Synchronization Built-In Protection Capability for Improved System Reliability * Cycle-by-Cycle and Hiccup Current-Limit Protection * Overvoltage and Thermal-Shutdown Protection * -40C to +125C Automotive Temperature Range * AEC-Q100 Qualified Ordering Information PIN-PACKAGE MAX DUTY CYCLE 16 TSSOP-EP* 50% The MAX15004A/B/MAX15005A/B are available in spacesaving 16-pin TSSOP and thermally enhanced 16-pin TSSOP-EP packages. All devices operate over the -40C to +125C automotive temperature range. MAX15004AAUE/V+ 16 TSSOP-EP* 50% MAX15004BAUE+ 50% 16 TSSOP-EP* Programmable Applications MAX15005AAUE/V+ 16 TSSOP-EP* Programmable MAX15005BAUE+ 16 TSSOP Programmable MAX15005BAUE/V+ 16 TSSOP Programmable Automotive Vacuum Fluorescent Display (VFD) Power Supply Isolated Flyback, Forward, Nonisolated SEPIC, Boost Converters Pin Configuration appears at end of data sheet. 19-0723; Rev 7; 2/17 PART MAX15004AAUE+ 16 TSSOP MAX15004BAUE/V+ 16 TSSOP MAX15005AAUE+ 50% Note: All devices are specified over the -40C to +125C temperature range. +Denotes a lead(Pb)-free/RoHS-compliant package. /V denotes an automotive qualified part. *EP = Exposed pad. MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Absolute Maximum Ratings IN to SGND............................................................-0.3V to +45V IN to PGND............................................................-0.3V to +45V ON/OFF to SGND.......................................-0.3V to (VIN + 0.3V) OVI, SLOPE, RTCT, SYNC, SS, FB, COMP, CS to SGND...................................... -0.3V to (VREG5 + 0.3V) VCC to PGND.........................................................-0.3V to +12V REG5 to SGND........................................................-0.3V to +6V OUT to PGND........................................... -0.3V to (VCC + 0.3V) SGND to PGND.....................................................-0.3V to +0.3V VCC Sink Current (clamped mode).....................................35mA OUT Current (< 10s transient)..........................................1.5A Continuous Power Dissipation* (TA = +70C) 16-Pin TSSOP-EP (derate 21.3mW/C above +70C).............................................................1702mW 16-Pin TSSOP (derate 9.4mW/C above +70C).........754mW Operating Junction Temperature Range........... -40C to +125C Junction Temperature.......................................................+150C Storage Temperature Range............................. -60C to +150C Lead Temperature (soldering, 10s).................................. +300C Soldering Temperature (reflow)........................................+260C *As per JEDEC51 Standard, Multilayer Board. Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Electrical Characteristics (VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS POWER SUPPLY Input Supply Range Operating Supply Current ON/OFF CONTROL Input-Voltage Threshold Input-Voltage Hysteresis Input Bias Current Shutdown Current INTERNAL 7.4V LDO (VCC) Output (VCC) Voltage Set Point Line Regulation UVLO Threshold Voltage UVLO Hysteresis Dropout Voltage Output Current Limit Internal Clamp Voltage INTERNAL 5V LDO (REG5) Output (REG5) Voltage Set Point Line Regulation VIN IQ VON VON/OFF rising IB-ON/OFF VON/OFF = 40V VHYST-ON ISHDN VVCC VUVLO-VCC VHYST-UVLO IVCC-ILIM VREG5 www.maximintegrated.com IREG5-ILIM 1.05 IVCC = 0 to 20mA (sourcing) 7.15 VCC rising 3.15 VIN = 8V to 40V VCC = 7.5V, IREG5 = 0 to 15mA (sourcing) VCC = 5.5V to 10V VCC = 4.5V, IREG5 = 15mA (sourcing) IREG5 sourcing V 3.1 mA 1.23 1.40 V mV 0.5 A 10 20 A 7.4 7.60 1 3.5 0.25 3.75 0.5 V mA 10.0 10.4 10.8 4.75 4.95 5.05 2 32 V mV 45 0.25 V mV/V 500 VIN = 4.5V, IVCC = 20mA (sourcing) IVCC sourcing 40.0 2 75 VON/OFF = 0V VVCC-CLAMP IVCC = 30mA (sinking) Dropout Voltage Output Current Limit 4.5 VIN = 40V, fOSC = 150kHz V V mV/V 0.5 V mA Maxim Integrated 2 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Electrical Characteristics (continued) (VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 1000 kHz OSCILLATOR (RTCT) Oscillator Frequency Range RTCT Peak Trip Level RTCT Valley Trip Level RTCT Discharge Current fOSC fOSC = 2 x fOUT for MAX15004A/B, fOSC = fOUT for MAX15005A/B 15 VTH,RTCT VTL,RTCT IDIS,RTCT Oscillator Frequency Accuracy (Note 2) 0.55 x VREG5 VRTCT = 2V 1.30 0.1 x VREG5 1.33 Minimum On-Time DMAX tON-MIN SYNC Lock-In Frequency Range (Note 4) SYNC High-Level Voltage SYNC Low-Level Voltage SYNC Input Current +4 RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz -4 +4 RT = 21k, CT = 100pF, fOSC (typ) = 500kHz -5 +5 RT = 7k, CT = 100pF, fOSC (typ) = 1MHz -7 +7 SYNC Minimum Input Pulse Width MAX15005A/B, RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz 50 78.5 % 80 81.5 110 170 ns 200 %fOSC 102 2 VSYNC = 0 to 5V mA % VIN = 14V VIH-SYNC ISYNC 1.36 -4 RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz VIL-SYNC V RT = 13.7k, CT = 4.7nF, fOSC (typ) = 18kHz MAX15004A/B Maximum PWM Duty Cycle (Note 3) V V -0.5 0.8 V +0.5 A 50 ns ERROR AMPLIFIER/SOFT-START Soft-Start Charging Current SS Reference Voltage SS Threshold for HICCUP Enable ISS VSS VSS = 0V 8 15 21 A 1.215 1.228 1.240 V VSS rising 1.1 FB Regulation Voltage VREF-FB COMP = FB, ICOMP = -500A to +500A 1.215 FB Input Offset Voltage VOS-FB COMP = 0.25V to 4.5V, ICOMP = -500A to +500A, VSS = 0 to 1.5V -5 FB Input Current COMP Sink Current www.maximintegrated.com ICOMP-SINK VFB = 0 to 1.5V VFB = 1.5V, VCOMP = 0.25V 1.228 -300 3 V 1.240 V +5 mV +300 5.5 nA mA Maxim Integrated 3 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Electrical Characteristics (continued) (VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5) PARAMETER COMP Source Current SYMBOL ICOMPSOURCE CONDITIONS VFB = 1V, VCOMP = 4.5V COMP High Voltage VOH-COMP VFB = 1V, ICOMP = 1mA (sourcing) COMP Low Voltage VOL-COMP VFB = 1.5V, ICOMP = 1mA (sinking) Open-Loop Gain Unity-Gain Bandwidth Phase Margin COMP Positive Slew Rate COMP Negative Slew Rate AEAMP MIN TYP 1.3 2.8 mA VREG5 - 0.5 VREG5 - 0.2 V 0.1 MAX 0.25 UNITS V 100 dB UGFEAMP 1.6 MHz 75 degrees SR+ 0.5 V/s SR- -0.5 V/s PMEAMP PWM COMPARATOR Current-Sense Gain ACS-PWM VCOMP/VCS (Note 5) PWM Propagation Delay to OUT tPD-PWM CS = 0.15V, from VCOMP falling edge: 3V to 0.5V to OUT falling (excluding leading-edge blanking time) PWM Comparator Current-Sense Leading-Edge Blanking Time tCS-BLANK 2.85 3 3.15 V/V 60 ns 50 ns CURRENT-LIMIT COMPARATOR Current-Limit Threshold Voltage Current-Limit Input Bias Current VILIM IB-CS ILIMIT Propagation Delay to OUT tPD-ILIM ILIM Comparator Current-Sense Leading-Edge Blanking Time tCS-BLANK 290 OUT= high, 0 VCS 0.3V 305 -2 From CS rising above VILIM (50mV overdrive) to OUT falling (excluding leading-edge blanking time) Number of Consecutive ILIMIT Events to HICCUP 317 mV +2 A 60 ns 50 ns 7 HICCUP Timeout Clock periods 512 SLOPE COMPENSATION (Note 6) Slope Capacitor Charging Current Slope Compensation Slope Compensation Tolerance (Note 2) Slope Compensation Range www.maximintegrated.com ISLOPE VSLOPE = 100mV 9.8 CSLOPE = 100pF -4 CSLOPE = 100pF CSLOPE = 22pF CSLOPE = 1000pF 10.5 11.2 25 +4 110 2.5 A mV/s % mV/s Maxim Integrated 4 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Electrical Characteristics (continued) (VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5) PARAMETER SYMBOL CONDITIONS MIN TYP MAX 3.5 UNITS OUTPUT DRIVER Driver Output Impedance Driver Peak Output Current ROUT-N VCC = 8V (applied externally), IOUT = 100mA (sinking) 1.7 ROUT-P VCC = 8V (applied externally), IOUT = 100mA (sourcing) 3 IOUT-PEAK OVERVOLTAGE COMPARATOR Overvoltage Comparator Input Threshold Overvoltage Comparator Hysteresis Overvoltage Comparator Delay OVI Input Current THERMAL SHUTDOWN Shutdown Temperature Thermal Hysteresis VOV-TH COUT = 10nF, sinking 1000 COUT = 10nF, sourcing VOVI rising IOVI 1.20 TSHDN THYST From OVI rising above 1.228V to OUT falling, with 50mV overdrive VOVI = 0 to 5V Temperature rising mA 750 VOV-HYST TDOVI 5 1.228 1.26 V 125 mV 1.6 s -0.5 +0.5 A 160 C 15 C Note 1: 100% production tested at +125C. Limits over the temperature range are guaranteed by design. Note 2: Guaranteed by design; not production tested. Note 3: For the MAX15005A/B, DMAX depends upon the value of RT. See Figure 3b and the Oscillator Frequency/External Synchronization section. Note 4: The external SYNC pulse triggers the discharge of the oscillator ramp. See Figure 2. During external SYNC, DMAX = 50% for the MAX15004A/B; for the MAX15005A/B, there is a shift in DMAX with fSYNC/fOSC ratio (see the Oscillator Frequency/ External Synchronization section). Note 5: The parameter is measured at the trip point of latch, with 0 VCS 0.3V, and FB = COMP. Note 6: Slope compensation = (2.5 x 10-9)/CSLOPE mV/s. See the Applications Information section. www.maximintegrated.com Maxim Integrated 5 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Typical Operating Characteristics VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.) 13 10 COUT = 0nF 7 110 135 VCC OUTPUT VOLTAGE vs. VIN SUPPLY VOLTAGE IVCC = 0mA IVCC = 1mA 7.0 IVCC = 20mA 6.5 6.0 5.5 5 10 15 20 25 30 35 VIN SUPPLY VOLTAGE (V) 150 OSCILLATOR FREQUENCY (kHz) 149 148 40 45 10 60 110 160 210 260 310 360 410 460 510 FREQUENCY (kHz) REG5 OUTPUT VOLTAGE vs. VCC VOLTAGE 5.000 4.975 IREG5 = 1mA (SOURCING) 4.950 4.925 4.900 4.875 4.850 IREG5 = 15mA (SOURCING) 4.825 4.800 4.775 4.750 4.725 4.700 5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0 10.5 VCC VOLTAGE (V) OSCILLATOR FREQUENCY (fOSC) vs. VIN SUPPLY VOLTAGE RT = 13.7k CT = 560pF 147 TA = +25C 146 MAX15005 TA = -40C 145 144 143 142 141 140 MAX15004 toc06 10 35 60 85 TEMPERATURE (C) TA = +125C TA = +135C 5.5 10.5 15.5 20.5 25.5 30.5 35.5 40.5 45.5 VIN SUPPLY VOLTAGE (V) www.maximintegrated.com TA = +25C TA = -40C 5 10 40 45 REG5 DROPOUT VOLTAGE vs. IREG5 0.30 0.28 0.25 VCC = 4.5 VIN = VON/OFF 0.23 0.20 0.18 0.15 0.13 0.10 0.08 0.05 0.03 0 15 20 25 30 35 SUPPLY VOLTAGE (V) TA = +125C TA = +135C TA = +25C TA = -40C 0 2 4 6 8 10 IREG5 (mA) 12 14 OSCILLATOR FREQUENCY (fOSC) vs. RT/CT 1000 OSCILLATOR FREQUENCY (kHz) -15 1 MAX15004 toc03 MAX15004 toc02 COUT = 10nF 16 TA = +135C MAX15004 toc07 19 MAX15004 toc08 -40 7.5 5.0 22 4 REG5 OUTPUT VOLTAGE (V) 20 10 0 25 20 19 18 17 16 15 14 13 12 11 10 9 8 7 6 5 SHUTDOWN SUPPLY CURRENT vs. SUPPLY VOLTAGE CT = 100pF CT = 220pF CT = 560pF MAX15004 toc09 50 40 30 28 VIN SHUTDOWN SUPPLY CURRENT (A) 70 60 MAX15005 VIN = 14V CT = 220pF REG5 LDO DROPOUT VOLTAGE (V) 80 31 VIN SUPPLY CURRENT (mA) MAX15004 toc01 100 90 MAX15004 toc04 VIN UVLO HYSTERESIS (mV) 120 110 VCC OUTPUT VOLTAGE (V) VIN SUPPLY CURRENT (ISUPPLY) vs. OSCILLATOR FREQUENCY (fOSC) VIN UVLO HYSTERESIS vs. TEMPERATURE CT = 1000pF 100 CT = 1500pF CT = 2200pF CT = 3300pF 10 1 10 100 1000 RT (k) Maxim Integrated 6 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Typical Operating Characteristics (continued) VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.) 80 75 70 65 CT = 3300pF CT = 2200pF CT = 1500pF 60 CT = 1000pF CT = 560pF 55 50 10 CT = 220pF 100 OUTPUT FREQUENCY (kHz) 70 55 50 52 51 50 49 48 47 CT = 560pF RT = 10k fOSC = fOUT = 180kHz CRTCT = 220pF RRTCT = 10k fOSC = fOUT = 418kHz -40 -15 10 35 60 85 TEMPERATURE (C) 110 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 fSYNC/fOSC RATIO GAIN MAX15004 toc12 75 73 71 69 260 220 180 PHASE 140 1 10 100 1k 10k 100k 1M 10M FREQUENCY (Hz) -15 10 35 60 85 TEMPERATURE (C) 110 135 CS-TO-OUT DELAY vs. TEMPERATURE MAX15004 toc15 VCS OVERDRIVE = 50mV 80 100 0.1 -40 90 300 60 DRIVER OUTPUT PEAK SOURCE AND SINK CURRENT 70 60 VCS OVERDRIVE = 190mV 50 40 30 20 10 0 -40 -15 10 35 60 85 TEMPERATURE (C) 110 135 POWER-UP SEQUENCE THROUGH VIN MAX15004 toc18 MAX15004 toc17 MAX15004 toc16 77 100 340 40 30 20 OVI TO OUT DELAY THROUGH OVERVOLTAGE COMPARATOR 79 65 135 MAX15004 toc14 90 80 70 60 50 10 0 -10 81 67 110 100 MAX15004 toc13 MAX15005 CT = 560pF RT = 13.7k fOSC = fOUT = 150kHz 83 ERROR AMPLIFIER OPEN-LOOP GAIN AND PHASE vs. FREQUENCY 65 60 53 45 1000 GAIN (dB) MAXIMUM DUTY CYCLE (%) 75 85 46 MAXIMUM DUTY CYCLE vs. fSYNC/fOSC RATIO 80 fOUT = 75kHz 54 MAXIMUM DUTY CYCLE (%) 85 55 PHASE (DEGREES) CS-TO-OUT DELAY (ns) MAXIMUM DUTY CYCLE (%) 90 MAX15005 MAXIMUM DUTY CYCLE vs. TEMPERATURE MAX15004 MAXIMUM DUTY CYCLE vs. TEMPERATURE MAX15004 toc11 CT = 100pF 95 MAXIMUM DUTY CYCLE (%) 100 MAX15004 toc10 MAX15005 MAXIMUM DUTY CYCLE vs. OUTPUT FREQUENCY (fOUT) COUT = 10nF VOUT VOVI VOUT 5V/div VOUT 2V/div VIN 10V/div VCC 5V/div VON/OFF = 5V REG5 5V/div IOUT 1A/div VOVI 500mV/div VOUT 5V/div 1s/div www.maximintegrated.com 400ns/div 2ms/div Maxim Integrated 7 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Typical Operating Characteristics (continued) VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.) POWER-UP SEQUENCE THROUGH ON/OFF POWER-DOWN SEQUENCE THROUGH VIN POWER-DOWN SEQUENCE THROUGH ON/OFF MAX15004 toc21 MAX15004 toc20 MAX15004 toc19 VON/OFF = 5V ON/OFF 5V/div VIN 10V/div ON/OFF 5V/div VCC 5V/div VCC 5V/div VCC 5V/div REG5 5V/div REG5 5V/div REG5 5V/div VOUT 5V/div VOUT 5V/div VOUT 5V/div 400ms/div 1ms/div 4ms/div LINE TRANSIENT FOR VIN STEP FROM 14V TO 5.5V LINE TRANSIENT FOR VIN STEP FROM 14V TO 40V MAX15004 toc22 MAX15004 toc23 VIN 10V/div VCC 5V/div VIN 20V/div VCC 5V/div REG5 5V/div REG5 5V/div VOUT 5V/div VOUT 5V/div 100s/div 100s/div HICCUP MODE FOR FLYBACK CIRCUIT (FIGURE 7) DRAIN WAVEFORM IN FLYBACK CONVERTER (FIGURE 7) MAX15004 toc25 MAX15004 toc24 ILOAD = 10mA VCS 200mV/div 10V/div VANODE 1V/div ISHORT 500mA/div 10s/div www.maximintegrated.com 4s/div Maxim Integrated 8 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Pin Description PIN NAME 1 IN FUNCTION 2 ON/OFF 3 OVI 4 SLOPE 5 N.C. 6 RTCT Oscillator-Timing Network Input. Connect a resistor from RTCT to REG5 and a capacitor from RTCT to SGND to set the oscillator frequency (see the Oscillator Frequency/External Synchronization section). 7 SGND Signal Ground. Connect SGND to SGND plane. 8 SYNC External-Clock Synchronization Input. Connect SYNC to SGND when not using an external clock. 9 SS Soft-Start Capacitor Input. Connect a capacitor from SS to SGND to set the soft-start time interval. 10 FB Internal Error-Amplifier Inverting Input. The noninverting input is internally connected to SS. 11 COMP 12 CS 13 REG5 5V Low-Dropout Regulator Output. Bypass REG5 with a 1F ceramic capacitor to SGND. 14 PGND Power Ground. Connect PGND to the power ground plane. 15 OUT Gate Driver Output. Connect OUT to the gate of the external n-channel MOSFET. 16 VCC 7.4V Low-Dropout Regulator Output--Driver Power Source. Bypass VCC with 0.1F and 1F or higher ceramic capacitors to PGND. Do not connect external supply or bootstrap to VCC. -- EP Exposed Pad (MAX15004A/MAX15005A only). Connect EP to the SGND plane to improve thermal performance. Do not use the EP as an electrical connection. Input Power Supply. Bypass IN with a minimum 0.1F ceramic capacitor to PGND. ON/OFF Input. Connect ON/OFF to IN for always-on operation. To externally program the UVLO threshold of the IN supply, connect a resistive divider between IN, ON/OFF, and SGND. Pull ON/OFF to SGND to disable the controller. Overvoltage Comparator Input. Connect a resistive divider between the output of the power supply, OVI, and SGND to set the output overvoltage threshold. Programmable Slope Compensation Capacitor Input. Connect a capacitor (CSLOPE) to SGND to set the amount of slope compensation. Slope compensation = (2.5 x 10-9)/CSLOPE mV/s with CSLOPE in farads. No Connection. Not internally connected. Error-Amplifier Output. Connect the frequency compensation network between FB and COMP. Current-Sense Input. The current-sense signal is compared to a signal proportional to the error-amplifier output voltage. www.maximintegrated.com Maxim Integrated 9 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Functional Diagram IN 1 MAX15004A/B MAX15005A/B 1.228V OFF ON/OFF 2 ON/OFF COMP PREREGULATOR OFF 3.5V UVLO REFERENCE 16 VCC 7.4V LDO REG UVB DRIVER 15 OUT 14 PGND VCC THERMAL SHUTDOWN SET RESET OV-COMP OVI 3 UVB ILIMIT COMP 5V LDO REG 13 REG5 50ns LEAD DELAY 12 CS 0.3V 1.228V PWMCOMP R OVRLD SLOPE 4 SLOPE COMPENSATION 2R RTCT 6 OSCILLATOR 10 FB SGND 7 RESET SYNC 8 11 COMP SS_OK CLK 7 CONSECUTIVE EVENTS COUNTER EAMP 1.228V 9 SS REF-AMP OVRLD www.maximintegrated.com Maxim Integrated 10 MAX15004A/B-MAX15005A/B Detailed Description The MAX15004A/B/MAX15005A/B are high-performance, current-mode PWM controllers for wide inputvoltage range isolated/nonisolated power supplies. These controllers are for use as general-purpose boost, flyback, and SEPIC controllers. The input voltage range of 4.5V to 40V makes it ideal in automotive applications such as vacuum fluorescent display (VFD) power supplies. The internal low-dropout regulator (VCC regulator) enables the MAX15004A/B/MAX15005A/B to operate directly from an automotive battery input. The input voltage can go lower than 4.5V after startup if IN is bootstrapped to a boosted output voltage. The undervoltage lockout (ON/OFF) allows the devices to program the input-supply startup voltage and ensures predictable operation during brownout conditions. The devices contain two internal regulators, VCC and REG5. The VCC regulator output voltage is set at 7.4V and REG5 regulator output voltage at 5V 2%. The input undervoltage lockout (UVLO) circuit monitors the VCC voltage and turns off the converter when the VCC voltage drops below 3.5V (typ). An external resistor and capacitor network programs the switching frequency from 15kHz to 500kHz. The MAX15004A/B/MAX15005A/B provide a SYNC input for synchronization to an external clock. The OUT (FET-driver output) duty cycle for the MAX15004A/B is 50%. The maximum duty cycle can be set on MAX15005A/B by selecting the right combination of RT and CT. The RTCT discharge current is trimmed to 2%, allowing accurate setting of the duty cycle for the MAX15005. An internal slope-compensation circuit stabilizes the current loop when operating at higher duty cycles and can be programmed externally. The MAX15004/MAX15005 include an internal error amplifier with 1% accurate reference to regulate the output in nonisolated topologies using a resistive divider. The internal reference connected to the noninverting input of the error amplifier can be increased in a controlled manner to obtain soft-start. A capacitor connected at SS to ground programs soft-start to reduce inrush current and prevent output overshoot. The MAX15004/MAX15005 include protection features like hiccup current limit, output overvoltage, and thermal shutdown. The hiccup current-limit circuit reduces the power delivered to the electronics powered by the MAX15004/ MAX15005 converter during severe fault conditions. The overvoltage circuit senses the output using the path different from the feedback path to provide meaningful overvoltage protection. During continuous high input operation, the www.maximintegrated.com 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers power dissipation into the MAX15004/MAX15005 could exceed its limit. Internal thermal shutdown protection safely turns off the converter when the junction heats up to 160C. Current-Mode Control Loop The advantages of current-mode control overvoltagemode control are twofold. First, there is the feed-forward characteristic brought on by the controller's ability to adjust for variations in the input voltage on a cycle-by-cycle basis. Secondly, the stability requirements of the currentmode controller are reduced to that of a single-pole system unlike the double pole in voltage-mode control. The MAX15004/MAX15005 offer peak current-mode control operation to make the power supply easy to design with. The inherent feed-forward characteristic is useful especially in an automotive application where the input voltage changes fast during cold-crank and load dump conditions. While the current-mode architecture offers many advantages, there are some shortcomings. For higher dutycycle and continuous conduction mode operation where the transformer does not discharge during the off duty cycle, subharmonic oscillations appear. The MAX15004/ MAX15005 offer programmable slope compensation using a single capacitor. Another issue is noise due to turn-on of the primary switch that may cause the premature end of the on cycle. The current-limit and PWM comparator inputs have leading-edge blanking. All the shortcomings of the current-mode control are addressed in the MAX15004/ MAX15005, making it ideal to design for automotive power conversion applications. Internal Regulators VCC and REG5 The internal LDO converts the automotive battery voltage input to a 7.4V output voltage (VCC). The VCC output is set at 7.4V and operates in a dropout mode at input voltages below 7.5V. The internal LDO is capable of delivering 20mA current, enough to provide power to internal control circuitry and the gate drive. The regulated VCC keeps the driver output voltage well below the absolute maximum gate voltage rating of the MOSFET especially during the double battery and load dump conditions. The second 5V LDO regulator from VCC to REG5 provides power to the internal control circuits. This LDO can also be used to source 15mA of external load current. Bypass VCC and REG5 with a parallel combination of 1F and 0.1F low-ESR ceramic capacitors. Additional capacitors (up to 22F) at VCC can be used although they are not necessary for proper operation of the MAX15004/ MAX15005. Maxim Integrated 11 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Startup Operation/UVLO/ON/OFF The MAX15004A/B/MAX15005A/B feature two undervoltage lockouts (UVLO). The internal UVLO monitors the VCC-regulator and turns on the converter once VCC rises above 3.5V. The internal UVLO circuit has about 0.5V hysteresis to avoid chattering during turn-on. VIN MAX15004A/B MAX15005A/B R1 ON/OFF An external undervoltage lockout can be achieved by controlling the voltage at the ON/OFF input. The ON/ OFF input threshold is set at 1.23V (rising) with 75mV hysteresis. 1.23V R2 Before any operation can commence, the ON/OFF voltage must exceed the 1.23V threshold. Calculate R1 in Figure 1 by using the following formula: V = R1 ON - 1 x R2 VUVLO where VUVLO is the ON/OFF's 1.23V rising threshold, and VON is the desired input startup voltage. Choose an R2 value in the 100k range. The UVLO circuits keep the PWM comparator, ILIM comparator, oscillator, and output driver shut down to reduce current consumption (see the Functional Diagram). The ON/OFF input can be used to disable the MAX15004/MAX15005 and reduce the standby current to less than 20A. Soft-Start The MAX15004/MAX15005 are provided with an externally adjustable soft-start function, saving a number of external components. The SS is a 1.228V reference bypass connection for the MAX15004A/B/MAX15005A/B and also controls the soft-start period. At startup, after VIN is applied and the UVLO thresholds are reached, the device enters soft-start. During soft-start, 15A is sourced into the capacitor (CSS) connected from SS to GND causing the reference voltage to ramp up slowly. The HICCUP mode of operation is disabled during soft-start. When VSS reaches 1.228V, the output as well as the HICCUP mode become fully active. Set the soft-start time (tSS) using following equation: t SS = Oscillator Frequency/External Synchronization Use an external resistor and capacitor at RTCT to program the MAX15004A/B/MAX15005A/B internal oscillator frequency from 15kHz to 1MHz. The MAX15004A/B output switching frequency is one-half the programmed oscillator frequency with a 50% maximum duty-cycle limit. The MAX15005A/B output switching frequency is the same as the oscillator frequency. The RC network connected to RTCT controls both the oscillator frequency and the maximum duty cycle. The CT capacitor charges and discharges from (0.1 x VREG5) to (0.55 x VREG5). It charges through RT and discharges through an internal trimmed controlled current sink. The maximum duty cycle is inversely proportional to the discharge time (tDISCHARGE). See Figure 3a and Figure 3b for a coarse selection of capacitor values for a given switching frequency and maximum duty cycle and then use the following equations to calculate the resistor value to fine-tune the switching frequency and verify the worst-case maximum duty cycle. t CHARGE = 1.23(V) x C SS 15 x 10 -6 (A ) where tSS is in seconds and CSS is in farads. The soft-start programmability is important to control the input inrush current issue and also to avoid the MAX15004/MAX15005 power supply from going into the unintentional hiccup during the startup. The required softstart time depends on the topology used, current-limit setting, output capacitance, and the load condition. www.maximintegrated.com Figure 1. Setting the MAX15004A/B/MAX15005A/B Undervoltage-Lockout Threshold t DISCHARGE = D MAX f OSC t RT = CHARGE 0.7 x CT 2.25(V) x RT x CT (1.33 x 10 -3 (A) x RT) - 3.375(V) 1 f OSC = t CHARGE + t DISCHARGE where fOSC is the oscillator frequency, RT is the resistance connected from RTCT to REG5, and CT is the capacitor connected from RTCT to SGND. For the Maxim Integrated 12 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers most accuracy, CT should include all additional stray capacitance (typically 25pF to 35pF). The MAX15004A/B is a 50% maximum duty-cycle part, while the MAX15005A/B is a 100% maximum duty-cycle part: f OUT = 1 f OSC 2 for the MAX15004A/B and: f OUT = f OSC for the MAX15005A/B. The MAX15004A/B/MAX15005A/B can be synchronized using an external clock at the SYNC input. For proper frequency synchronization, SYNC's input frequency must be at least 102% of the programmed internal oscillator frequency. Connect SYNC to SGND when not using an external clock. A rising clock edge on SYNC is interpreted as a synchronization input. If the SYNC signal is lost, the internal oscillator takes control of the switching rate, returning the switching frequency to that set by RC network connected to RTCT. This maintains output regulation even with intermittent SYNC signals. n-Channel MOSFET Driver OUT drives the gate of an external n-channel MOSFET. The driver is powered by the internal regulator (VCC), internally set to approximately 7.4V. The regulated VCC voltage keeps the OUT voltage below the maximum gate voltage rating of the external MOSFET. OUT can source 750mA and sink 1000mA peak current. The average current sourced by OUT depends on the switching frequency and total gate charge of the external MOSFET. Error Amplifier The MAX15004A/B/MAX15005A/B include an internal error amplifier. The noninverting input of the error amplifier is connected to the internal 1.228V reference and feedback is provided at the inverting input. High 100dB open-loop gain and 1.6MHz unity-gain bandwidth allow good closed-loop bandwidth and transient response. MAX15004A/B (DMAX = 50%) WITH SYNC INPUT WITHOUT SYNC INPUT RTCT CLKINT SYNC OUT D = 50% D = 50% MAX15005A/B (DMAX = 81%) WITH SYNC INPUT WITHOUT SYNC INPUT RTCT CLKINT SYNC OUT D = 81.25% D = 80% Figure 2. Timing Diagram for Internal Oscillator vs. External SYNC and DMAX Behavior www.maximintegrated.com Maxim Integrated 13 MAX15004A/B-MAX15005A/B MAX15005 MAXIMUM DUTY CYCLE vs. OUTPUT FREQUENCY (fOUT) 100 OSCILLATOR FREQUENCY (kHz) MAXIMUM DUTY CYCLE (%) 90 85 80 75 CT = 3300pF 70 CT = 2200pF CT = 1500pF 60 CT = 1000pF CT = 560pF 55 50 10 OSCILLATOR FREQUENCY (fOSC) vs. RT/CT 1000 CT = 100pF 95 65 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers CT = 100pF CT = 220pF CT = 560pF CT = 1000pF 100 CT = 1500pF CT = 2200pF CT = 3300pF CT = 220pF 100 OUTPUT FREQUENCY (kHz) 1000 10 1 10 100 1000 RT (k) Figure 3a. MAX15005 Maximum Duty Cycle vs. Output Frequency. Figure 3b. Oscillator Frequency vs. RT/CT Moreover, the source and sink current capability of 2mA provides fast error correction during the output load transient. For Figure 5, calculate the power-supply output voltage using the following equation: Current Limit R VOUT= 1 + A VREF RB where VREF = 1.228V. The amplifier's noninverting input is internally connected to a soft-start circuit that gradually increases the reference voltage during startup. This forces the output voltage to come up in an orderly and well-defined manner under all load conditions. Slope Compensation The MAX15004A/B/MAX15005A/B use an internal ramp generator for slope compensation. The internal ramp signal resets at the beginning of each cycle and slews at the rate programmed by the external capacitor connected to SLOPE. The amount of slope compensation needed depends on the downslope of the current waveform. Adjust the MAX15004A/B/MAX15005A/B slew rate up to 110mV/s using the following equation: Slope compensation (mV s) = 2.5 x 10 -9 (A) C SLOPE where CSLOPE is the external capacitor at SLOPE in farads. www.maximintegrated.com The current-sense resistor (RCS), connected between the source of the MOSFET and ground, sets the current limit. The CS input has a voltage trip level (VCS) of 305mV. The current-sense threshold has 5% accuracy. Set the currentlimit threshold 20% higher than the peak switch current at the rated output power and minimum input voltage. Use the following equation to calculate the value of RS: = R S VCS (IPK x 1.2) where IPRI is the peak current that flows through the MOSFET at full load and minimum VIN. When the voltage produced by this current (through the current-sense resistor) exceeds the current-limit comparator threshold, the MOSFET driver (OUT) quickly terminates the on-cycle. In most cases, a short-time constant RC filter is required to filter out the leading-edge spike on the sense waveform. The amplitude and width of the leading edge depends on the gate capacitance, drain capacitance (including interwinding capacitance), and switching speed (MOSFET turn-on time). Set the RC time constant just long enough to suppress the leading edge. For a given design, measure the leading spike at the highest input and rated output load to determine the value of the RC filter. Maxim Integrated 14 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Inductor Selection in Boost Configuration VIN Using the following equation, calculate the minimum inductor value so that the converter remains in continuous mode operation at minimum output current (IOMIN): REG5 MAX15004A/B MAX15005A/B L MIN = N R1 where: D= RCS 0.3V CURRENT-LIMIT COMPARATOR CCS RS Figure 4. Reducing Current-Sense Threshold The low 305mV current-limit threshold reduces the power dissipation in the current-sense resistor. The current-limit threshold can be further reduced by adding a DC offset to the CS input from REG5 voltage. Do not reduce the current-limit threshold below 150mV as it may cause noise issues. See Figure 4. For a new value of the current-limit threshold (VILIM_LOW), calculate the value of R1 using the following equation: 4.75 x R CS R1 = 0.290 - VILIM_LOW and The higher value of IOMIN reduces the required inductance; however, it increases the peak and RMS currents in the switching MOSFET and inductor. Use IOMIN from 10% to 25% of the full load current. The VD is the forward voltage drop of the external Schottky diode, D is the duty cycle, and VDS is the voltage drop across the external switch. Select the inductor with low DC resistance and with a saturation current (ISAT) rating higher than the peak switch current limit of the converter. Input Capacitor Selection in Boost Configuration The input current for the boost converter is continuous and the RMS ripple current at the input capacitor is low. Calculate the minimum input capacitor value and maximum ESR using the following equations: C IN = Boost Converter The MAX15004A/B/MAX15005A/B can be configured for step-up conversion. The boost converter output can be fed back to IN through a Schottky diode (see Figure 5) so the controller can function during low voltage conditions such as cold-crank. Use a Schottky diode (DVIN) in the VIN path to avoid backfeeding the input source. Use the equations in the following sections to calculate inductor (LMIN), input capacitor (CIN), and output capacitor (COUT) when using the converter in boost operation. VOUT + VD - VIN VOUT + VD - VDS I OMIN = (0.1x I O ) to (0.25 x I O ) Applications Information www.maximintegrated.com VIN 2 x D x 2 x f OUT x VOUT x I OMIN IL x D 4 x f OUT x VQ ESR = where : VESR IL (V - VDS ) x D IL = IN L x f OUT VDS is the total voltage drop across the external MOSFET plus the voltage drop across the inductor ESR. IL is peak-to-peak inductor ripple current as calculated above. VQ is the portion of input ripple due to the capacitor Maxim Integrated 15 MAX15004A/B-MAX15005A/B VIN 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers DVIN CIN CVIN 1F CREG5 0.1F REG5 6 IN MAX15004A/B MAX15005A/B VCC OUT CFF CF CS 11 10 VOUT 18V D3 16 COUT CVCC 4.7F RTCT CT RF L 1 13 RT DVBS COMP 15 Q RCS 12 RA CCS FB SLOPE SS CSLOPE RB PGND 9 4 RS CSS Figure 5. Application Schematic discharge and VESR is the contribution due to ESR of the capacitor. Assume the input capacitor ripple contribution due to ESR (VESR) and capacitor discharge (VQ) is equal when using a combination of ceramic and aluminum capacitors. During the converter turn-on, a large current is drawn from the input source especially at high output to input differential. The MAX15004/MAX15005 are provided with a programmable soft-start; however, a large storage capacitor at the input may be necessary to avoid chattering due to finite hysteresis. Output Capacitor Selection in Boost Configuration For the boost converter, the output capacitor supplies the load current when the main switch is on. The required output capacitance is high, especially at higher duty cycles. Also, the output capacitor ESR needs to be low enough to minimize the voltage drop due to the ESR while supporting the load current. Use the following equations to calculate the output capacitor, for a specified output ripple. All ripple values are peak-to-peak. ESR = VESR IO I x D MAX C OUT = O VQ x f OUT IO is the load current, VQ is the portion of the ripple due to the capacitor discharge, and VESR is the contribution due to the ESR of the capacitor. DMAX is the maximum duty cycle at the minimum input voltage. Use a combination of low-ESR ceramic and high-value, low-cost aluminum capacitors for lower output ripple and noise. www.maximintegrated.com Maxim Integrated 16 MAX15004A/B-MAX15005A/B Calculating Power Loss in Boost Converter The MAX15004A/MAX15005A devices are available in a thermally enhanced package and can dissipate up to 1.7W at +70C ambient temperature. The total power dissipation in the package must be limited so that the junction temperature does not exceed its absolute maximum rating of +150C at maximum ambient temperature; however, Maxim recommends operating the junction at about +125C for better reliability. The average supply current (IDRIVE-GATE) required by the switch driver is: IDRIVE-GATE = Q g x f OUT where Qg is total gate charge at 7.4V, a number available from MOSFET data sheet. The supply current in the MAX15004A/B/MAX15005A/B is dependent on the switching frequency. See the Typical Operating Characteristics to find the supply current ISUPPLY of the MAX15004A/B/MAX15005A/B at a given operating frequency. The total power dissipation (PT) in the device due to supply current (ISUPPLY) and the current required to drive the switch (IDRIVEGATE) is calculated using following equation. PT =VINMAX x (I SUPPLY + IDRIVE -GATE ) MOSFET Selection in Boost Converter The MAX15004A/B/MAX15005A/B drive a wide variety of n-channel power MOSFETs. Since VCC limits the OUT output peak gate-drive voltage to no more than 11V, a 12V (max) gate voltage-rated MOSFET can be used without an additional clamp. Best performance, especially at low-input voltages (5VIN), is achieved with low-threshold n-channel MOSFETs that specify on-resistance with a gate-source voltage (VGS) of 2.5V or less. When selecting the MOSFET, key parameters can include: 1) Total gate charge (Qg). 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MOSFET must be greater than the maximum output voltage setting plus a diode drop. The 10V additional margin is recommended for spikes at the MOSFET drain due to the inductance in the rectifier diode and output capacitor path. In addition, Qg helps predict the current needed to drive the gate at the selected operating frequency when the internal LDO is driving the MOSFET. Slope Compensation in Boost Configuration The MAX15004A/B/MAX15005A/B use an internal ramp generator for slope compensation to stabilize the current loop when operating at duty cycles above 50%. It is advisable to add some slope compensation even at lower than 50% duty cycle to improve the noise immunity. The slope compensations should be optimized because too much slope compensation can turn the converter into the voltage-mode control. The amount of slope compensation required depends on the downslope of the inductor current when the main switch is off. The inductor downslope depends on the input to output voltage differential of the boost converter, inductor value, and the switching frequency. Theoretically, the compensation slope should be equal to 50% of the inductor downslope; however, a little higher than 50% slope is advised. Use the following equation to calculate the required compensating slope (mc) for the boost converter: mc = (VOUT - VIN ) x R S x 10 -3 (mV s) 2L The internal ramp signal resets at the beginning of each cycle and slews at the rate programmed by the external capacitor connected to SLOPE. Adjust the MAX15004A/B/ MAX15005A/B slew rate up to 110mV/s using the following equation: C SLOPE = 2.5 x 10 -9 mc(mV s) 2) Reverse-transfer capacitance or charge (CRSS). where CSLOPE is the external capacitor at SLOPE in farads. 3) On-resistance (RDS(ON)). Flyback Converter 4) Maximum drain-to-source voltage (VDS(MAX)). 5) Maximum gate frequencies threshold voltage (VTH(MAX)). At high switching, dynamic characteristics (parameters 1 and 2 of the above list) that predict switching losses have more impact on efficiency than RDS(ON), which predicts DC losses. Qg includes all capacitances associated with charging the gate. The VDS(MAX) of the selected www.maximintegrated.com The choice of the conversion topology is the first stage in power-supply design. The topology selection criteria include input voltage range, output voltage, peak currents in the primary and secondary circuits, efficiency, form factor, and cost. For an output power of less than 50W and a 1:2 input voltage range with small form factor requirements, the flyback topology is the best choice. It uses a minimum Maxim Integrated 17 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers of components, thereby reducing cost and form factor. The flyback converter can be designed to operate either in continuous or discontinuous mode of operation. In discontinuous mode of operation, the transformer core completes its energy transfer during the off-cycle, while in continuous mode of operation, the next cycle begins before the energy transfer is complete. The discontinuous mode of operation is chosen for the present example for the following reasons: The secondary inductance determines the time required to discharge the core. Use the following equations to calculate the secondary inductance: It maximizes the energy storage in the magnetic component, thereby reducing size. where: Simplifies the dynamic stability compensation design (no right-half plane zero). VD = Secondary diode forward voltage drop Higher unity-gain bandwidth. LS (VOUT + VD ) x (D OFFMIN) 2 2 x I OUT x f OUT(MAX) D OFF = t OFF t ON + t OFF DOFFMIN = Minimum DOFF IOUT = Maximum output rated current A major disadvantage of discontinuous mode operation is the higher peak-to-average current ratio in the primary and secondary circuits. Higher peak-to-average current means higher RMS current, and therefore, higher loss and lower efficiency. For low-power converters, the advantages of using discontinuous mode easily surpass the possible disadvantages. Moreover, the drive capability of the MAX15004/MAX15005 is good enough to drive a large switching MOSFET. With the presently available MOSFETs, power output of up to 50W is easily achievable with a discontinuous mode flyback topology using the MAX15004/MAX15005 in automotive applications. Step 2) The rising current in the primary builds the energy stored in the core during on-time, which is then released to deliver the output power during the off-time. Primary inductance is then calculated to store enough energy during the on-time to support the maximum output power. Transformer Design Step 3) Calculate the secondary to primary turns ratio (NSP) and the bias winding to primary turns ratio (NBP) using the following equations: Step-by-step transformer specification design for a discontinuous flyback example is explained below. LP = D= Step 2) Calculate primary winding inductance for sufficient energy to support the maximum load. Step 3) Calculate the secondary and bias winding turns ratios. Step 4) Calculate the RMS current in the primary and estimate the secondary RMS current. Step 5) Consider proper sequencing of windings and transformer construction for low leakage. Step 1) As discussed earlier, the core must be discharged during the off-cycle for discontinuous mode operation. www.maximintegrated.com t ON t ON + t OFF DMAX = Maximum D. Follow the steps below for the discontinuous mode transformer: Step 1) Calculate the secondary winding inductance for guaranteed core discharge within a minimum offtime. VINMIN 2 x D MAX 2 x 2 x POUT x f OUT(MAX) = N SP NS = NP LS LP and = NBP NBIAS 11.7 = NP VOUT + 0.35 The forward bias drops of the secondary diode and the bias rectifier diode are assumed to be 0.35V and 0.7V, respectively. Refer to the diode manufacturer's data sheet to verify these numbers. Step 4) The transformer manufacturer needs the RMS current maximum values in the primary, secondary, and bias windings to design the wire diameter for the different windings. Use only wires with a diameter smaller Maxim Integrated 18 MAX15004A/B-MAX15005A/B than 28AWG to keep skin effect losses under control. To achieve the required copper cross-section, multiple wires must be used in parallel. Multifilar windings are common in high-frequency converters. Maximum RMS currents in the primary and secondary occur at 50% duty cycle (minimum input voltage) and maximum output power. Use the following equations to calculate the primary and secondary RMS currents: = IPRMS I SRMS = POUT D MAX x 0.5 x D MAX x x VINMIN 3 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers total gate charge, and the transition loss during turn-off. There are no transition losses during turn-on since the primary current starts from zero in the discontinuous conduction mode. MOSFET derating may be necessary to avoid damage during system turn-on and any other fault conditions. Use the following equation to estimate the power dissipation due to the power MOSFET: PMOS = (1.4 x R DSON x I 2 PRMS ) + (Q g x VIN x f OUTMAX ) + xI x t xf V ( INMAX PK OFF OUTMAX ) 4 I OUT D OFFMAX 0.5 x D OFFMAX 3 The bias current for most MAX15004/MAX15005 applications is about 20mA and the selection of wire depends more on convenience than on current capacity. Step 5) The winding technique and the windings sequence is important to reduce the leakage inductance spike at switch turn-off. For example, interleave the secondary between two primary halves. Keep the bias winding close to the secondary, so that the bias voltage tracks the output voltage. MOSFET Selection MOSFET selection criteria include the maximum drain voltage, peak/RMS current in the primary and the maximum-allowable power dissipation of the package without exceeding the junction temperature limits. The voltage seen by the MOSFET drain is the sum of the input voltage, the reflected secondary voltage through transformer turns ratio and the leakage inductance spike. The MOSFET's absolute maximum VDS rating must be higher than the worst-case (maximum input voltage and output load) drain voltage. N = VINMAX + P x (VOUT + VD ) + VSPIKE VDSMAX N S Lower maximum VDS requirement means a shorter channel, lower RDS(ON), lower gate charge, and smaller package. A lower NP/NS ratio allows a low VDSMAX specification and keeps the leakage inductance spike under control. A resistor/diode/capacitor snubber network can be also used to suppress the leakage inductance spike. The DC losses in the MOSFET can be calculated using the value for the primary RMS maximum current. Switching losses in the MOSFET depend on the operating frequency, www.maximintegrated.com + C DS x VDS 2 x f OUTMAX 2 where: Qg = Total gate charge of the MOSFET (C) at 7.4V VIN = Input voltage (V) tOFF = Turn-off time (s) CDS = Drain-to-source capacitance (F) Output Filter Design The output capacitance requirements for the flyback converter depend on the peak-to-peak ripple acceptable at the load. The output capacitor supports the load current during the switch on-time. During the off-cycle, the transformer secondary discharges the core replenishing the lost charge and simultaneously supplies the load current. The output ripple is the sum of the voltage drop due to charge loss during the switch on-time and the ESR of the output capacitor. The high switching frequency of the MAX15004/ MAX15005 reduces the capacitance requirement. An additional small LC filter may be necessary to suppress the remaining low-energy high-frequency spikes. The LC filter also helps attenuate the switching frequency ripple. Care must be taken to avoid any compensation problems due to the insertion of the additional LC filter. Design the LC filter with a corner frequency at more than a decade higher than the estimated closed-loop, unitygain bandwidth to minimize its effect on the phase margin. Use 1F to 10F low-ESR ceramic capacitors and calculate the inductance using following equation: L 1 4 x 10 3 x f C 2 x C where fC = estimated converter closed-loop unity-gain frequency. Maxim Integrated 19 MAX15004A/B-MAX15005A/B SEPIC Converter The MAX15004A/B/MAX15005A/B can be configured for SEPIC conversion when the output voltage must be lower and higher than the input voltage when the input voltage varies through the operating range. The duty-cycle equation: VO D = VIN 1 - D indicates that the output voltage is lower than the input for a duty cycle lower than 0.5 while VOUT is higher than the input at a duty cycle higher than 0.5. The inherent advantage of the SEPIC topology over the boost converter is a complete isolation of the output from the source during a fault at the output. The SEPIC converter output can be fed back to IN through a Schottky diode (see Figure 6) so the controller can function during low voltage conditions such as cold-crank. Use a Schottky diode (DVIN) in the VIN path to avoid backfeeding the input source. 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Inductor Selection in SEPIC Converter Use the following equations to calculate the inductance values. Assume both L1 and L2 are equal and that the inductor ripple current (IL) is equal to 20% of the input current at nominal input voltage to calculate the inductance value. V x D MAX = L L= = IN-MIN 1 L2 2 x f OUT x IL 0.2 x I OUT -MAX x D MAX IL = (1 - D MAX ) x where fOUT is the converter switching frequency and is the targeted system efficiency. Use the coupled inductors MSD-series from Coilcraft or PF0553-series from Pulse Engineering, Inc. Make sure the inductor saturating current rating (ISAT) is 30% higher than the peak inductor current calculated using the following equation. Use the current-sense resistor calculated based on the ILPK value from the equation below (see the Current Limit section). The SEPIC converter design includes sizing of inductors, a MOSFET, series capacitance, and the rectifier diode. I x D MAX The inductance is determined by the allowable ripple = ILPK OUT -MAX + I OUT -MAX + IL (1 - D MAX ) x current through all the components mentioned above. Lower ripple current means lower peak and RMS currents MOSFET, Diode, and Series Capacitor and lower losses. The higher inductance value needed Selection in a SEPIC Converter for a lower ripple current means a larger-sized inductor, For the SEPIC configuration, choose an n-channel which is a more expensive solution. The inductors (L1 and MOSFET with a VDS rating at least 20% higher than the L2) can be independent, however, winding them on the sum of the output and input voltages. When operating at same core reduces the ripple currents. a high switching frequency, the gate charge and switchCalculate the maximum duty cycle using the following ing losses become significant. Use low gate-charge equation and choose the RT and CT values accordMOSFETs. The RMS current of the MOSFET is: ingly for a given switching frequency (see the Oscillator Frequency/External Synchronization section). D VOUT + VD D MAX = VIN-MIN + VOUT + VD - (VDS + VCS ) where VD is the forward voltage of the Schottky diode, VCS (0.305V) is the current-sense threshold of the MAX15004/MAX15005, and VDS is the voltage drop across the switching MOSFET during the on-time. www.maximintegrated.com = IMOS -RMS (A) (I ) 2 + (ILDC ) 2 + (ILPK x ILDC ) x MAX LPK 3 where ILDC = (ILPK - IL). Use Schottky diodes for higher conversion efficiency. The reverse voltage rating of the Schottky diode must be higher than the sum of the maximum input voltage (VIN-MAX) and the output voltage. Since the average current flowing through the diode is equal to the output current, choose the diode with forward current rating of IOUT-MAX. The Maxim Integrated 20 MAX15004A/B-MAX15005A/B current sense (RCS) can be calculated using the currentlimit threshold (0.305V) of MAX15004/MAX15005 and ILPK. Use a diode with a forward current rating more than the maximum output current limit if the SEPIC converter needs to be output short-circuit protected. R CS = 0.305 ILPK Select RCS 20% below the value calculated above. Calculate the output current limit using the following equation: D I OUT -LIM = x (ILPK (1 - D) - IL ) where D is the duty cycle at the highest input voltage (VIN-MAX). The series capacitor should be chosen for minimum ripple voltage (VCP) across the capacitor. We recommend using a maximum ripple VCP to be 5% of the minimum input voltage (VIN-MIN) when operating at the minimum input voltage. The multilayer ceramic capacitor X5R and X7R series are recommended due to their high ripple current capability and low ESR. Use the following equation to calculate the series capacitor CP value. I x D MAX CP = OUT -MAX VCP x f OUT 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Power Dissipation The MAX15004/MAX15005 maximum power dissipation depends on the thermal resistance from the die to the ambient environment and the ambient temperature. The thermal resistance depends on the device package, PCB copper area, other thermal mass, and airflow. Calculate the temperature rise of the die using following equation: TJ = TC + (PT x JC) or: TJ = TA + (PT x JA) where JC is the junction-to-case thermal impedance (3C/W) of the 16-pin TSSOP-EP package and PT is power dissipated in the device. Solder the exposed pad of the package to a large copper area to spread heat through the board surface, minimizing the case-toambient thermal impedance. Measure the temperature of the copper area near the device (TC) at worst-case condition of power dissipation and use 3C/W as JC thermal impedance. The case-to-ambient thermal impedance (JA) is dependent on how well the heat is transferred from the PCB to the ambient. Use a large copper area to keep the PCB temperature low. The JA is 38C/W for TSSOP-16-EP and 90C/W for TSSOP-16 package with the condition specified by the JEDEC51 standard for a multilayer board. where VCP is 0.05 x VIN-MIN. For a further discussion of SEPIC converters, go to http://pdfserv.maximintegrated.com/en/an/AN1051. pdf. www.maximintegrated.com Maxim Integrated 21 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers VIN 2.5V TO 16V L1 L11 = L22 = 7.5mH C7 6.8F VOUT D1 LL4148 VOUT (8V/2A) D2 STP745G C4 22F D3 BAT54C 1 IN VCC C1 100nF C1 6.8F 16 C2 6.8F C5 22F C6 22F C3 6.8F CVCC 1F MAX15005A/B 2 ON ON/OFF OFF 3 OUT 4 REG5 RT 15k CT 150pF 6 7 REG5 13 C10 1F N.C. RTCT CS RCS 100 12 CCS 100pF RS 0.025 SGND COMP SYNC 8 14 SLOPE REG5 5 STD20NF06L OVI PGND CSLOPE 47pF RG 1 15 11 R3 1.8k SYNC RSYNC 10k FB SS EP 10 C3 47nF VOUT C4 680pF R2 15k R1 2.7k 9 CSS 150nF Figure 6. SEPIC Application Circuit www.maximintegrated.com Maxim Integrated 22 MAX15004A/B-MAX15005A/B Layout Recommendations Typically, there are two sources of noise emission in a switching power supply: high di/dt loops and high dv/dt surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the heatsink of the MOSFET connected to the device drain presents a dv/dt source; therefore, minimize the surface area of the heatsink as much as possible. Keep all PCB traces carrying switching currents as short as possible to minimize current loops. Use a ground plane for best results. Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Refer to the MAX15005 EV kit data sheet for a specific layout example. Use a multilayer board whenever possible for better noise immunity. Follow these guidelines for good PCB layout: 1) Use a large copper plane under the package and solder it to the exposed pad. To effectively use this copper area as a heat exchanger between the PCB and ambient, expose this copper area on the top and bottom side of the PCB. 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers 3) Isolate the power components and high-current path from the sensitive analog circuitry. 4) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. 5) Connect SGND and PGND together close to the device at the return terminal of VCC bypass capacitor. Do not connect them together anywhere else. 6) Keep the power traces and load connections short. This practice is essential for high efficiency. Use thick copper PCBs (2oz vs. 1oz) to enhance full-load efficiency. 7) Ensure that the feedback connection to FB is short and direct. 8) Route high-speed switching nodes away from the sensitive analog areas. Use an internal PCB layer for SGND as an EMI shield to keep radiated noise away from the device, feedback dividers, and analog bypass capacitors. 9) Connect SYNC pin to SGND when not used. 2) Do not connect the connection from SGND (pin 7) to the EP copper plane underneath the IC. Use midlayer-1 as an SGND plane when using a multilayer board. www.maximintegrated.com Maxim Integrated 23 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Typical Operating Circuits C12 R7 510 220pF VIN (5.5V TO 16V) C1 330F 50V C11 2200pF 100V R16 10 C18 4700pF 100V 1 R17 100k 1% R18 47.5k 1% VCC 2 VIN R11 182k 1% C4 100pF R1 8.45k 1% 5 6 C5 1200pF 7 R19 10k 1 2 VGRID (60V/12mA) C15 22F 60V 16 C3 1F 16V R9 NU C14 NU R10 36k FILAMENT+ (3V/650mA) 15 C16 330F 6.3V R15 100 R3 50 N C17 2.2F 10V OVI 14 FILAMENTD5 REG5 SLOPE REG5 N.C. CS RTCT 13 C10 1F R5 1k 12 C9 560pF R6 0.06 1% SGND COMP 8 D2 D4 PGND 4 R8 100k ON/OFF OUT 3 C13 10F 200V D2 R2 560 D1 MAX15005A/B R12 12.1k 1% REG5 IN C2 0.1mF 50V VANODE (110V/55mA) 11 R2 402k 1% SYNC JU1 FB SS EP 10 C6 4700pF 9 C8 0.1F VANODE C7 47pF R13 118k 1% R14 1.3k 1% Figure 7. VFD Flyback Application Circuit www.maximintegrated.com Maxim Integrated 24 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Typical Operating Circuits (continued) VIN (4.5V TO 16V) C1 10F 25V L1 10H/IHLP5050 VISHAY 1 C11 0.1F IN VCC MAX15005A/B R11 301k 2 VOUT R10 100k R8 153k D1 B340LB OUT 3 4 5 R2 13k 6 C3 180pF 7 REG5 N.C. CS RTCT 2 14 13 C10 1F R3 1k 12 C4 100pF R4 0.025 SGND COMP 1 Q Si736DP REG5 SLOPE SYNC 8 C6 56F/25V SVP-SANYO R1 5 15 VOUT (18V/2A) OVI PGND C2 100pF C10 1F/16V CERAMIC ON/OFF R9 10k REG5 16 11 R5 100k SYNC JU1 FB SS EP 10 C9 0.1F 9 C7 0.1F VOUT C8 330pF R6 136k R7 10k Figure 8. Boost Application Circuit www.maximintegrated.com Maxim Integrated 25 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Pin Configurations TOP VIEW IN 1 + ON/OFF 2 OVI 3 16 VCC IN 1 15 OUT ON/OFF 2 + 16 VCC 15 OUT OVI 3 14 PGND 14 PGND MAX15004A 13 REG5 MAX15005A SLOPE 4 RTCT 6 11 COMP RTCT 6 11 COMP SGND 7 10 FB SGND 7 10 FB SYNC 8 9 SS SYNC 8 9 SS SLOPE 4 N.C. 5 12 CS EP N.C. 5 PROCESS: BiCMOS 12 CS Package Information For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a "+", "#", or "-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE www.maximintegrated.com 13 REG5 TSSOP TSSOP-EP Chip Information MAX15004B MAX15005B PACKAGE CODE OUTLINE NO. LAND PATTERN NO. 16 TSSOP U16+2 21-0066 90-0117 16 TSSOP-EP U16E+3 21-0108 90-0120 Maxim Integrated 26 MAX15004A/B-MAX15005A/B 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers Revision History REVISION NUMBER REVISION DATE PAGES CHANGED 0 1/07 Initial release 1 11/07 Updated Features, revised equations on pages 13, 20, and 21, revised Figure 8 with correct MOSFET, and updated package outline 1, 13, 20, 21, 25, 28 2 12/10 Added MAX15005BAUE/V+ automotive part, updated Features, updated Package Information, style edits 1-5, 9, 13, 21, 25-29 3 1/11 Added MAX15004AAUE/V+, MAX15004BAUE/V+, MAX15005AAUE/V+ automotive parts to the Ordering Information 1 4 1/15 Updated Benefits and Features section 1 5 9/15 Miscellaneous updates 6 12/15 Deleted last sentence in the Startup Operation/UVLO/ON/OFF section 12 7 2/17 Corrected fOSC formula and moved section to page 12 13 DESCRIPTION -- 1, 6, 9-11, 14-16, 18, 20-22 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated's website at www.maximintegrated.com. Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc. (c) 2017 Maxim Integrated Products, Inc. 27 Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Maxim Integrated: MAX15004AAUE+ MAX15004AAUE+T MAX15004BAUE+ MAX15004BAUE+T MAX15005AAUE+ MAX15005AAUE+T MAX15005AAUE/V+ MAX15005AAUE/V+T MAX15005BAUE+ MAX15005BAUE+T MAX15005BAUE/V+ MAX15005BAUE/V+T MAX15004AAUE/V+ MAX15004AAUE/V+T MAX15004BAUE/V+ MAX15004BAUE/V+T