Zero-Drift, Single-Supply Rail-to-Rail Input/Output Operational Amplifier AD8628 FEATURES Lowest Auto-Zero Amplifier Noise Low Offset Voltage: 1 V Input Offset Drift: 0.02 V/C Rail-to-Rail Input and Output Swing 5 V Single-Supply Operation High Gain, CMRR, and PSRR: 120 dB Very Low Input Bias Current: 100 pA Low Supply Current: 1.0 mA Overload Recovery Time: 10 s No External Components Required APPLICATIONS Automotive Sensors Pressure and Position Sensors Strain Gage Amplifiers Medical Instrumentation Thermocouple Amplifiers Precision Current Sensing Photodiode Amplifier PIN CONFIGURATIONS 5-Lead SOT-23 (RJ Suffix) 5 V+ OUT 1 V- 2 AD8628 4 -IN +IN 3 8-Lead SOIC (R Suffix) 8 NC NC 1 -IN 2 +IN 3 V- 4 AD8628 7 V+ 6 OUT 5 NC NC = NO CONNECT GENERAL DESCRIPTION This new breed of amplifier has ultralow offset, drift, and bias current. The AD8628 is a wide bandwidth auto-zero amplifier featuring rail-to-rail input and output swings and low noise. Operation is fully specified from 2.7 V to 5 V single supply ( 1.35 V to 2.5 V dual supply). The AD8628 family provides benefits previously found only in expensive auto-zeroing or chopper-stabilized amplifiers. Using Analog Devices' new topology, these zero-drift amplifiers combine low cost with high accuracy and low noise. (No external capacitors are required.) In addition, the AD8628 greatly reduces the digital switching noise found in most chopper-stabilized amplifiers. With an offset voltage of only 1 mV, drift less than 0.005 mV/C, and noise of only 0.5 mV p-p (0 Hz to 10 Hz), the AD8628 is perfectly suited for applications where error sources cannot be tolerated. Position and pressure sensors, medical equipment, and strain gage amplifiers benefit greatly from nearly zero drift over their operating temperature range. Many systems can take advantage of the rail-to-rail input and output swings provided by the AD8628 family to reduce input biasing complexity and maximize SNR. The AD8628 family is specified for the extended industrial temperature range (-40C to +125C). The AD8628 amplifier is available in the tiny SOT-23 and the popular 8-lead narrow SOIC plastic packages. The SOT-23 package devices are available only in tape and reel. REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective companies. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 (c) 2003 Analog Devices, Inc. All rights reserved. AD8628-SPECIFICATIONS ELECTRICAL CHARACTERISTICS (V = 5.0 V, V S Parameter Symbol INPUT CHARACTERISTICS Offset Voltage VOS Input Bias Current IB Input Offset Current IOS Input Voltage Range Common-Mode Rejection Ratio CMRR Large Signal Voltage Gain* AVO Offset Voltage Drift DVOS/DT OUTPUT CHARACTERISTICS Output Voltage High VOH Output Voltage Low VOL Short-Circuit Limit ISC Output Current IO POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier INPUT CAPACITANCE Differential Common Mode DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density PSRR ISY CM = 2.5 V, TA = 25C, unless otherwise noted.) Conditions -40C TA +125C -40C TA +125C VCM = 0 V to 5 V -40C TA +125C RL = 10 kW, VO = 0.3 V to 4.7 V -40C TA +125C -40C TA +125C RL = 100 kW to Ground -40C TA +125C RL = 10 kW to Ground -40C TA +125C RL = 100 kW to V+ -40C TA +125C RL = 10 kW to V+ -40C TA +125C -40C TA +125C Unit 1 5 10 100 1.5 200 250 5 140 130 145 135 0.002 0.02 mV mV pA nA pA pA V dB dB dB dB mV/C 0 120 115 125 120 4.99 4.99 4.95 4.95 4.996 4.995 4.98 4.97 1 2 10 15 50 40 30 15 V V V V mV mV mV mV mA mA mA mA 25 115 130 0.85 1.0 5 5 20 20 1.1 1.2 dB mA mA 1.5 10 pF pF RL = 10 kW 1.0 0.05 2.5 V/ms ms MHz 0.1 Hz to 10 Hz 0.1 Hz to 1.0 Hz f = 1 kHz f = 10 Hz 0.5 0.16 22 5 mV p-p mV p-p nV//Hz fA//Hz GBP en p-p en p-p en in Max 50 -40C TA +125C VS = 2.7 V to 5.5 V -40C TA +125C VO = 0 V -40C TA +125C Typ 30 -40C TA +125C CIN SR Min *Gain testing is highly dependent upon test bandwidth. Specifications subject to change without notice. -2- REV. A AD8628 ELECTRICAL CHARACTERISTICS (V = 2.7 V, V S Parameter Symbol INPUT CHARACTERISTICS Offset Voltage VOS Input Bias Current IB Input Offset Current IOS Input Voltage Range Common-Mode Rejection Ratio CMRR Large Signal Voltage Gain AVO Offset Voltage Drift DVOS/DT OUTPUT CHARACTERISTICS Output Voltage High VOH Output Voltage Low VOL Short-Circuit Limit ISC Output Current IO POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier INPUT CAPACITANCE Differential Common Mode DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density PSRR ISY CM = 1.35 V, VO = 1.4 V, TA = 25C, unless otherwise noted.) Conditions -40C TA +125C -40C TA +125C VCM = 0 V to 2.7 V -40C TA +125C RL = 10 kW , VO = 0.3 V to 2.4 V -40C TA +125C -40C TA +125C RL = 100 kW to Ground -40C TA +125C RL = 10 kW to Ground -40C TA +125C RL = 100 kW to V+ -40C TA +125C RL = 10 kW to V+ -40C TA +125C -40C TA +125C 1 5 10 100 1.5 200 250 5 2.68 2.68 2.67 2.67 10 115 Unit 130 120 140 130 0.002 0.02 mV mV pA nA pA pA V dB dB dB dB mV/C 2.695 2.695 2.68 2.675 1 2 10 15 15 10 10 5 V V V V mV mV mV mV mA mA mA mA 130 0.75 0.9 5 5 20 20 1.0 1.2 dB mA mA pF pF RL = 10 kW 1 0.05 2 V/ms ms MHz 0.1 Hz to 10 Hz f = 1 kHz f = 10 Hz 0.5 22 5 mV p-p nV//Hz fA//Hz Specifications subject to change without notice. REV. A Max 1.5 10 GBP en p-p en in 0 115 110 110 105 -40C TA +125C VS = 2.7 V to 5.5 V -40C TA +125C VO = 0 V -40C TA +125C Typ 30 1.0 50 -40C TA +125C CIN SR Min -3- AD8628 ABSOLUTE MAXIMUM RATINGS 1 Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V Input Voltage . . . . . . . . . . . . . . . . GND - 0.3 V to VS- + 0.3 V Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . 5.0 V Output Short-Circuit Duration to GND . . . . . . . . . . Indefinite Storage Temperature Range R, RJ Packages . . . . . . . . . . . . . . . . . . . . . -65C to +150C Operating Temperature Range . . . . . . . . . . -40C to +125C Junction Temperature Range R, RJ Packages . . . . . . . . . . . . . . . . . . . . . -65C to +150C Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300C Package Type JA* JC Unit 5-Lead SOT-23 (RT-5) 8-Lead SOIC (R) 230 158 146 43 C/W C/W *JA is specified for worst-case conditions, i.e., JA is specified for device soldered in circuit board for surface-mount packages. 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Differential input voltage is limited to 5 V or the supply voltage, whichever is less. ORDERING GUIDE Model Temperature Range Package Description Package Option AD8628ART-R2 AD8628ART-REEL7 AD8628AR AD8628AR-REEL AD8628AR-REEL7 -40C to +125C -40C to +125C -40C to +125C -40C to +125C -40C to +125C 5-Lead SOT-23 5-Lead SOT-23 8-Lead SOIC 8-Lead SOIC 8-Lead SOIC RJ-5 RJ-5 R-8 R-8 R-8 Branding AYA AYA CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD8628 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. -4- REV. A Typical Performance Characteristics-AD8628 180 60 1,500 VS = 2.7V TA = 25C VS = 5V 140 120 100 80 60 40 50 40 30 20 +25C 10 -1.5 -0.5 0.5 1.5 INPUT OFFSET VOLTAGE (V) 0 2.5 TPC 1. Input Offset Voltage Distribution at 2.7 V 1 2 3 4 5 INPUT COMMON-MODE VOLTAGE (V) 125C 500 0 -500 -1,500 6 TPC 2. Input Bias Current vs. Input Common-Mode Voltage at 5 V VS = 5V VCM = 2.5V TA = 25C 70 60 50 40 30 20 6 VS = 5V TA = 25C OUTPUT VOLTAGE (mV) 80 1 2 3 4 5 INPUT COMMON-MODE VOLTAGE (V) 1k VS = 5V TA = -40C TO +125 C 6 NUMBER OF AMPLIFIERS 90 0 TPC 3. Input Bias Current vs. Input Common-Mode Voltage at 5 V 7 100 NUMBER OF AMPLIFIERS 0 150C 1,000 -1,000 -40C 20 0 -2.5 VS = 5V +85C INPUT BIAS CURRENT (pA) INPUT BIAS CURRENT (pA) NUMBER OF AMPLIFIERS 160 5 4 3 2 100 10 SOURCE SINK 1 0.1 1 10 1k INPUT BIAS CURRENT (pA) OUTPUT VOLTAGE (mV) 10 SOURCE SINK 1 0.1 0.01 0.1 1 0.001 LOAD CURRENT (mA) 10 TPC 7. Output Voltage to Supply Rail vs. Load Current at 2.7 V REV. A 4 6 TCVOS (nV/C) 8 10 0.1 1 0.001 0.01 LOAD CURRENT (mA) 10 TPC 5. Input Offset Voltage Drift TPC 6. Output Voltage to Supply Rail vs. Load Current at 5 V 1,500 1,250 VS = 2.7V 100 2 0 .5 TPC 4. Input Offset Voltage Distribution at 5 V 0.01 0.0001 0.01 0.0001 0 -1.5 -0.5 2 0.5 1.5 INPUT OFFSET VOLTAGE (V) 1,150 VS = 5V VCM = 2.5V TA = -40C TO +150 C 800 450 100 0 -50 -25 TA = 25C 5V SUPPLY CURRENT (A) 0 -2.5 1,000 2.7V 750 500 250 0 25 50 75 100 125 150 175 TEMPERATURE (C) TPC 8. Input Bias Current vs. Temperature -5- 0 -50 0 50 100 150 TEMPERATURE (C) TPC 9. Supply Current vs. Temperature 200 AD8628 70 70 VS = 2.7V CL = 20pF RL = M = 52.1 50 600 400 200 40 0 30 45 20 90 10 135 0 180 10 225 0 1 2 3 4 SUPPLY VOLTAGE (V) 30 10k 6 5 TPC 10. Supply Current vs. Supply Voltage 45 20 90 10 135 0 180 10 225 100k 1M FREQUENCY (Hz) 30 10k 10M AV = 100 30 20 AV = 10 10 0 AV = 1 -10 50 30 AV = 10 20 10 -10 -20 -30 100k 1M FREQUENCY (Hz) 10M AV = 1 0 -20 10k AV = 100 240 180 AV = 100 150 120 90 AV = 10 60 1k 10k 100k 1M FREQUENCY (Hz) 0 100 10M 1M 10k 100k FREQUENCY (Hz) 1k 10M 100M TPC 15. Output Impedance vs. Frequency at 2.7 V TPC 14. Closed-Loop Gain vs. Frequency at 5 V VS = 5V AV = 1 210 30 TPC 13. Closed-Loop Gain vs. Frequency at 2.7 V 300 VS = 2.7V 270 40 -30 1k 10M 300 VS = 5V CL = 20pF RL = 2k 60 40 100k 1M FREQUENCY (Hz) TPC 12. Open-Loop Gain and Phase vs. Frequency OUTPUT IMPEDANCE () 50 CLOSED-LOOP GAIN (dB) CLOSED-LOOP GAIN (dB) 0 30 70 VS = 2.7V CL = 20pF RL = 2k 60 0 0 0 0 0 0 AV = 1 210 180 AV = 100 150 120 90 AV = 10 60 30 0 100 1k 1M 10k 100k FREQUENCY (Hz) 10M TPC 16. Output Impedance vs. Frequency at 5 V 100M VOLTAGE (1V/DIV) 240 VOLTAGE (500mV/DIV) OUTPUT IMPEDANCE () 40 TPC 11. Open-Loop Gain and Phase vs. Frequency 70 270 50 20 20 0 VS = 5V CL = 20pF RL = M = 52.1 60 OPEN-LOOP GAIN (dB) 800 OPEN-LOOP GAIN (dB) SUPPLY CURRENT (A) 60 PHASE SHIFT (Degrees) TA = 25C VS = 1.35V CL = 300pF RL = AV = 1 0 0 0 VS = 2.5V CL = 300pF RL = AV = 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 TIME (4s/DIV) 0 0 TPC 17. Large Signal Transient Response at 2.7 V -6- 0 0 0 0 0 0 0 0 0 TIME (5s/DIV) 0 0 0 TPC 18. Large Signal Transient Response at 5 V REV. A PHASE SHIFT (Degrees) 1,000 AD8628 0 0 VS = 1.35V CL = 50pF RL = AV = 1 0 0 0 0 0 VS = 1.35V RL = 2k TA = 25C 90 80 OVERSHOOT (%) 0 0 VOLTAGE (50mV/DIV) VOLTAGE (50mV/DIV) 0 100 VS = 2.5V CL = 50pF RL = AV = 1 0 0 0 70 60 OS- 50 40 OS+ 30 0 20 0 0 0 0 0 0 0 0 0 0 0 0 TIME (4s/DIV) 0 0 0 0 0 TPC 19. Small Signal Transient Response at 2.7 V 0 0 0 0 0 0 0 TIME (4s/DIV) 0 0 0 0 70 0 VIN OS- 0 VOLTAGE (V) 60 50 OS+ 40 30 0V0 0 0V0 VIN 0 0 VOUT 0 0 0V0 10 100 CAPACITIVE LOAD (pF) 1k 0 0 0 0 0 0 0 0 0 TIME (2s/DIV) 0 0 0 0 TPC 23. Positive Overvoltage Recovery CMRR (dB) 0 0 0 0 0 0 0 0 TIME (10s/DIV) 0 0 0 140 VS = 2.7V 0 0 TPC 24. Negative Overvoltage Recovery 140 VS = 2.5V VIN = 1kHz @ 3V p-p CL = 0pF RL = 10k AV = 1 0 0 0 120 120 100 100 80 80 60 60 CMRR (dB) 1 VOUT 0 TPC 22. Small Signal Overshoot vs. Load Capacitance at 5 V VOLTAGE (1V/DIV) VS = 2.5V AV = -50 RL = 10k CL = 0 CH1 = 50mV/DIV CH2 = 1V/DIV 0 10 0 1k 0V0 0 20 10 100 CAPACITIVE LOAD (pF) 0 VS = 2.5V AV = -50 RL = 10k CL = 0 CH1 = 50mV/DIV CH2 = 1V/DIV VOLTAGE (V) VS = 2.5V RL = 2k TA = 25C 1 TPC 21. Small Signal Overshoot vs. Load Capacitance at 2.7 V TPC 20. Small Signal Transient Response at 5 V 80 OVERSHOOT (%) 10 40 20 VS = 5V 40 20 0 0 -20 -20 -40 -40 0 0 0 0 0 0 0 0 0 0 0 TIME (200s/DIV) 0 0 TPC 25. No Phase Reversal REV. A 0 -60 100 1k 10k 100k FREQUENCY (Hz) 1M 10M TPC 26. CMRR vs. Frequency at 2.7 V -7- -60 100 1k 10k 100k FREQUENCY (Hz) 1M 10M TPC 27. CMRR vs. Frequency at 5 V AD8628 140 140 VS = 1.35V 120 100 80 60 PSRR (dB) +PSRR 40 PSRR 20 0 60 +PSRR 40 PSRR 20 0 -20 -20 -40 -60 100 1k 10k 100k FREQUENCY (Hz) 1M 1.5 1.0 0.5 -60 100 10M TPC 28. PSRR vs. Frequency 1k 10k 100k FREQUENCY (Hz) 1M 0 100 10M 0.60 4.0 3.5 VOLTAGE (V) VS = 5V RL = 10k TA = 25C AV = 1 3.0 2.5 2.0 1.5 0 100 1k 10k 100k FREQUENCY (Hz) 0.45 0.30 0.30 0.15 0 -0.15 -0.30 -0.45 1 2 3 4 5 6 TIME (s) 7 8 9 -0.60 0 10 75 60 45 30 15 0 0.5 1.0 1.5 FREQUENCY (kHz) 2.0 TPC 34. Voltage Noise Density at 2.7 V from 0 Hz to 2.5 kHz 2.5 2 3 4 5 6 TIME (s) 7 8 9 10 120 VS = 2.7V NOISE AT 10kHz = 42.4nV VOLTAGE NOISE DENSITY (nV/ Hz) VOLTAGE NOISE DENSITY (nV/ Hz) 90 105 1 TPC 33. 0.1 Hz to 10 Hz Noise at 5 V 120 VS = 2.7V NOISE AT 1kHz = 21.3nV 0 -0.15 TPC 32. 0.1 Hz to 10 Hz Noise at 2.7 V 120 105 0 -0.45 0 TPC 31. Maximum Output Swing vs. Frequency at 5 V 0.15 -0.30 -0.60 1M 1M VS = 5V 0.45 1.0 0.5 10k 100k FREQUENCY (H) 0.60 VS = 2.7V 4.5 1k TPC 30. Maximum Output Swing vs. Frequency TPC 29. PSRR vs. Frequency 5.5 5.0 VOLTAGE NOISE DENSITY (nV/ Hz) 2.5 VS = 2.7V RL = 10k TA = 25C 2.0 AV = 1 -40 VOLTAGE (V) PSRR (dB) OUTPUT SWING (V p-p) 100 80 OUTPUT SWING (V p-p) 3.0 VS = 2.5V 120 90 75 60 45 30 15 105 VS = 5V NOISE AT 1kHz = 22.1nV 90 75 60 45 30 15 0 5 10 15 FREQUENCY (kHz) 20 TPC 35. Voltage Noise Density at 2.7 V from 0 Hz to 25 kHz -8- 25 0.5 1.0 1.5 FREQUENCY (kHz) 2.0 2.5 TPC 36. Voltage Noise Density at 5 V from 0 Hz to 2.5 kHz REV. A AD8628 120 105 90 75 60 45 30 15 105 90 75 60 45 30 15 130 120 5 10 15 FREQUENCY - kHz 20 100 90 80 70 60 0 25 150 VS = 2.7V TA = -40C TO +150 C 100 50 ISC- 0 ISC+ -50 -100 -50 -25 0 25 50 75 100 125 150 175 TEMPERATURE (C) TPC 40. Output Short-Circuit Current vs. Temperature 5 FREQUENCY - Hz 100 ISC- 50 0 -50 0 25 50 75 TEMPERATURE (C) 100 125 VS = 5V VCC - V OH @ 1k 100 VOL - V EE @ 1k VCC - V OH @ 10k 10 VOL - V EE @ 10k VCC - V OH @ 100k 1 VOL - V EE @ 100k ISC+ -100 -50 -25 0 25 50 75 100 125 150 175 TEMPERATURE (C) TPC 41. Output Short-Circuit Current vs. Temperature VCC - V OH @ 1k 100 VOL - V EE @ 1k VCC - V OH @ 10k 10 VOL - V EE @ 10k VCC - V OH @ 100k 1 VOL - V EE @ 100k 25 50 75 100 125 150 175 TEMPERATURE (C) TPC 43. Output-to-Rail Voltage vs. Temperature REV. A -25 TPC 39. Power Supply Rejection vs. Temperature 1k VS = 5V TA = -40C TO +150 C VS = 2.7V 0 10 150 1k 0.10 -50 -25 50 -50 TPC 38. Voltage Noise OUTPUT SHORT-CIRCUIT CURRENT (mA) TPC 37. Voltage Noise Density at 5 V from 0 Hz to 25 kHz VS = 2.7V TO 5V TA = -40C TO +125 C 100 OUTPUT-TO-RAIL VOLTAGE (mV) 0 OUTPUT-TO-RAIL VOLTAGE (mV) 140 0 0 OUTPUT SHORT-CIRCUIT CURRENT (mA) 150 VS = 5V POWER SUPPLY REJECTION (dB) VS = 5V NOISE AT 10kHz = 36.4nV VOLTAGE NOISE DENSITY (nV/ Hz) VOLTAGE NOISE DENSITY (nV/ Hz) 120 -9- 0.10 -50 -25 0 25 50 75 100 125 150 175 TEMPERATURE (C) TPC 42. Output-to-Rail Voltage vs. Temperature AD8628 FUNCTIONAL DESCRIPTION 1/f Noise The AD8628 is a single-supply, ultrahigh precision rail-to-rail input and output operational amplifier. The typical offset voltage of less than 1 mV allows this amplifier to be easily configured for high gains without risk of excessive output voltage errors. The extremely small temperature drift of 2 nV/C ensures a minimum of offset voltage error over its entire temperature range of -40C to +125C, making the AD8628 amplifier ideal for a variety of sensitive measurement applications in harsh operating environments. The AD8628 achieves a high degree of precision through a patented combination of auto-zeroing and chopping. This unique topology allows the AD8628 to maintain its low offset voltage over a wide temperature range and over its operating lifetime. AD8628 also optimizes the noise and bandwidth over previous generations of auto-zero amplifiers, offering the lowest voltage noise of any auto-zero amplifier by more than 50%. 1/f noise, also known as "pink noise," is a major contributor of errors in dc-coupled measurements. This 1/f noise error term can be in the range of several mV or more, and, when amplified with the closed-loop gain of the circuit, can show up as a large output offset. For example, when an amplifier with a 5 mV p-p 1/f noise is configured for a gain of 1,000, its output will have 5 mV of error due to the 1/f noise. But AD8628 eliminates 1/f noise internally and therefore greatly reduces output errors. Here is how it works: 1/f noise appears as a slowly varying offset to AD8628 inputs. Auto-zeroing corrects any dc or low frequency offset, thus the 1/f noise component is essentially removed, leaving AD8628 free of 1/f noise. 120 VOLTAGE NOISE DENSITY (nV/ Hz) 105 Previous designs used either auto-zeroing or chopping to add precision to the specifications of an amplifier. Auto-zeroing results in low noise energy at the auto-zeroing frequency at the expense of higher low frequency noise due to aliasing of wideband noise into the auto-zeroed frequency band. Chopping results in lower low frequency noise at the expense of larger noise energy at the chopping frequency. AD8628 uses both auto-zeroing and chopping in a patented ping-pong arrangement to obtain lower low frequency noise together with lower energy at the chopping and auto-zeroing frequencies, maximizing the signal-to-noise ratio (SNR) for the majority of applications without the need for additional filtering. The relatively high clock frequency of 15 kHz simplifies filter requirements for a wide, useful, noisefree bandwidth. AD8628 is one of the few auto-zero amplifiers offered in the 5-lead SOT-23 package. It greatly improves the ac parameters of the previous auto-zero amplifiers. It has low noise over a relatively wide bandwidth (0 Hz to 10 kHz) and can be used where the highest dc precision is required. In systems with signal bandwidths up to 5 kHz to 10 kHz, the AD8628 provides true 16-bit accuracy making it the best choice for very high resolution systems. LTC2050 (89.7nV/ Hz) 90 75 60 45 LMC2001 (31.1nV/ Hz) 30 15 0 AD8628 (19.4nV/ Hz) 0 2 MK AT 1kHz FOR ALL 3 GRAPHS 6 4 FREQUENCY (kHz) 8 10 12 Figure 1. Noise Spectral Density of AD8628 vs. Competition One of the biggest advantages that AD8628 brings to systems applications over competitive auto-zero amplifiers is its very low noise. The comparison shown in Figure 1 indicates an inputreferred noise density of 19.4 nV//Hz at 1 kHz for AD8628 that is much better than the LTC2050 and LMC2001. The noise is flat from dc to 1.5 kHz, slowly increasing up to 20 kHz. The lower noise at low frequency is desirable where auto-zero amplifiers are widely used. -10- REV. A AD8628 50 AD8628 Peak-to-Peak Noise vs. Competition Because of the ping-pong action between auto-zeroing and chopping, the peak-to-peak noise of the AD8628 is much lower than its competition. Figures 2 and 3 show this comparison. 45 40 35 NOISE (dB) 0 en p-p = 0.5V BW = 0.1Hz to 10Hz 0 VOLTAGE (0.5V/DIV) 0 30 25 20 15 0 10 0 5 0 10 0 20 30 40 50 60 FREQUENCY (Hz) 70 80 90 100 Figure 5a. Simulation Transfer Function of Test Circuit 0 50 0 0 0 0 0 0 0 0 TIME (1s/DIV) 0 0 0 0 45 Figure 2. AD8628 Peak-to-Peak Noise 40 35 NOISE (dB) 0 en p-p = 2.3V BW = 0.1Hz to 10Hz 0 VOLTAGE (0.5V/DIV) 0 30 25 20 15 0 10 0 5 0 0 FREQUENCY (kHz) Figure 5b. Actual Transfer Function of Test Circuit 0 Measured noise spectrum of test circuit showing noise between 5 kHz and 45 kHz is successfully rolled off by first order filter. 0 0 0 0 0 0 0 0 TIME (1s/DIV) 0 0 0 0 Total Integrated Input-Referred Noise for First Order Filter (AD8628 vs. Competition) Figure 3. LTC2050 Peak-to-Peak Noise Noise Behavior with First Order Low-Pass Filter 10 AD8628 was simulated as a low-pass filter and then configured as shown in Figure 4. The behavior of the AD8628 matches the simulated data. It was verified that noise is rolled off by first order filtering. RMS NOISE (V) LTC2050 IN OUT 100k 470pF AD8551 AD8628 1 1k Figure 4. Test Circuit: First Order Low-Pass Filter--x101 Gain and 3 kHz Corner Frequency 0.1 10 100 1k 3dB FILTER BANDWIDTH (Hz) 10k Figure 6. 3 dB Filter Bandwidth in Hz For a first order filter, the total integrated noise from the AD8628 is lower than the LTC2050. REV. A -11- AD8628 0 Input Overvoltage Protection CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 0 VIN 0 0 0V VOLTAGE ( V) Although the AD8628 is a rail-to-rail input amplifier, care should be taken to ensure that the potential difference between the inputs does not exceed the supply voltage. Under normal negative feedback operating conditions, the amplifier will correct its output to ensure the two inputs are at the same voltage. However, if either input exceeds either supply rail by more than 0.3 V, large currents will begin to flow through the ESD protection diodes in the amplifier. These diodes are connected between the inputs and each supply rail to protect the input transistors against an electrostatic discharge event and are normally reverse biased. However, if the input voltage exceeds the supply voltage, these ESD diodes will become forward biased. Without current limiting, excessive amounts of current could flow through these diodes, causing permanent damage to the device. If inputs are subject to overvoltage, appropriate series resistors should be inserted to limit the diode current to less than 5 mA maximum. 0 0V 0 0 0 VOUT 0 0 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 Figure 7. Positive Input Overload Recovery for AD8628 0 Output Phase Reversal Overload Recovery Time CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 0 VIN 0 VOLTAGE ( V) Output phase reversal occurs in some amplifiers when the input common-mode voltage range is exceeded. As common-mode voltage is moved outside of the common-mode range, the outputs of these amplifiers will suddenly jump in the opposite direction to the supply rail. This is the result of the differential input pair shutting down, causing a radical shifting of internal voltages that results in the erratic output behavior. The AD8628 amplifier has been carefully designed to prevent any output phase reversal, provided both inputs are maintained within the supply voltages. If one or both inputs could exceed either supply voltage, a resistor should be placed in series with the input to limit the current to less than 5 mA. This will ensure the output will not reverse its phase. 0V 0 0 0V 0 0 0 VOUT 0 0 Many auto-zero amplifiers are plagued by long overload recovery time, often in milliseconds, due to the complicated settling behavior of the internal nulling loops after saturation of the outputs. AD8628 has been designed so that internal settling occurs within two clock cycles after output saturation happens. This results in a much shorter recovery time, less than 10 ms, when compared to other auto-zero amplifiers. The wide bandwidth of the AD8628 enhances performance when it is used to drive loads that inject transients into the outputs. This is a common situation when an amplifier is used to drive the input of switched capacitor ADCs. 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 Figure 8. Positive Input Overload Recovery for LTC2050 0 CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 VIN 0 VOLTAGE (V) 0 0V 0 0 0V 0 0 0 VOUT 0 0 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 Figure 9. Positive Input Overload Recovery for LMC2001 -12- REV. A AD8628 0 The results shown in Figures 7-12 are summarized in Table I. 0V 0 VOLTAGE (V) Table I. Overload Recovery Time CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 0 VIN 0 0 VOUT 0 0 Positive Overload Recovery (s) Negative Overload Recovery (s) AD8628 LTC2050 LMC2001 6 650 40,000 9 25,000 35,000 Infrared Sensors 0V 0 0 0 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 Figure 10. Negative Input Overload Recovery for AD8628 0 0V CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 0 VIN 0 VOUT 0 VOLTAGE (V) Product Type Recovery 0 0V 0 Infrared (IR) sensors, particularly thermopiles, are increasingly being used in temperature measurement for applications as wideranging as automotive climate controls, human ear thermometers, home insulation analysis, and automotive repair diagnostics. The relatively small output signal of the sensor demands high gain with very low offset voltage and drift to avoid dc errors. If interstage ac coupling is used (Figure 13), low offset and drift prevents the input amplifier's output from drifting close to saturation. The low input bias currents generate minimal errors from the sensor's output impedance. As with pressure sensors, the very low amplifier drift with time and temperature eliminates additional errors once the temperature measurement has been calibrated. The low 1/f noise improves SNR for dc measurements taken over periods often exceeding 1/5 second. Figure 15 shows a circuit that can amplify ac signals from 100 mV to 300 mV up to the 1 V to 3 V level, gain of 10,000 for accurate A/D conversion. 0 10k 100 0 100k 100k 5V 5V 0 0 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 100V - 300V Figure 11. Negative Input Overload Recovery for LTC2050 IR DETECTOR 10F AD8628 AD8628 fC 10k 1.6Hz 0 TO BIAS VOLTAGE 0V 0 CH 1 = 50mV/DIV CH 2 = 1V/DIV AV = -50 VOLTAGE ( V) 0 Figure 13. Preamplifier for Thermopile VIN 0 VOUT 0 0 0 0V 0 0 0 0 0 0 0 0 0 TIME (500s/DIV) 0 0 0 0 Figure 12. Negative Input Overload Recovery for LMC2001 REV. A -13- AD8628 Precision Current Shunts Output Amplifier for High Precision DACs A precision shunt current sensor benefits from the unique attributes of auto-zero amplifiers when used in a differencing configuration (Figure 14). Shunt current sensors are used in precision current sources for feedback control systems. They are also used in a variety of other applications, including battery fuel gauging, laser diode power measurement and control, torque feedback controls in electric power steering, and precision power metering. AD8628 is used as an output amplifier for a 16-bit high precision DAC in unipolar configuration. In this case, the selected op amp needs to have very low offset voltage (the DAC LSB is 38 mV when operated with a 2.5 V reference) to eliminate the need for output offset trims. Input bias current (typically a few tens of pico amp) must also be very low since it generates an additional zero code error when multiplied by the DAC output impedance (approximately 6 kW). Rail-to-rail input and output provide full-scale output with very little error. Output impedance of the DAC is constant and code-independent, but the high input impedance of the AD8628 minimizes gain errors. The amplifier's wide bandwidth also serves well in this case. The amplifier with settling time of 1 ms adds another time constant to the system, increasing the settling time of the output. The settling time of the AD5541 is 1 ms. The combined settling time is approximately 1.4 ms, as can be derived from the equation: SUPPLY I 100k e = 1,000 RS I 100mV/mA RS 0.1 RL 100 C 5V AD8628 100k tS (TOTAL ) = 100 (tS DAC ) + (tS AD8628) 2 2 C 5V 2.5V Figure 14. Low-Side Current Sensing In such applications, it is desirable to use a shunt with very low resistance to minimize the series voltage drop; this minimizes wasted power and allows the measurement of high currents without saving power. A typical shunt might be 0.1 W. At measured current values of 1 A, the shunt's output signal is hundreds of millivolts, or even volts, and amplifier error sources are not critical. However, at low measured current values in the 1 mA range, the 100 mV output voltage of the shunt demands a very low offset voltage and drift to maintain absolute accuracy. Low input bias currents are also needed, so that "injected" bias current does not become a significant percentage of the measured current. High open-loop gain, CMRR, and PSRR all help to maintain the overall circuit accuracy. As long as the rate of change of the current is not too fast, an auto-zero amplifier can be used with excellent results. 0.1F SERIAL INTERFACE 0.1F VDD CS DIN 10F REF(REF*) REFS* AD5541/AD5542 LDAC* OUT AD8628 SCLK UNIPOLAR OUTPUT DGND AGND *AD5542 ONLY Figure 15. AD8628 Used as an Output Amplifier -14- REV. A AD8628 OUTLINE DIMENSIONS 5-Lead Small Outline Transistor Package [SOT-23] (RT-5) Dimensions shown in millimeters 2.90 BSC 5 4 2.80 BSC 1.60 BSC 1 2 3 PIN 1 0.95 BSC 1.90 BSC 1.30 1.15 0.90 1.45 MAX 0.15 MAX 0.50 0.35 SEATING PLANE 0.22 0.08 10 5 0 0.60 0.45 0.30 COMPLIANT TO JEDEC STANDARDS MO-178AA 8-Lead Standard Small Outline Package [SOIC] (R-8) Dimensions shown in millimeters and (inches) 5.00 (0.1968) 4.80 (0.1890) 4.00 (0.1574) 3.80 (0.1497) 8 5 1 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY SEATING 0.10 PLANE 6.20 (0.2440) 5.80 (0.2284) 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) 0.50 (0.0196) 45 0.25 (0.0099) 8 0.25 (0.0098) 0 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067) COMPLIANT TO JEDEC STANDARDS MS-012AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN REV. A -15- AD8628 REVISION HISTORY Location Page 6/03--Data Sheet changed from Rev. 0 to Rev. A Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Change to FUNCTIONAL DESCRIPTION section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 -16- REV. A C02735-0-6/03(A) Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3