LMV793MA/NOPB - NATIONAL SEMICONDUCTOR

LMV793,LMV794

LMV793/LMV794 88 MHz, Low Noise, 1.8V CMOS Input, Decompensated

Operational Amplifiers

Literature Number: SNOSAX6C

June 23, 2008

LMV793/LMV794

88 MHz, Low Noise, 1.8V CMOS Input, Decompensated

Operational Amplifiers

General Description

The LMV793 (single) and the LMV794 (dual) CMOS input

operational amplifiers offer a low input voltage noise density

of 5.8 nV/ while consuming only 1.15 mA (LMV793) of

quiescent current. The LMV793/LMV794 are stable at a gain

of 10 and have a gain bandwidth product (GBW) of 88 MHz.

The LMV793/LMV794 have a supply voltage range of 1.8V to

5.5V and can operate from a single supply. The LMV793/

LMV794 each feature a rail-to-rail output stage capable of

driving a 600Ω load and sourcing as much as 60 mA of cur-

rent.

The LMV793/LMV794 provide optimal performance in low

voltage and low noise systems. A CMOS input stage, with

typical input bias currents in the range of a few femto-Am-

peres, and an input common mode voltage range, which

includes ground, make the LMV793/LMV794 ideal for low

power sensor applications where high speeds are needed.

The LMV793/LMV794 are manufactured using National’s ad-

vanced VIP50 process. The LMV793 is offered in either a 5-

Pin SOT23 or an 8-Pin SOIC package. The LMV794 is offered

in either the 8-Pin SOIC or the 8-Pin MSOP.

Features

(Typical 5V supply, unless otherwise noted)

■Input referred voltage noise 5.8 nV/√Hz

■Input bias current 100 fA

■Gain bandwidth product 88 MHz

■Supply current per channel

—LMV793 1.15 mA

—LMV794 1.30 mA

■Rail-to-rail output swing

—@ 10 kΩ load 25 mV from rail

—@ 2 kΩ load 45 mV from rail

■Guaranteed 2.5V and 5.0V performance

■Total harmonic distortion 0.04% @1 kHz, 600Ω

■Temperature range −40°C to 125°C

Applications

■ADC interface

■Photodiode amplifiers

■Active filters and buffers

■Low noise signal processing

■Medical instrumentation

■Sensor interface applications

Typical Application

20216369

Photodiode Transimpedance Amplifier

20216339

Input Referred Voltage Noise vs. Frequency

LMV793/LMV794 88 MHz, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifiers

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

ESD Tolerance (Note 2)

Human Body Model 2000V

Machine Model 200V

Charge-Device Model 1000V

VIN Differential ±0.3V

Supply Voltage (V+ – V−)6.0V

Input/Output Pin Voltage V+ +0.3V, V− −0.3V

Storage Temperature Range −65°C to 150°C

Junction Temperature (Note 3) +150°C

Soldering Information

Infrared or Convection (20 sec) 235°C

Wave Soldering Lead Temp (10 sec) 260°C

Operating Ratings (Note 1)

Temperature Range (Note 3) −40°C to 125°C

Supply Voltage (V+ – V−)

−40°C ≤ TA ≤ 125°C 2.0V to 5.5V

0°C ≤ TA ≤ 125°C 1.8V to 5.5V

Package Thermal Resistance (θJA (Note 3))

5-Pin SOT-23 180°C/W

8-Pin SOIC 190°C/W

8-Pin MSOP 236°C/W

2.5V Electrical Characteristics (Note 4)

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at

the temperature extremes.

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)

Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift

(Note 7)

LMV793 −1.0 μV/°C

LMV794 −1.8

IBInput Bias Current VCM = 1.0V

(Notes 8, 9)

−40°C ≤ TA ≤ 85°C 0.05 1

25 pA

−40°C ≤ TA ≤ 125°C 0.05 1

100

IOS Input Offset Current VCM = 1.0V

(Note 9)

10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 1.4V 80

94 dB

PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

-0.3

1.5

1.5 V

AVOL Open Loop Voltage Gain VOUT = 0.15V to 2.2V,

RL = 2 kΩ to V+/2

LMV793 85

LMV794 82

VOUT = 0.15V to 2.2V,

RL = 10 kΩ to V+/2

110

VOUT Output Voltage Swing High RL = 2 kΩ to V+/2 25 75

mV from

either rail

RL = 10 kΩ to V+/2 20 65

Output Voltage Swing Low RL = 2 kΩ to V+/2 30 75

RL = 10 kΩ to V+/2 15 65

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LMV793/LMV794

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)

Units

IOUT Output Current Sourcing to V−

VIN = 200 mV (Note 10)

Sinking to V+

VIN = –200 mV (Note 10)

ISSupply Current Per Amplifier LMV793 0.95 1.30

1.65 mA

LMV794 1.1 1.50

1.85

SR Slew Rate AV = +10, Rising (10% to 90%) 32 V/μs

AV = +10, Falling (90% to 10%) 24

GBW Gain Bandwidth AV = +10, RL = 10 kΩ 88 MHz

enInput Referred Voltage Noise Density f = 1 kHz 6.2 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 600Ω 0.01 %

5V Electrical Characteristics (Note 4)

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at

the temperature extremes.

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)

Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift

(Note 7)

LMV793 −1.0 μV/°C

LMV794 −1.8

IBInput Bias Current VCM = 2.0V

(Notes 8, 9)

−40°C ≤ TA ≤ 85°C 0.1 1

25 pA

−40°C ≤ TA ≤ 125°C 0.1 1

100

IOS Input Offset Current VCM = 2.0V

(Note 9)

10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 3.7V 80

100 dB

PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

-0.3

AVOL Open Loop Voltage Gain VOUT = 0.3V to 4.7V,

RL = 2 kΩ to V+/2

LMV793 85

LMV794 82

VOUT = 0.3V to 4.7V,

RL = 10 kΩ to V+/2

110

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LMV793/LMV794

Symbol Parameter Conditions Min

(Note 6)

Typ

(Note 5)

Max

(Note 6)

Units

VOUT Output Voltage Swing High RL = 2 kΩ to V+/2 LMV793 35 75

mV from

either rail

LMV794 35 75

RL = 10 kΩ to V+/2 25 65

Output Voltage Swing Low RL = 2 kΩ to V+/2 LMV793 42 75

LMV794 45 80

RL = 10 kΩ to V+/2 20 65

IOUT Output Current Sourcing to V−

VIN = 200 mV (Note 10)

Sinking to V+

VIN = –200 mV (Note 10)

ISSupply Current per Amplifier LMV793 1.15 1.40

1.75 mA

LMV794 per Channel 1.30 1.70

2.05

SR Slew Rate AV = +10, Rising (10% to 90%) 35 V/μs

AV = +10, Falling (90% to 10%) 28

GBW Gain Bandwidth AV = +10, RL = 10 kΩ 88 MHz

enInput Referred Voltage Noise Density f = 1 kHz 5.8 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 600Ω 0.01 %

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics

Tables.

Note 2: Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)

Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).

Note 3: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is

PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.

Note 4: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating

of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ >

TA.

Note 5: Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will

also depend on the application and configuration. The typical values are not tested and are not guaranteed on shipped production material.

Note 6: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the statistical quality

control (SQC) method.

Note 7: Offset voltage average drift is determined by dividing the change in VOS by temperature change.

Note 8: Positive current corresponds to current flowing into the device.

Note 9: This parameter is guaranteed by design and/or characterization and is not tested in production.

Note 10: The short circuit test is a momentary test, the short circuit duration is 1.5 ms.

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LMV793/LMV794

Connection Diagrams

5-Pin SOT-23 (LMV793)

20216301

Top View

8-Pin SOIC (LMV793)

20216385

Top View

8-Pin SOIC/MSOP (LMV794)

20216302

Top View

Ordering Information

Package Part Number Package Marking Transport Media NSC Drawing

5-Pin SOT23 LMV793MF AS4A 1k Units Tape and Reel MF05A

LMV793MFX 3k Units Tape and Reel

8-Pin SOIC

LMV793MA LMV793MA 95 Units/Rail

M08A

LMV793MAX 2.5k Units Tape and Reel

LMV794MA LMV794MA 95 Units/Rail

LMV794MAX 2.5k Units Tape and Reel

8-Pin MSOP LMV794MM AN4A 1k Units Tape and Reel MUA08A

LMV794MMX 3.5k Units Tape and Reel

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LMV793/LMV794

Typical Performance Characteristics Unless otherwise specified, TA = 25°C, V– = 0, V+ = Supply Voltage

= 5V, VCM = V+/2.

Supply Current vs. Supply Voltage (LMV793)

20216305

Supply Current vs. Supply Voltage (LMV794)

20216381

VOS vs. VCM

20216309

VOS vs. VCM

20216351

VOS vs. VCM

20216311

VOS vs. Supply Voltage

20216312

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LMV793/LMV794

Slew Rate vs. Supply Voltage

20216352

Input Bias Current vs. VCM

20216362

Input Bias Current vs. VCM

20216387

Sourcing Current vs. Supply Voltage

20216320

Sinking Current vs. Supply Voltage

20216319

Sourcing Current vs. Output Voltage

20216350

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LMV793/LMV794

Sinking Current vs. Output Voltage

20216354

Positive Output Swing vs. Supply Voltage

20216317

Negative Output Swing vs. Supply Voltage

20216315

Positive Output Swing vs. Supply Voltage

20216316

Negative Output Swing vs. Supply Voltage

20216314

Positive Output Swing vs. Supply Voltage

20216318

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LMV793/LMV794

Negative Output Swing vs. Supply Voltage

20216313

Time Domain Voltage Noise

20216338

Input Referred Voltage Noise vs. Frequency

20216339

Overshoot and Undershoot vs. CLOAD

20216330

THD+N vs. Frequency

20216326

THD+N vs. Frequency

20216304

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LMV793/LMV794

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20216374

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20216375

Open Loop Gain and Phase

20216306

Closed Loop Output Impedance vs. Frequency

20216332

Small Signal Transient Response, AV = +10

20216353

Large Signal Transient Response, AV = +10

20216355

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LMV793/LMV794

Small Signal Transient Response, AV = +10

20216357

Large Signal Transient Response, AV = +10

20216363

PSRR vs. Frequency

20216370

CMRR vs. Frequency

20216356

Input Common Mode Capacitance vs. VCM

20216376

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LMV793/LMV794

Application Information

ADVANTAGES OF THE LMV793/LMV794

Wide Bandwidth at Low Supply Current

The LMV793/LMV794 are high performance op amps that

provide a GBW of 88 MHz with a gain of 10 while drawing a

low supply current of 1.15 mA. This makes them ideal for pro-

viding wideband amplification in data acquisition applications.

With the proper external compensation the LMV793/LMV794

can be operated at gains of ±1 and still maintain much faster

slew rates than comparable unity gain stable amplifiers. The

increase in bandwidth and slew rate is obtained without any

additional power consumption over the LMV796.

Low Input Referred Noise and Low Input Bias Current

The LMV793/LMV794 have a very low input referred voltage

noise density (5.8 nV/ at 1 kHz). A CMOS input stage en-

sures a small input bias current (100 fA) and low input referred

current noise (0.01 pA/ ). This is very helpful in maintain-

ing signal integrity, and makes the LMV793/LMV794 ideal for

audio and sensor based applications.

Low Supply Voltage

The LMV793 and LMV794 have performance guaranteed at

2.5V and 5V supply. These parts are guaranteed to be oper-

ational at all supply voltages between 2.0V and 5.5V, for

ambient temperatures ranging from −40°C to 125°C, thus uti-

lizing the entire battery lifetime. The LMV793/LMV794 are

also guaranteed to be operational at 1.8V supply voltage, for

temperatures between 0°C and 125°C optimizing their usage

in low-voltage applications.

RRO and Ground Sensing

Rail-to-rail output swing provides the maximum possible dy-

namic range. This is particularly important when operating at

low supply voltages. An innovative positive feedback scheme

is used to boost the current drive capability of the output

stage. This allows the LMV793/LMV794 to source more than

40 mA of current at 1.8V supply. This also limits the perfor-

mance of these parts as comparators, and hence the usage

of the LMV793 and the LMV794 in an open-loop configuration

is not recommended. The input common-mode range in-

cludes the negative supply rail which allows direct sensing at

ground in single supply operation.

Small Size

The small footprint of the LMV793 and the LMV794 package

saves space on printed circuit boards, and enables the design

of smaller electronic products, such as cellular phones,

pagers, or other portable systems. Long traces between the

signal source and the op amp make the signal path more

susceptible to noise pick up.

The physically smaller LMV793/LMV794 packages, allow the

op amp to be placed closer to the signal source, thus reducing

noise pickup and maintaining signal integrity.

USING THE DECOMPENSATED LMV793

Advantages of Decompensated Op Amps

A unity gain stable op amp, which is fully compensated, is

designed to operate with good stability down to gains of ±1.

The large amount of compensation does provide an op amp

that is relatively easy to use; however, a decompensated op

amp is designed to maximize the bandwidth and slew rate

without any additional power consumption. This can be very

advantageous.

The LMV793/LMV794 require a gain of ±10 to be stable.

However, with an external compensation network (a simple

RC network) these parts can be stable with gains of ±1 and

still maintain the higher slew rate. Looking at the Bode plots

for the LMV793 and its closest equivalent unity gain stable op

amp, the LMV796, one can clearly see the increased band-

width of the LMV793. Both plots are taken with a parallel

combination of 20 pF and 10 kΩ for the output load.

20216322

FIGURE 1. LMV793 AVOL vs. Frequency

20216323

FIGURE 2. LMV796 AVOL vs. Frequency

Figure 1 shows the much larger 88 MHz bandwidth of the

LMV793 as compared to the 17 MHz bandwidth of the

LMV796 shown in Figure 2. The decompensated LMV793

has five times the bandwidth of the LMV796.

What is a Decompensated Op Amp?

The differences between the unity gain stable op amp and the

decompensated op amp are shown in Figure 3. This Bode plot

assumes an ideal two pole system. The dominant pole of the

decompensated op amp is at a higher frequency, f1, as com-

pared to the unity-gain stable op amp which is at fd as shown

in Figure 3. This is done in order to increase the speed capa-

bility of the op amp while maintaining the same power dissi-

pation of the unity gain stable op amp. The LMV793/LMV794

have a dominant pole at 1.6 kHz. The unity gain stable

LMV796/LMV797 have their dominant pole at 300 Hz.

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LMV793/LMV794

20216324

FIGURE 3. Open Loop Gain for Unity-Gain Stable Op Amp

and Decompensated Op Amp

Having a higher frequency for the dominate pole will result in:

1. The DC open-loop gain (AVOL) extending to a higher

frequency.

2. A wider closed loop bandwidth.

3. Better slew rate due to reduced compensation

capacitance within the op amp.

The second open loop pole (f2) for the LMV793/LMV794 oc-

curs at 45 MHz. The unity gain (fu’) occurs after the second

pole at 51 MHz. An ideal two pole system would give a phase

margin of 45° at the location of the second pole. The LMV793/

LMV794 have parasitic poles close to the second pole, giving

a phase margin closer to 0°. Therefore it is necessary to op-

erate the LMV793/LMV794 at a closed loop gain of 10 or

higher, or to add external compensation in order to assure

stability.

For the LMV796, the gain bandwidth product occurs at 17

MHz. The curve is constant from fd to fu which occurs before

the second pole.

For the LMV793/LMV794, the GBW = 88 MHz and is constant

between f1 and f2. The second pole at f2 occurs before AVOL

= 1. Therefore fu’ occurs at 51 MHz, well before the GBW

frequency of 88 MHz. For decompensated op amps the unity

gain frequency and the GBW are no longer equal. Gmin is the

minimum gain for stability and for the LMV793/LMV794 this

is a gain of 18 to 20 dB.

Input Lead-Lag Compensation

The recommended technique which allows the user to com-

pensate the LMV793/LMV794 for stable operation at any gain

is lead-lag compensation. The compensation components

added to the circuit allow the user to shape the feedback

function to make sure there is sufficient phase margin when

the loop gain is as low as 0 dB and still maintain the advan-

tages over the unity gain op amp. Figure 4 shows the lead-

lag configuration. Only RC and C are added for the necessary

compensation.

20216325

FIGURE 4. LMV793 with Lead-Lag Compensation for

Inverting Configuration

To cover how to calculate the compensation network values

it is necessary to introduce the term called the feedback factor

or F. The feedback factor F is the feedback voltage VA-VB

across the op amp input terminals relative to the op amp out-

put voltage VOUT.

From feedback theory the classic form of the feedback equa-

tion for op amps is:

A is the open loop gain of the amplifier and AF is the loop gain.

Both are highly important in analyzing op amps. Normally AF

>>1 and so the above equation reduces to:

Deriving the equations for the lead-lag compensation is be-

yond the scope of this datasheet. The derivation is based on

the feedback equations that have just been covered. The in-

verse of feedback factor for the circuit in Figure 4 is:

(1)

where 1/F's pole is located at

(2)

1/F's zero is located at

(3)

(4)

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LMV793/LMV794

The circuit gain for Figure 4 at low frequencies is −RF/RIN, but

F, the feedback factor is not equal to the circuit gain. The

feedback factor is derived from feedback theory and is the

same for both inverting and non-inverting configurations. Yes,

the feedback factor at low frequencies is equal to the gain for

the non-inverting configuration.

(5)

From this formula, we can see that

•1/F's zero is located at a lower frequency compared with

1/F's pole.

•1/F's value at low frequency is 1 + RF/RIN.

•This method creates one additional pole and one

additional zero.

•This pole-zero pair will serve two purposes:

—To raise the 1/F value at higher frequencies prior to its

intercept with A, the open loop gain curve, in order to

meet the Gmin = 10 requirement. For the LMV793/

LMV794 some overcompensation will be necessary for

good stability.

—To achieve the previous purpose above with no

additional loop phase delay.

Please note the constraint 1/F ≥ Gmin needs to be satisfied

only in the vicinity where the open loop gain A and 1/F inter-

sect; 1/F can be shaped elsewhere as needed. The 1/F pole

must occur before the intersection with the open loop gain A.

In order to have adequate phase margin, it is desirable to fol-

low these two rules:

Rule 1 1/F and the open loop gain A should intersect at the

frequency where there is a minimum of 45° of phase

margin. When over-compensation is required the in-

tersection point of A and 1/F is set at a frequency

where the phase margin is above 45°, therefore in-

creasing the stability of the circuit.

Rule 2 1/F’s pole should be set at least one decade below

the intersection with the open loop gain A in order to

take advantage of the full 90° of phase lead brought

by 1/F’s pole which is F’s zero. This ensures that the

effect of the zero is fully neutralized when the 1/F and

A plots intersect each other.

Calculating Lead-Lag Compensation for LMV793/

LMV794

Figure 5 is the same plot as Figure 1, but the AVOL and phase

curves have been redrawn as smooth lines to more readily

show the concepts covered, and to clearly show the key pa-

rameters used in the calculations for lead-lag compensation.

20216348

FIGURE 5. LMV793/LMV794 Simplified Bode Plot

To obtain stable operation with gains under 10 V/V the open

loop gain margin must be reduced at high frequencies to

where there is a 45° phase margin when the gain margin of

the circuit with the external compensation is 0 dB. The pole

and zero in F, the feedback factor, control the gain margin at

the higher frequencies. The distance between F and AVOL is

the gain margin; therefore, the unity gain point (0 dB) is where

F crosses the AVOL curve.

For the example being used RIN = RF for a gain of −1. There-

fore F = 6 dB at low frequencies. At the higher frequencies

the minimum value for F is 18 dB for 45° phase margin. From

Equation 5 we have the following relationship:

Now set RF = RIN = R. With these values and solving for RC

we have RC = R/5.9. Note that the value of C does not affect

the ratio between the resistors. Once the value of the resistors

are set, then the position of the pole in F must be set. A

2 kΩ resistor is used for RF and RIN in this design. Therefore

the value for RC is set at 330Ω, the closest standard value for

2 kΩ/5.9.

Rewriting Equation 2 to solve for the minimum capacitor value

gives the following equation:

C = 1/(2πfpRC)

The feedback factor curve, F, intersects the AVOL curve at

about 12 MHz. Therefore the pole of F should not be any

larger than 1.2 MHz. Using this value and Rc = 330Ω the min-

imum value for C is 390 pF. Figure 6 shows that there is too

much overshoot, but the part is stable. Increasing C to 2.2 nF

did not improve the ringing, as shown in Figure 7.

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LMV793/LMV794

20216303

FIGURE 6. First Try at Compensation, Gain = −1

20216307

FIGURE 7. C Increased to 2.2 nF, Gain = −1

Some over-compensation appears to be needed for the de-

sired overshoot characteristics. Instead of intersecting the

AVOL curve at 18 dB, 2 dB of over-compensation will be used,

and the AVOL curve will be intersected at 20 dB. Using Equa-

tion 5 for 20 dB, or 10 V/V, the closest standard value of RC

is 240Ω. The following two waveforms show the new resistor

value with C = 390 pF and 2.2 nF. Figure 9 shows the final

compensation and a very good response for the 1 MHz

square wave.

20216308

FIGURE 8. RC = 240Ω and C = 390 pF, Gain = −1

20216310

FIGURE 9. RC = 240Ω and C = 2.2 nF, Gain = −1

To summarize, the following steps were taken to compensate

the LMV793 for a gain of −1:

1. Values for Rc and C were calculated from the Bode plot

to give an expected phase margin of 45°. The values

were based on RIN = RF = 2 kΩ. These calculations gave

Rc = 330Ω and C = 390 pF.

2. To reduce the ringing C was increased to 2.2 nF which

moved the pole of F, the feedback factor, farther away

from the AVOL curve.

3. There was still too much ringing so 2 dB of over-

compensation was added to F. This was done by

decreasing RC to 240Ω.

The LMV796 is the fully compensated part which is compa-

rable to the LMV793. Using the LMV796 in the same setup,

but removing the compensation network, provide the re-

sponse shown in Figure 10 .

20216321

FIGURE 10. LMV796 Response

For large signal response the rise and fall times are dominat-

ed by the slew rate of the op amps. Even though both parts

are quite similar the LMV793 will give rise and fall times about

2.5 times faster than the LMV796. This is possible because

the LMV793 is a decompensated op amp and even though it

is being used at a gain of −1, the speed is preserved by using

a good technique for external compensation.

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LMV793/LMV794

Non-Inverting Compensation

For the non-inverting amp the same theory applies for estab-

lishing the needed compensation. When setting the inverting

configuration for a gain of −1, F has a value of 2. For the non-

inverting configuration both F and the actual gain are the

same, making the non-inverting configuration more difficult to

compensate. Using the same circuit as shown in Figure 4, but

setting up the circuit for non-inverting operation (gain of +2)

results in similar performance as the inverting configuration

with the inputs set to half the amplitude to compensate for the

additional gain. Figure 11 below shows the results.

20216382

FIGURE 11. RC = 240Ω and C = 2.2 nF, Gain = +2

20216383

FIGURE 12. LMV796 Response Gain = +2

The response shown in Figure 11 is close to the response

shown in Figure 9. The part is actually slightly faster in the

non-inverting configuration. Decreasing the value of RC to

around 200Ω can decrease the negative overshoot but will

have slightly longer rise and fall times. The other option is to

add a small resistor in series with the input signal. Figure 12

shows the performance of the LMV796 with no compensation.

Again the decompensated parts are almost 2.5 times faster

than the fully compensated op amp.

The most difficult op amp configuration to stabilize is the gain

of +1. With proper compensation the LMV793/LMV794 can

be used in this configuration and still maintain higher speeds

than the fully compensated parts. Figure 13 shows the gain =

1, or the buffer configuration, for these parts.

20216384

FIGURE 13. LMV793 with Lead-Lag Compensation for

Non-Inverting Configuration

Figure 13 is the result of using Equation 5 and additional ex-

perimentation in the lab. RP is not part of Equation 5, but it is

necessary to introduce another pole at the input stage for

good performance at gain = +1. Equation 5 is shown below

with RIN = ∞.

Using 2 kΩ for RF and solving for RC gives RC = 2000/6.9 =

290Ω. The closest standard value for RC is 300Ω. After some

fine tuning in the lab RC = 330Ω and RP = 1.5 kΩ were

choosen as the optimum values. RP together with the input

capacitance at the non-inverting pin inserts another pole into

the compensation for the LMV793/LMV794. Adding this pole

and slightly reducing the compensation for 1/F (using a slight-

ly higher resistor value for RC) gives the optimum response

for a gain of +1. Figure 14 is the response of the circuit shown

in Figure 13. Figure 15 shows the response of the LMV796 in

the buffer configuration with no compensation and RP = RF =

20216388

FIGURE 14. RC = 330Ω and C = 10 nF, Gain = +1

www.national.com 16

LMV793/LMV794

20216389

FIGURE 15. LMV796 Response Gain = +1

With no increase in power consumption the decompensated

op amp offers faster speed over the compensated equivalent

part. These examples used RF = 2 kΩ. This value is high

enough to be easily driven by the LMV793/LMV794, yet small

enough to minimize the effects from the parasitic capacitance

of both the PCB and the op amp.

Note: When using the LMV793/LMV794, proper high fre-

quency PCB layout must be followed. The GBW of these parts

is 88 MHz, making the PCB layout significantly more critical

than when using the compensated counterparts which have

a GBW of 17 MHz.

TRANSIMPEDANCE AMPLIFIER

An excellent application for either the LMV793 or the LMV794

is as a transimpedance amplifier. With a GBW product of 88

MHz these parts are ideal for high speed data transmission

by light. The circuit shown on the front page of the datasheet

is the circuit used to test the LMV793/LMV794 as tran-

simpedance amplifiers. The only change is that VB is tied to

the VCC of the part, thus the direction of the diode is reversed

from the circuit shown on the front page.

Very high speed components were used in testing to check

the limits of the LMV793/LMV794 in a transimpedance con-

figuration. The photo diode part number is PIN-HR040 from

OSI Optoelectronics. The diode capacitance for this part is

only about 7 pF for the 2.5V bias used (VCC to virtual ground).

The rise time for this diode is 1 nsec. A laser diode was used

for the light source. Laser diodes have on and off times under

5 nsec. The speed of the selected optical components al-

lowed an accurate evaluation of the LMV793 as a tran-

simpedance amplifier. Nationals Evaluation Board for decom-

pensated op amps, PN 551013271-001 A, was used and only

minor modifications were necessary and no traces had to be

cut.

20216361

FIGURE 16. Transimpedance Amplifier

Figure 16 is the complete schematic for a transimpedance

amplifier. Only the supply bypass capacitors are not shown.

CD represents the photo diode capacitance which is given on

its datasheet. CCM is the input common mode capacitance of

the op amp and, for the LMV793 it is shown in the last drawing

of the Typical Performance Characteristics section of this

datasheet. In Figure 16 the inverting input pin of the LMV793

is kept at virtual ground. Even though the diode is connected

to the 2.5V line, a power supply line is AC ground, thus CD is

connected to ground.

Figure 17 shows the schematic needed to derive F, the feed-

back factor, for a transimpedance amplifier. In this figure CD

+ CCM = CIN. Therefore it is critical that the designer knows

the diode capacitance and the op amp input capacitance. The

photo diode is close to an ideal current source once its ca-

pacitance is included in the model. What kind of circuit is this?

Without CF there is only an input capacitor and a feedback

resistor. This circuit is a differentiator! Remember, differen-

tiator circuits are inherently unstable and must be compen-

sated. In this case CF compensates the circuit.

20216364

FIGURE 17. Transimpedance Feedback Model

17 www.national.com

LMV793/LMV794

Using feedback theory, F = VA/VOUT, this becomes a voltage

divider giving the following equation:

The noise gain is 1/F. Because this is a differentiator circuit,

a zero must be inserted. The location of the zero is given by:

CF has been added for stability. The addition of this part adds

a pole to the circuit. The pole is located at:

To attain maximum bandwidth and still have good stability the

pole is to be located on the open loop gain curve which is A.

If additional compensation is required one can always in-

crease the value of CF, but this will also reduce the bandwidth

of the circuit. Therefore A = 1/F, or AF = 1. For A the equation

is:

The expression fGBW is the gain bandwidth product of the part.

For a unity gain stable part this is the frequency where A = 1.

For the LMV793 fGBW = 88 MHz. Multiplying A and F results

in the following equation:

For the above equation s = jω. To find the actual amplitude of

the equation the square root of the square of the real and

imaginary parts are calculated. At the intersection of F and A,

we have:

After a bit of algebraic manipulation the above equation re-

duces to:

In the above equation the only unknown is CF. In trying to

solve this equation the fourth power of CF must be dealt with.

An excel spread sheet with this equation can be used and all

the known values entered. Then through iteration, the value

of CF when both sides are equal will be found. That is the

correct value for CF, and of course the closest standard value

is used for CF.

Before moving the lab, the transfer function of the tran-

simpedance amplifier must be found and the units must be in

Ohms.

The LMV793 was evaluated for RF = 10 kΩ and 100 kΩ, rep-

resenting a somewhat lower gain configuration and with the

100 kΩ feedback resistor a fairly high gain configuration. The

RF = 10 kΩ is covered first. Looking at the Input Common

Mode Capacitance vs. VCM chart for CCM for the operating

point selected CCM = 15 pF. Note that for split supplies VCM =

2.5V, CIN = 22 pF and fGBW = 88 MHz. Solving for CF the cal-

culated value is 1.75 pF, so 1.8 pF is selected for use.

Checking the frequency of the pole finds that it is at 8.8 MHz,

which is right at the minimum gain recommended for this part.

Some over compensation was necessary for stability and the

final selected value for CF is 2.7 pF. This moves the pole to

5.9 MHz. Figure 18 and Figure 19 show the rise and fall times

obtained in the lab with a 1V output swing. The laser diode

was difficult to drive due to thermal effects making the starting

and ending point of the pulse quite different, therefore the two

separate scope pictures.

20216390

FIGURE 18. Fall Time

www.national.com 18

LMV793/LMV794

20216391

FIGURE 19. Rise Time

In Figure 18 the ringing and the hump during the on time is

from the laser. The higher drive levels for the laser gave ring-

ing in the light source as well as light changing from the

thermal characteristics. The hump is due to the thermal char-

acteristics.

Solving for CF using a 100 kΩ feedback resistor, the calcu-

lated value is 0.54 pF. One of the problems with more gain is

the very small value for CF. A 0.5 pF capacitor was used, its

measured value being 0.64 pF. For the 0.64 pF location the

pole is at 2.5 MHz. Figure 20 shows the response for a 1V

output.

20216392

FIGURE 20. High Gain Response

A transimpedance amplifier is an excellent application for the

LMV793. Even with the high gain using a 100 kΩ feedback

resistor, the bandwidth is still well over 1 MHz. Other than a

little over compensation for the 10 kΩ feedback resistor con-

figuration using the LMV793 was quite easy. Of course a very

good board layout was also used for this test. For information

on photo diodes please contact OSI Optoelectronics, (310)

978-0516. For further information on transimpedance ampli-

fiers please contact your National Semiconductor represen-

tative.

19 www.national.com

LMV793/LMV794

Physical Dimensions inches (millimeters) unless otherwise noted

5-Pin SOT23

NS Package Number MF05A

8-Pin SOIC

NS Package Number M08A

www.national.com 20

LMV793/LMV794

8-Pin MSOP

NS Package Number MUA08A

21 www.national.com

LMV793/LMV794

Notes

LMV793/LMV794 88 MHz, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifiers

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