LTC3129-1
1
31291fc
For more information www.linear.com/LTC3129-1
Typical applicaTion
FeaTures DescripTion
15V, 200mA Synchronous
Buck-Boost DC/DC Converter
with 1.3µA Quiescent Current
The LT C
®
3129-1 is a high efficiency, 200mA buck-boost
DC/DC converter with a wide VIN and VOUT range. It
includes an accurate RUN pin threshold to allow predict-
able regulator turn-on and a maximum power point control
(MPPC) capability that ensures maximum power extraction
from non-ideal power sources such as photovoltaic panels.
The LTC3129-1 employs an ultralow noise, 1.2MHz PWM
switching architecture that minimizes solution footprint by
allowing the use of tiny, low profile inductors and ceramic
capacitors. Built-in loop compensation and soft-start
simplify the design. For high efficiency operation at light
loads, automatic Burst Mode operation can be selected,
reducing the quiescent current to just 1.3µA. To further
reduce part count and improve light load efficiency, the
LTC3129-1 includes an internal voltage divider to provide
eight selectable fixed output voltages.
Additional features include a power good output, less than
10nA of shutdown current and thermal shutdown.
The LTC3129-1 is available in thermally enhanced
3mm×3mm QFN and 16-lead MSOP packages. For an
adjustable output voltage, see the functionally equivalent
LTC3129.
L, LT , LT C , LT M , Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective owners.
applicaTions
n Regulates VOUT Above, Below or Equal to VIN
n Wide VIN Range: 2.42V to 15V, 1.92V to 15V After
Start-Up (Bootstrapped)
n Fixed Output Voltage with Eight User-Selectable
Settings from 2.5V to 15V
n 200mA Output Current in Buck Mode
n Single Inductor
n 1.3µA Quiescent Current
n Programmable Maximum Power Point Control
n 1.2MHz Ultralow Noise PWM
n Current Mode Control
n Pin Selectable Burst Mode
®
Operation
n Up to 95% Efficiency
n Accurate RUN Pin Threshold
n Power Good Indicator
n 10nA Shutdown Current
n Thermally Enhanced 3mm × 3mm QFN and
16-Lead MSOP Packages
n Industrial Wireless Sensor Nodes
n Post-Regulator for Harvested Energy
n Solar Panel Post-Regulator/Charger
n Intrinsically Safe Power Supplies
n Wireless Microphones
n Avionics-Grade Wireless Headsets
BST1
VOUT
VOUT
5V AT
100mA VIN < VOUT
200mA VIN > VOUT
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD
GND
VCC
VCC
VIN
VIN
2.42V TO 15V
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 10µH
10µF
10µF
2.2µF
31291 TA01a
PGND
AA OR AAA
BATTERIES
Efficiency and Power Loss vs Load
100
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
POWER LOSS (mW)
70
80
90
50
30
20
60
40
10
0
1000
100
0.1
10
1
0.01
3129 TA01b
0.1 100 1000101
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 15V
EFFICIENCY
POWER LOSS
VOUT = 5V
LTC3129-1
2
31291fc
For more information www.linear.com/LTC3129-1
absoluTe MaxiMuM raTings
VIN, VOUT Voltages ..................................... –0.3V to 18V
SW1 DC Voltage .............................. –0.3V to (VIN + 0.3V)
SW2 DC Voltage............................–0.3V to (VOUT + 0.3V)
SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V
BST1 Voltage .....................(SW1 – 0.3V) to (SW1 + 6V)
BST2 Voltage .....................(SW2 – 0.3V) to (SW2 + 6V)
RUN, PGOOD Voltage ................................. –0.3V to 18V
VCC, PWM, MPPC, VS1, VS2,
VS3 Voltages ............................................... –0.3V to 6V
PGOOD Sink Current ..............................................15mA
Operating Junction Temperature Range
(Notes 2, 5) ............................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
MSE Lead Temperature (Soldering, 10 sec) .......... 300°C
(Notes 1, 8)
16 15 14 13
5678
TOP VIEW
UD PACKAGE
16-LEAD (3mm × 3mm) PLASTIC QFN
9
10
11
12
4
3
2
1
BST1
VIN
VCC
RUN
VOUT
PGOOD
PWM
VS1
SW1
PGND
SW2
BST2
MPPC
GND
VS3
VS2
17
PGND
TJMAX = 125°C, θJC = 7.5°C/W, θJA = 68°C/W (NOTE 6)
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
1
2
3
4
5
6
7
8
VCC
RUN
MPPC
GND
VS3
VS2
VS1
PWM
16
15
14
13
12
11
10
9
VIN
BST1
SW1
PGND
SW2
BST2
VOUT
PGOOD
TOP VIEW
MSE PACKAGE
16-LEAD PLASTIC MSOP
17
PGND
TJMAX = 125°C, θJC = 10°C/W, θJA = 40°C/W (NOTE 6)
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
pin conFiguraTion
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3129EUD-1#PBF LTC3129EUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3129IUD-1#PBF LTC3129IUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3129EMSE-1#PBF LTC3129EMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C
LTC3129IMSE-1#PBF LTC3129IMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C
Consult LT C Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
LTC3129-1
3
31291fc
For more information www.linear.com/LTC3129-1
elecTrical characTerisTics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
PARAMETER CONDITIONS MIN TYP MAX UNITS
VIN Start-Up Voltage l2.25 2.42 V
Input Voltage Range VCC > 2.42V (Back-Driven) l1.92 15 V
VIN UVLO Threshold (Rising) VCC > 2.42V (Back-Driven) l1.8 1.9 2.0 V
VIN UVLO Hysteresis l80 100 130 mV
VOUT Voltages VS1 = VS2 = VS3 = 0V
VS1 = VCC, VS2 = VS3 = 0V
VS2 = VCC, VS1 = VS3 = 0V
VS1 = VS2 = VCC, VS3 = 0V
VS1 = VS2 = 0V, VS3 = VCC
VS2 = 0V, VS1 = VS3 = VCC
VS1 = 0V, VS2 = VS3 = VCC
VS1 = VS2 = VS3 = VCC
l
l
l
l
l
l
l
l
2.425
3.2175
3.998
4.875
6.727
7.995
11.64
14.50
2.5
3.3
4.1
5.0
6.9
8.2
12
15.0
2.575
3.383
4.203
5.125
7.073
8.405
12.40
15.50
V
V
V
V
V
V
V
V
Quiescent Current (VIN) – Shutdown RUN = 0V, Including Switch Leakage 10 100 nA
Quiescent Current (VIN) UVLO Either VIN or VCC Below Their UVLO Threshold, or
RUN Below the Threshold to Enable Switching
1.9 3 µA
Quiescent Current – Burst Mode Operation Measured on VIN, VOUT > VREG
PWM = 0V, RUN = VIN
1.3 2.0 µA
N-Channel Switch Leakage on VIN and VOUT SW1 = 0V, VIN = 15V
SW2 = 0V, VOUT = 15V
RUN = 0V
10 50 nA
N-Channel Switch On-Resistance VCC = 4V 0.75 Ω
Inductor Average Current Limit VOUT > UV Threshold (Note 4)
VOUT < UV Threshold (Note 4)
l
l
220
80
275
130
350
200
mA
mA
Inductor Peak Current Limit (Note 4) l400 500 680 mA
Maximum Boost Duty Cycle VOUT < VREG as Set by VS1-VS3. Percentage of
Period SW2 is Low in Boost Mode (Note 7)
l85 89 95 %
Minimum Duty Cycle VOUT > VREG as Set by VS1-VS3. Percentage of
Period SW1 is High in Buck Mode (Note 7)
l0 %
Switching Frequency PWM = VCC l1.0 1.2 1.4 MHz
SW1 and SW2 Minimum Low Time (Note 3) 90 ns
MPPC Voltage l1.12 1.175 1.22 V
MPPC Input Current MPPC = 5V 1 10 nA
RUN Threshold to Enable VCC l0.5 0.9 1.15 V
RUN Threshold to Enable Switching (Rising) VCC > 2.4V l1.16 1.22 1.28 V
RUN (Switching) Threshold Hysteresis 50 80 120 mV
RUN Input Current RUN = 15V 1 10 nA
VS1, VS2, VS3 Input High l1.2 V
VS1, VS2, VS3 Input Low l0.4 V
VS1, VS2, VS3 Input Current VS1, VS2, VS3 = VCC = 5V 1 10 nA
PWM Input High l1.6 V
PWM Input Low l0.5 V
PWM Input Current PWM = 5V 0.1 1 µA
Soft-Start Time 3 ms
VCC Voltage VIN > 4.85V l3.4 4.1 4.7 V
VCC Dropout Voltage (VIN – VCC) VIN = 3.0V, Switching
VIN = 2.0V (VCC in UVLO)
35
0
60
2
mV
mV
LTC3129-1
4
31291fc
For more information www.linear.com/LTC3129-1
elecTrical characTerisTics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3129-1 is tested under pulsed load conditions such
that TJ ≈ TA. The LTC3129E-1 is guaranteed to meet specifications
from 0°C to 85°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3129I-1 is guaranteed over the full –40°C to 125°C operating junction
temperature range. The junction temperature (TJ, in °C) is calculated from
the ambient temperature (TA, in °C) and power dissipation (PD, in watts)
according to the formula:
TJ = TA + (PD • θJA),
where θJA (in °C/W) is the package thermal impedance.
Note that the maximum ambient temperature consistent with these
specifications is determined by specific operating conditions in
conjunction with board layout, the rated thermal package thermal
resistance and other environmental factors.
PARAMETER CONDITIONS MIN TYP MAX UNITS
VCC UVLO Threshold (Rising) l2.1 2.25 2.42 V
VCC UVLO Hysteresis 60 mV
VCC Current Limit VCC = 0V l4 20 60 mA
VCC Back-Drive Voltage (Maximum) l5.5 V
VCC Input Current (Back-Driven) VCC = 5.5V (Switching) 2 4 mA
VCC Leakage to VIN if VCC>VIN VCC = 5.5V, VIN = 1.8V, Measured on VIN –27 µA
VOUT UV Threshold (Rising) l0.95 1.15 1.35 V
VOUT UV Hysteresis 150 mV
VOUT Current – Shutdown RUN = 0V, VOUT = 15V Including Switch Leakage 10 100 nA
VOUT Current – Sleep PWM = 0V, VOUT ≥ VREG VOUT/27 µA
VOUT Current – Active PWM = VCC, VOUT = 15V (Note 4) 5 9 µA
PGOOD Threshold, Falling Referenced to Programmed VOUT Voltage –5.5 –7.5 –10 %
PGOOD Hysteresis Referenced to Programmed VOUT Voltage 2.5 %
PGOOD Voltage Low ISINK = 1mA 250 300 mV
PGOOD Leakage PGOOD = 15V 1 50 nA
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
Note 3: Specification is guaranteed by design and not 100% tested in
production.
Note 4: Current measurements are made when the output is not switching.
Note 5: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
Note 6: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a much higher thermal resistance.
Note 7: Switch timing measurements are made in an open-loop test
configuration. Timing in the application may vary somewhat from these
values due to differences in the switch pin voltage during non-overlap
durations when switch pin voltage is influenced by the magnitude and
duration of the inductor current.
Note 8: Voltage transients on the switch pin(s) beyond the DC limits
specified in the Absolute Maximum Ratings are non-disruptive to normal
operation when using good layout practices as described elsewhere in the
data sheet and Application Notes and as seen on the product demo board.
LTC3129-1
5
31291fc
For more information www.linear.com/LTC3129-1
Typical perForMance characTerisTics
Efficiency, VOUT = 2.5V
TA = 25°C, unless otherwise noted.
Power Loss, VOUT = 2.5V Efficiency, VOUT = 3.3V
Power Loss, VOUT = 3.3V
Efficiency, VOUT = 5V Power Loss, VOUT = 5V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
31291 G01
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
BURST
PWM
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
31291 G02
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
BURST
PWM
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
31291 G03
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
BURST
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
31291 G04
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
0.1
0.01
1
100.1
31291 G04b
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
BURST
Efficiency, VOUT = 4.1V Power Loss, VOUT = 4.1V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
31291 G05
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
31291 G06
10001 100
PWM
BURST VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
Efficiency, VOUT = 6.9V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
80
70
60
30
20
10
0
50
40
100.1
31291 G04a
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
80
70
60
30
20
10
0
50
40
100.1
31291 G06a
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
LTC3129-1
6
31291fc
For more information www.linear.com/LTC3129-1
Typical perForMance characTerisTics
TA = 25°C, unless otherwise noted.
Maximum Output Current
vs VIN and VOUT
No Load Input Current
vs VIN and VOUT (PWM = 0V)Power Loss, VOUT = 15V
VIN (V)
2
IOUT (mA)
250
200
150
100
50
0133 4
31291 G11
1510 1411 1285 96 7
VOUT = 2.5V
VOUT = 3.3V
VOUT = 4.1V
VOUT = 5V
VOUT = 6.9V
VOUT = 8.2V
VOUT = 12V
VOUT = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
0.1
0.01
1
100.1
31291 G06b
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
BURST
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
80
70
60
30
20
10
0
50
40
100.1
31291 G06c
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
0.1
0.01
1
100.1
31291 G06d
10001 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
BURST
Efficiency, VOUT = 12V Power Loss, VOUT = 12V Efficiency, VOUT = 15V
Power Loss, VOUT = 6.9V Efficiency, VOUT = 8.2V Power Loss, VOUT = 8.2V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
31291 G07
10001 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
31291 G09
10001 100
PWM
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
31291 G08
10001 100
PWM
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
31291 G10
10001 100
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
VIN (V)
2
IIN (µA)
5
4
3
2
1
012 14
31291 G12
16104 86
VOUT = 2.5V
VOUT = 3.3V
VOUT = 4.1V
VOUT = 5V
VOUT = 6.9V
VOUT = 8.2V
VOUT = 12V
VOUT = 15V
LTC3129-1
7
31291fc
For more information www.linear.com/LTC3129-1
Accurate RUN Threshold
vs Temperature (Normalized to 25°C)
Maximum Output vs Temperature
(Normalized to 25°C)
VCC Dropout Voltage vs Temperature
(PWM Mode, Switching)
VCC Dropout Voltage vs VIN
(PWM Mode, Switching)
Fixed Frequency PWM
Waveforms
Typical perForMance characTerisTics
TA = 25°C, unless otherwise noted.
Burst Mode Threshold
vs VIN and VOUT Switch RDS(ON) vs Temperature
Output Voltage vs Temperature
(Normalized to 25°C)
TEMPERATURE (°C)
–45
R
DS(ON)
(Ω)
1.3
1.2
1.1
1.0
0.8
0.7
0.6
0.5
0.9
0.4 –20
31291 G14
130
55 10580305
VCC = 2.5V
VCC = 3V
VCC = 4V
VCC = 5V
TEMPERATURE (°C)
–45
CHANGE IN V
OUT
(%)
1.0
0.5
0
–0.5
–1.0 –20
31291 G15
130
55 10580305
Average Input Current Limit
vs MPPC Voltage
TEMPERATURE (°C)
–45
DROPOUT (mV)
60
50
30
20
40
0
10
–20
31291 G20
130
55 10580305
VIN (V)
2
DROPOUT (mV)
60
50
30
20
40
0
10
2.25
31291 G21
4
3 3.5 3.753.252.752.5
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 200mA
SW2
5V/DIV
SW1
5V/DIV
31291 G22
500ns/DIV
IL
200mA/DIV
TEMPERATURE (°C)
–45
CHANGE IN RUN THRESHOLD (%)
2
0
–1
1
–2 –20
31291 G17
13055 10580305
MPPC PIN VOLTAGE (V)
1.13
PERCENTAGE OF FULL INPUT CURRENT (%)
100
90
70
60
50
40
30
20
10
80
01.135
31291 G18
1.171.1651.161.1551.145 1.151.14
TEMPERATURE (°C)
–45
CHANGE IN MAXIMUM OUTPUT CURRENT (%)
15
10
0
–5
5
–15
–10
–20
31291 G19
13055 10580305
VIN (V)
2
LOAD (mA)
80
70
60
40
30
20
10
50
04
3129 G13
1610 141286
VOUT = 2.5V
VOUT = 3.3V
VOUT = 4.1V
VOUT = 5V
VOUT = 6.9V
VOUT = 8.2V
VOUT = 12V
VOUT = 15V
LTC3129-1
8
31291fc
For more information www.linear.com/LTC3129-1
Typical perForMance characTerisTics
TA = 25°C, unless otherwise noted.
Step Load Transient Response in
Burst Mode Operation
PGOOD Response to a Drop
On VOUT MPPC Response to a Step Load
L = 10µH
VIN = 7V
VOUT = 5V
COUT = 22µF
IOUT = 5mA to 125mA STEP
31291 G28
500µs/DIV
IVOUT
100mA/DIV
VOUT
100mV/DIV
VOUT = 5V
31291 G29
1ms/DIV
PGOOD
2V/DIV
VOUT
2V/DIV
VIN = 5VOC
VMPPC SET TO 3.5V
CIN = 22µF, RIN = 10Ω,
VOUT = 5V, COUT = 22µF
IOUT = 25mA to 125mA STEP
31291 G30
2ms/DIV
IVOUT
100mA/DIV
VOUT
2V/DIV
VIN
2V/DIV
Fixed Frequency Ripple on VOUT Burst Mode Waveforms Burst Mode Ripple on VOUT
Start-Up Waveforms
Step Load Transient Response in
Fixed Frequency
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 200mA
COUT = 10µF
31291 G23
200ns/DIV
IL
200mA/DIV
VOUT
20mV/DIV
VIN = 7V
VOUT = 5V
IOUT = 50mA
COUT = 22µA
31291 G26
1ms/DIV
IVIN
200mA/DIV
VOUT
5V/DIV
VCC
5V/DIV
RUN
5V/DIV
L = 10µH
VIN = 7V
VOUT = 5V
COUT = 10µF
IOUT = 50mA to 150mA STEP
31291 G27
500µs/DIV
IVOUT
100mA/DIV
VOUT
100mV/DIV
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 5mA
COUT = 22µF
31291 G24
50µs/DIV
IL
200mA/DIV
SW2
5V/DIV
SW1
5V/DIV
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 5mA
COUT = 22µF
31291 G25
100µs/DIV
IL
100mA/DIV
VOUT
100mV/DIV
LTC3129-1
9
31291fc
For more information www.linear.com/LTC3129-1
BST1 (Pin 1/Pin 15): Boot-Strapped Floating Supply for
High Side NMOS Gate Drive. Connect to SW1 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used.
VIN (Pin 2/Pin 16): Input Voltage for the Converter. Connect
a minimum of 4.7µF ceramic decoupling capacitor from
this pin to the ground plane, as close to the pin as possible.
VCC (Pin 3/Pin 1): Output Voltage of the Internal Voltage
Regulator. This is the supply pin for the internal circuitry.
Bypass this output with a minimum of 2.2µF ceramic ca-
pacitor close to the pin. This pin may be back-driven by
an external supply, up to a maximum of 5.5V.
RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull
this pin above 1.1V to enable the VCC regulator and above
1.28V to enable the converter. Connecting this pin to a
resistor divider from VIN to ground allows programming a
VIN start threshold higher than the 1.8V (typical) VIN UVLO
threshold. In this case, the typical VIN turn-on threshold is
determined by VIN = 1.22V • [1+(R3/Pin R4)] (see Figure 2).
MPPC (Pin 5/Pin 3): Maximum Power Point Control
Programming Pin. Connect this pin to a resistor divider
from VIN to ground to enable the MPPC functionality.
If the VOUT load is greater than what the power source
can provide, the MPPC will reduce the inductor current
to regulate VIN to a voltage determined by: VIN = 1.175V
• [1 + (R5/R6)] (see Figure 3). By setting the VIN regula-
tion voltage appropriately, maximum power transfer from
the limited source is assured. Note this pin is very noise
sensitive, therefore minimize trace length and stray capaci-
tance. Please refer to the Applications Information section
for more detail on programming the MPPC for different
sources. If this function is not needed, tie the pin to VCC.
GND (Pin 6/Pin 4): Signal Ground. Provide a short direct
PCB path between GND and the ground plane where the
exposed pad is soldered.
VS3 (Pin 7/Pin 5): Output Voltage Select Pin. Connect this
pin to ground or VCC to program the output voltage (see
Table 1). This pin should not float or go below ground.
If this pin is externally driven above VCC, a 1M resistor
should be added in series.
VS2 (Pin 8/Pin 6): Output Voltage Select Pin. Connect this
pin to ground or VCC to program the output voltage (see
Table 1). This pin should not float or go below ground.
VS1 (Pin 9/Pin 7): Output Voltage Select Pin. Connect this
pin to ground or VCC to program the output voltage (see
Table 1). This pin should not float or go below ground.
PWM (Pin 10/Pin 8): Mode Select Pin.
PWM = Low (ground): Enables automatic Burst Mode
operation.
PWM = High (tie to VCC): Fixed frequency PWM
operation.
This pin should not be allowed to float. It has internal 5M
pull-down resistor.
PGOOD (Pin 11/Pin 9): Open drain output that pulls to
ground when FB drops too far below its regulated voltage.
Connect a pull-up resistor from this pin to a positive sup-
ply. This pin can sink up to the absolute maximum rating
of 15mA when low. Refer to the Operation section of the
data sheet for more detail.
VOUT (Pin 12/Pin 10): Output voltage of the converter, set
by the VS1-VS3 programming pins according to Table 1.
Connect a minimum value of 4.7µF ceramic capacitor from
this pin to the ground plane, as close to the pin as possible.
BST2 (Pin 13/Pin 11): Boot-Strapped Floating Supply for
High Side NMOS Gate Drive. Connect to SW2 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used
SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
pin FuncTions
(QFN/MSOP)
LTC3129-1
10
31291fc
For more information www.linear.com/LTC3129-1
pin FuncTions
(QFN/MSOP)
PGND (Pin 15/Pin 13, Exposed Pad Pin 17/Pin 17): Power
Ground. Provide a short direct PCB path between PGND
and the ground plane. The exposed pad must also be
soldered to the PCB ground plane. It serves as a power
ground connection, and as a means of conducting heat
away from the die.
SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
Table 1. VOUT Program Settings
VS3 PIN VS2 PIN VS1 PIN VOUT
0 0 0 2.5V
0 0 VCC 3.3V
0 VCC 0 4.1V
0 VCC VCC 5V
VCC 0 0 6.9V
VCC 0 VCC 8.2V
VCC VCC 0 12V
VCC VCC VCC 15V
block DiagraM
31291 BD
LDO
VREF
START
VREF
VCC
VCC
VCC_GD
START
START
4.1V
1.175V
VREF
+
–
SD
UVLO
+
–
–
+
+
–
–
+THERMAL
SHUTDOWN
+
–
–
+
PWM
600mV
–7.5%
OSC
GND
SLEEP
100mV
RESET
ENABLE
ILIM
IZERO
ISENSE
20mA
ISENSE
VREF_GD
500mA
PGND
CLAMP
+
–
1.175V
FB
–
+
–
+
–
+
–
+
DRIVER
DRIVER
DRIVER
DRIVER
ISENSE
ISENSE
DRV_C
1.1V
UV
VS1
VOUT
VCC
VOUT
VS2
VS3DRV_D
DRV_B
DRV_A
ISENSE
LOGIC
PGOOD
SOFT-START
+
–
MPPC
RUN
VCC
VIN
VIN
BST1 SW1 SW2
D
C
A
B
BST2
PWM
SLEEP
VIN
0.9V
1.22V
1.175V
5M
1.175V
VC
VOUT
SELECT
INPUTS
LTC3129-1
11
31291fc
For more information www.linear.com/LTC3129-1
operaTion
INTRODUCTION
The LTC3129-1 is a 1.3µA quiescent current, monolithic,
current mode, buck-boost DC/DC converter that can operate
over a wide input voltage range of 1.92V to 15V and provide
up to 200mA to the load. Eight fixed, user-programmable
output voltages can be selected using the three digital
programming pins. Internal, low RDS(ON) N-channel power
switches reduce solution complexity and maximize effi-
ciency. A proprietary switch control algorithm allows the
buck-boost converter to maintain output voltage regulation
with input voltages that are above, below or equal to the
output voltage. Transitions between the step-up or step-
down operating modes are seamless and free of transients
and sub-harmonic switching, making this product ideal
for noise sensitive applications. The LTC3129-1 operates
at a fixed nominal switching frequency of 1.2MHz, which
provides an ideal trade-off between small solution size and
high efficiency. Current mode control provides inherent
input line voltage rejection, simplified compensation and
rapid response to load transients.
Burst Mode capability is also included in the LTC3129-1
and is user-selected via the PWM input pin. In Burst Mode
operation, the LTC3129-1 provides exceptional efficiency at
light output loading conditions by operating the converter
only when necessary to maintain voltage regulation. The
Burst Mode quiescent current is a miserly 1.3µA. At higher
loads, the LTC3129-1 automatically switches to fixed fre-
quency PWM mode when Burst Mode operation is selected.
(Please refer to the Typical Performance Characteristic
curves for the mode transition point at different input and
output voltages). If the application requires extremely low
noise, continuous PWM operation can also be selected
via the PWM pin.
A MPPC (maximum power point control) function is also
provided that allows the input voltage to the converter to
be servo’d to a programmable point for maximum power
when operating from various non-ideal power sources
such as photovoltaic cells. The LTC3129-1 also features
an accurate RUN comparator threshold with hysteresis,
allowing the buck-boost DC/DC converter to turn on and
off at user-selected VIN voltage thresholds. With a wide
voltage range, 1.3µA Burst Mode current and program-
mable RUN and MPPC pins, the LTC3129-1 is well suited
for many diverse applications.
PWM MODE OPERATION
If the PWM pin is high or if the load current on the con-
verter is high enough to command PWM mode operation
with PWM low, the LTC3129-1 operates in a fixed 1.2MHz
PWM mode using an internally compensated average
current mode control loop. PWM mode minimizes output
voltage ripple and yields a low noise switching frequency
spectrum. A proprietary switching algorithm provides
seamless transitions between operating modes and
eliminates discontinuities in the average inductor cur-
rent, inductor ripple current and loop transfer function
throughout all modes of operation. These advantages
result in increased efficiency, improved loop stability and
lower output voltage ripple in comparison to the traditional
buck-boost converter.
Figure 1 shows the topology of the LTC3129-1 power stage
which is comprised of four N-channel DMOS switches and
their associated gate drivers. In PWM mode operation
both switch pins transition on every cycle independent of
the input and output voltages. In response to the internal
control loop command, an internal pulse width modulator
generates the appropriate switch duty cycle to maintain
regulation of the output voltage.
A
VCC
BST1
C
BST1
C
BST2
L
BST2VIN VOUT
SW1 SW2
VCC
VCC VCC
LTC3129-1
PGND PGND
31291 F01
B
D
C
Figure 1. Power Stage Schematic
LTC3129-1
12
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For more information www.linear.com/LTC3129-1
operaTion
When stepping down from a high input voltage to a lower
output voltage, the converter operates in buck mode and
switch D remains on for the entire switching cycle except
for the minimum switch low duration (typically 90ns). Dur-
ing the switch low duration, switch C is turned on which
forces SW2 low and charges the flying capacitor, CBST2.
This ensures that the switch D gate driver power supply
rail on BST2 is maintained. The duty cycle of switches A
and B are adjusted to maintain output voltage regulation
in buck mode.
If the input voltage is lower than the output voltage, the
converter operates in boost mode. Switch A remains on
for the entire switching cycle except for the minimum
switch low duration (typically 90ns). During the switch
low duration, switch B is turned on which forces SW1
low and charges the flying capacitor, CBST1. This ensures
that the switch A gate driver power supply rail on BST1 is
maintained. The duty cycle of switches C and D are adjusted
to maintain output voltage regulation in boost mode.
Oscillator
The LTC3129-1 operates from an internal oscillator with a
nominal fixed frequency of 1.2MHz. This allows the DC/DC
converter efficiency to be maximized while still using small
external components.
Current Mode Control
The LTC3129-1 utilizes average current mode control for
the pulse width modulator. Current mode control, both
average and the better known peak method, enjoy some
benefits compared to other control methods including:
simplified loop compensation, rapid response to load
transients and inherent line voltage rejection.
Referring to the Block Diagram, a high gain, internally
compensated transconductance amplifier monitors VOUT
through an internal voltage divider. The error amplifier out-
put is used by the current mode control loop to command
the appropriate inductor current level. The inverting input
of the internally compensated average current amplifier is
connected to the inductor current sense circuit. The aver-
age current amplifier’s output is compared to the oscillator
ramps, and the comparator outputs are used to control
the duty cycle of the switch pins on a cycle-by-cycle basis.
The voltage error amplifier monitors the output voltage,
VOUT through the internal voltage divider and makes adjust-
ments to the current command as necessary to maintain
regulation. The voltage error amplifier therefore controls
the outer voltage regulation loop. The average current
amplifier makes adjustments to the inductor current as
directed by the voltage error amplifier output via VC and is
commonly referred to as the inner current loop amplifier.
The average current mode control technique is similar to
peak current mode control except that the average current
amplifier, by virtue of its configuration as an integrator,
controls average current instead of the peak current. This
difference eliminates the peak to average current error
inherent to peak current mode control, while maintaining
most of the advantages inherent to peak current mode
control.
Average current mode control requires appropriate com-
pensation for the inner current loop, unlike peak current
mode control. The compensation network must have high
DC gain to minimize errors between the actual and com-
manded average current level, high bandwidth to quickly
change the commanded current level following transient
load steps and a controlled mid-band gain to provide a
form of slope compensation unique to average current
mode control. The compensation components required
to ensure proper operation have been carefully selected
and are integrated within the LTC3129-1.
Inductor Current Sense and Maximum Output Current
As part of the current control loop required for current
mode control, the LTC3129-1 includes a pair of current
sensing circuits that measure the buck-boost converter
inductor current.
The voltage error amplifier output, VC, is internally clamped
to a nominal level of 0.6V. Since the average inductor
current is proportional to VC, the 0.6V clamp level sets
the maximum average inductor current that can be pro-
grammed by the inner current loop. Taking into account
the current sense amplifier’s gain, the maximum average
LTC3129-1
13
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For more information www.linear.com/LTC3129-1
inductor current is approximately 275mA (typical). In
buck mode, the output current is approximately equal to
the inductor current IL.
IOUT(BUCK) ≈ IL • 0.89
The 90ns SW1/SW2 forced low time on each switching
cycle briefly disconnects the inductor from VOUT and VIN
resulting in about 11% less output current in either buck
or Boost mode for a given inductor current. In boost mode,
the output current is related to average inductor current
and duty cycle by:
IOUT(BOOST) ≈ IL • (1 – D) • Efficiency
where D is the converter duty cycle.
Since the output current in boost mode is reduced by the
duty cycle (D), the output current rating in buck mode is
always greater than in boost mode. Also, because boost
mode operation requires a higher inductor current for a
given output current compared to buck mode, the efficiency
in boost mode will be lower due to higher IL2 • RDS(ON)
losses in the power switches. This will further reduce the
output current capability in boost mode. In either operating
mode, however, the inductor peak-to-peak ripple current
does not play a major role in determining the output cur-
rent capability, unlike peak current mode control.
With peak current mode control, the maximum output
current capability is reduced by the magnitude of inductor
ripple current because the peak inductor current level is the
control variable, but the average inductor current is what
determines the output current. The LTC3129-1 measures
and controls average inductor current, and therefore, the
inductor ripple current magnitude has little effect on the
maximum current capability in contrast to an equivalent
peak current mode converter. Under most conditions in
buck mode, the LTC3129-1 is capable of providing a mini-
mum of 200mA to the load. In boost mode, as described
previously, the output current capability is related to the
boost ratio or duty cycle (D). For example, for a 3.6V VIN
to 5V output application, the LTC3129-1 can provide up
to 150mA to the load. Refer to the Typical Performance
Characteristics section for more detail on output current
capability.
operaTion
Overload Current Limit and IZERO Comparator
The internal current sense waveform is also used by the
peak overload current (IPEAK) and zero current (IZERO) com-
parators. The IPEAK current comparator monitors ISENSE
and turns off switch A if the inductor current level exceeds
its maximum internal threshold, which is approximately
500mA. An inductor current level of this magnitude will
occur during a fault, such as an output short-circuit, or
during large load or input voltage transients.
The LTC3129-1 features near discontinuous inductor
current operation at light output loads by virtue of the
IZERO comparator circuit. By limiting the reverse current
magnitude in PWM mode, a balance between low noise
operation and improved efficiency at light loads is achieved.
The IZERO comparator threshold is set near the zero current
level in PWM mode, and as a result, the reverse current
magnitude will be a function of inductance value and out-
put voltage due to the comparator's propagation delay. In
general, higher output voltages and lower inductor values
will result in increased reverse current magnitude.
In automatic Burst Mode operation (PWM pin low), the
IZERO comparator threshold is increased so that reverse
inductor current does not normally occur. This maximizes
efficiency at very light loads.
Burst Mode OPERATION
When the PWM pin is held low, the LTC3129-1 is con-
figured for automatic Burst Mode operation. As a result,
the buck-boost DC/DC converter will operate with normal
continuous PWM switching above a predetermined mini-
mum output load and will automatically transition to power
saving Burst Mode operation below this output load level.
Note that if the PWM pin is low, reverse inductor current is
not allowed at any load. Refer to the Typical Performance
Characteristics section of this data sheet to determine the
Burst Mode transition threshold for various combinations
of VIN and VOUT. If PWM is low, at light output loads, the
LTC3129-1
14
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For more information www.linear.com/LTC3129-1
operaTion
LTC3129-1 will go into a standby or sleep state when the
output voltage achieves its nominal regulation level. The
sleep state halts PWM switching and powers down all
nonessential functions of the IC, significantly reducing the
quiescent current of the LTC3129-1 to just 1.3µA typical.
This greatly improves overall power conversion efficiency
when the output load is light. Since the converter is not
operating in sleep, the output voltage will slowly decay
at a rate determined by the output load resistance and
the output capacitor value. When the output voltage has
decayed by a small amount, the LTC3129-1 will wake and
resume normal PWM switching operation until the volt-
age on VOUT is restored to the previous level. If the load
is very light, the LTC3129-1 may only need to switch for
a few cycles to restore VOUT and may sleep for extended
periods of time, significantly improving efficiency. If the
load is suddenly increased above the burst transition
threshold, the part will automatically resume continuous
PWM operation until the load is once again reduced.
Note that Burst Mode operation is inhibited until soft-start
is done, the MPPC pin is greater than 1.175V and VOUT
has reached regulation.
Soft-Start
The LTC3129-1 soft-start circuit minimizes input current
transients and output voltage overshoot on initial power up.
The required timing components for soft-start are internal
to the LTC3129-1 and produce a nominal soft-start dura-
tion of approximately 3ms. The internal soft-start circuit
slowly ramps the error amplifier output, VC. In doing so,
the current command of the IC is also slowly increased,
starting from zero. It is unaffected by output loading or
output capacitor value. Soft-start is reset by the UVLO on
both VIN and VCC, the RUN pin and thermal shutdown.
VCC Regulator
An internal low dropout regulator (LDO) generates a nomi-
nal 4.1V VCC rail from VIN. The VCC rail powers the internal
control circuitry and the gate drivers of the LTC3129-1. The
VCC regulator is disabled in shutdown to reduce quiescent
current and is enabled by raising the RUN pin above its
logic threshold. The VCC regulator includes current-limit
protection to safeguard against accidental short-circuiting
of the VCC rail.
Undervoltage Lockout (UVLO)
There are two undervoltage lockout (UVLO) circuits within
the LTC3129-1 that inhibit switching; one that monitors VIN
and another that monitors VCC. Either UVLO will disable
operation of the internal power switches and keep other
IC functions in a reset state if either VIN or VCC are below
their respective UVLO thresholds.
The VIN UVLO comparator has a falling voltage threshold
of 1.8V (typical). If VIN falls below this level, IC operation
is disabled until VIN rises above 1.9V (typical), as long as
the VCC voltage is above its UVLO threshold.
The VCC UVLO has a falling voltage threshold of 2.19V
(typical). If the VCC voltage falls below this threshold, IC
operation is disabled until VCC rises above 2.25V (typical)
as long as VIN is above its nominal UVLO threshold level.
Depending on the particular application, either of these
UVLO thresholds could be the limiting factor affecting the
minimum input voltage required for operation. Because the
VCC regulator uses VIN for its power input, the minimum
input voltage required for operation is determined by the
VCC minimum voltage, as input voltage (VIN) will always
be higher than VCC in the normal (non-bootstrapped)
configuration. Therefore, the minimum VIN for the part
to start up is 2.25V (typical).
In applications where VCC is bootstrapped (powered
through a Schottky diode by either VOUT or an auxiliary
power rail), the minimum input voltage for operation will
be limited only by the VIN UVLO threshold (1.8V typical).
Please note that if the bootstrap voltage is derived from
the LTC3129-1 VOUT and not an independent power rail,
then the minimum input voltage required for initial start-up
is still 2.25V (typical).
Note that if either VIN or VCC are below their UVLO
thresholds, or if RUN is below its accurate threshold of
1.22V (typical), then the LTC3129-1 will remain in a soft
shutdown state, where the VIN quiescent current will be
only 1.9µA typical.
LTC3129-1
15
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For more information www.linear.com/LTC3129-1
operaTion
VOUT Undervoltage
There is also an undervoltage comparator that monitors
the output voltage. Until VOUT reaches 1.15V (typical), the
average current limit is reduced by a factor of two. This
reduces power dissipation in the device in the event of a
shorted output. In addition, N-channel switch D, which
feeds VOUT, will be disabled until VOUT exceeds 1.15V.
RUN Pin Comparator
In addition to serving as a logic level input to enable cer-
tain functions of the IC, the RUN pin includes an accurate
internal comparator that allows it to be used to set custom
rising and falling ON/OFF thresholds with the addition of
an optional external resistor divider. When RUN is driven
above its logic threshold (0.9V typical), the VCC regulator
is enabled, which provides power to the internal control
circuitry of the IC. If the voltage on RUN is increased
further so that it exceeds the RUN comparator’s accurate
analog threshold (1.22V typical), all functions of the buck-
boost converter will be enabled and a start-up sequence
will ensue (assuming the VIN and VCC UVLO thresholds
are satisfied).
If RUN is brought below the accurate comparator threshold,
the buck-boost
converter will inhibit switching, but the VCC
regulator and control circuitry will remain powered unless
RUN is brought below its logic threshold. Therefore, in
order to completely shut down the IC and reduce the VIN
current to 10nA (typical), it is necessary to ensure that
RUN is brought below its worst case low logic threshold of
0.5V. RUN is a high voltage input and can be tied directly
to VIN to continuously enable the IC when the input supply
is present. Also note that RUN can be driven above VIN
or VOUT as long as it stays within the operating range of
the IC (up to 15V).
With the addition of an optional resistor divider as shown
in Figure 2, the RUN pin can be used to establish a user-
programmable turn-on and turn-off threshold. This feature
can be utilized to minimize battery drain below a certain
input voltage, or to operate the converter in a hiccup mode
from very low current sources.
Note that once RUN is above 0.9V typical, the quiescent
input current on VIN (or VCC if back-driven) will increase to
about 1.9µA typical until the VIN and VCC UVLO thresholds
are satisfied.
The converter is enabled when the voltage on RUN exceeds
1.22V (nominal). Therefore, the turn on voltage threshold
on VIN is given by:
VIN(TURN-ON) = 1.22V • (1 + R3/R4)
The RUN comparator includes a built-in hysteresis of
approximately 80mV, so that the turn off threshold will
be 1.14V.
There may be cases due to PCB layout, very large value
resistors for R3 and R4, or proximity to noisy components
where noise pickup may cause the turn-on or turn-off of the
IC to be intermittent. In these cases, a small filter capaci-
tor can be added across R4 to ensure proper operation.
PGOOD Comparator
The LTC3129-1 provides an open-drain PGOOD output that
pulls low if VOUT falls more than 7.5% (typical) below its
programmed value. When VOUT rises to within 5% (typical)
of its programmed value, the internal PGOOD pull-down
will turn off and PGOOD will go high if an external pull-
up resistor has been provided. An internal filter prevents
nuisance trips of PGOOD due to short transients on VOUT.
Note that PGOOD can be pulled up to any voltage, as long
as the absolute maximum rating of 18V is not exceeded,
and as long as the maximum sink current rating is not
exceeded when PGOOD is low. Note that PGOOD will
also be driven low if VCC is below its UVLO threshold or
Figure 2. Accurate RUN Pin Comparator
LTC3129-1
ENABLE SWITCHING
ENABLE LDO AND
CONTROL CIRCUITS
LOGIC THRESHOLD
ACCURATE THRESHOLD
31291 F02
+
–
–
+
0.9V
RUN
1.22V
VIN
R3
R4
LTC3129-1
16
31291fc
For more information www.linear.com/LTC3129-1
operaTion
if the part is in shutdown (RUN below its logic threshold)
while VCC is being held up (or back-driven). PGOOD is
not affected by VIN UVLO or the accurate RUN threshold.
In cases where VCC is not being back-driven in shutdown,
PGOOD will not be held low indefinitely. The internal PGOOD
pull-down will be disabled as the VCC voltage decays below
approximately 1V.
Maximum Power-Point Control (MPPC)
The MPPC input of the LTC3129-1 can be used with an
optional external voltage divider to dynamically adjust
the commanded inductor current in order to maintain
a minimum input voltage when using high resistance
sources, such as photovoltaic panels, so as to maximize
input power transfer and prevent VIN from dropping too
low under load. Referring to Figure 3, the MPPC pin is
internally connected to the noninverting input of a gm
amplifier, whose inverting input is connected to the 1.175V
reference. If the voltage at MPPC, using the external volt-
age divider, falls below the reference voltage, the output of
the amplifier pulls the internal VC node low. This reduces
the commanded average inductor current so as to reduce
the input current and regulate VIN to the programmed
minimum voltage, as given by:
VIN(MPPC) = 1.175V • (1 + R5/R6)
The MPPC feature provides capabilities to the LTC3129-1
that can ease the design of intrinsically safe power sup-
plies. For an example of an application that must operate
from a supply with intentional series resistance, refer to
the application example on the bottom of page 25.
Note that external compensation should not be required
for MPPC loop stability if the input filter capacitor, CIN, is
at least 22µF. See Typical Applications for an example of
external compensation that can be added in applications
where CIN must be less than the recommended minimum
value.
The divider resistor values can be in the megohm range to
minimize the input current in very low power applications.
However, stray capacitance and noise pickup on the MPPC
pin must also be minimized.
The MPPC pin controls the converter in a linear fashion
when using sources that can provide a minimum of 5mA
to 10mA of continuous input current. For operation from
weaker input sources, refer to the Application Information
section to see how the programmable RUN pin can be used
to control the converter in a hysteretic manner to provide
an effective MPPC function for sources that can provide
as little as 5µA or less.
If the MPPC function is not required, the MPPC pin should
be tied to VCC.
VOUT Programming Pins
The LTC3129-1 has a precision internal voltage divider on
VOUT, eliminating the need for high-value external feedback
resistors. This not only eliminates two external compo-
nents, it minimizes no-load quiescent current by using very
Figure 3. MPPC Amplifier with External Resistor Divider
LTC3129-1
1.175V
VC
CURRENT
COMMAND
VOLTAGE
ERROR AMP
31291 F03
MPPC
R5
R6
RS
VSOURCE
*CIN
VIN
VIN
+
–
+
–
+
–
* CIN SHOULD BE AT
LEAST 22µF FOR
MPPC APPLICATIONS
LTC3129-1
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For more information www.linear.com/LTC3129-1
operaTion
high resistance values that would not be practical due to the
effects of noise and board leakages that would cause VOUT
regulation errors. The tap point on this divider is digitally
selected by using the VS1, VS2 and VS3 pins to program
one of eight fixed output voltages. The VS pins should be
grounded or connected to VCC to select the desired output
voltage, according to the following table. The VS1, VS2
and VS3 pins can also be driven by external logic signals
as long as the absolute maximum voltage ratings are not
exceeded. Note however that driving any of the voltage
select pins high to a voltage less than the VCC operating
voltage will result in increased quiescent current. Also
note that if the VS3 pin is driven above VCC, an external
1M resistor should be added in series. For other output
voltages, refer to the LTC3129 which has a feedback pin,
allowing any output voltage from 1.4V to 15.75V.
VOUT Program Settings for the LTC3129-1
VS3 PIN VS2 PIN VS1 PIN VOUT
0 0 0 2.5V
0 0 VCC 3.3V
0 VCC 0 4.1V
0 VCC VCC 5.0V
VCC 0 0 6.9V
VCC 0 VCC 8.2V
VCC VCC 0 12V
VCC VCC VCC 15V
Note that in shutdown, or if VCC is below its UVLO thresh-
old, the internal voltage divider on VOUT is automatically
disconnected to eliminate any current draw on VOUT.
Thermal Considerations
The power switches of the LTC3129-1 are designed to op-
erate continuously with currents up to the internal current
limit thresholds. However, when operating at high current
levels, there may be significant heat generated within the
IC. In addition, the VCC regulator can also generate wasted
heat when VIN is very high, adding to the total power Figure 4. Typical 2-Layer PC Board Layout (MSE Package)
dissipation of the IC. As described elsewhere in this data
sheet, bootstrapping of the VCC for 5V output applications
can essentially eliminate the VCC power dissipation term
and significantly improve efficiency. As a result, careful
consideration must be given to the thermal environment
of the IC in order to provide a means to remove heat from
the IC and ensure that the LTC3129-1 is able to provide
its full rated output current. Specifically, the exposed die
attach pad of both the QFN and MSE packages must be
soldered to a copper layer on the PCB to maximize the
conduction of heat out of the IC package. This can be ac-
complished by utilizing multiple vias from the die attach
pad connection underneath the IC package to other PCB
layer(s) containing a large copper plane. A typical board
layout incorporating these concepts is shown in Figure 4.
If the IC die temperature exceeds approximately 180°C,
overtemperature shutdown will be invoked and all switching
will be inhibited. The part will remain disabled until the die
temperature cools by approximately 10°C. The soft-start
circuit is re-initialized in over temperature shutdown to
provide a smooth recovery when the IC die temperature
cools enough to resume operation.
GND VIN
GND
31291 F04
VOUT
COUT
CIN
VCC
CBST1
CBST2
L
LTC3129-1
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For more information www.linear.com/LTC3129-1
applicaTions inForMaTion
A standard application circuit for the LTC3129-1 is shown
on the front page of this data sheet. The appropriate selec-
tion of external components is dependent upon the required
performance of the IC in each particular application given
considerations and trade-offs such as PCB area, input
and output voltage range, output voltage ripple, transient
response, required efficiency, thermal considerations and
cost. This section of the data sheet provides some basic
guidelines and considerations to aid in the selection of
external components and the design of the applications
circuit, as well as more application circuit examples.
VCC Capacitor Selection
The VCC output of the LTC3129-1 is generated from VIN
by a low dropout linear regulator. The VCC regulator has
been designed for stable operation with a wide range
of output capacitors. For most applications, a low ESR
capacitor of at least 2.2µF should be used. The capacitor
should be located as close to the VCC pin as possible and
connected to the VCC pin and ground through the shortest
traces possible. VCC is the regulator output and is also the
internal supply pin for the LTC3129-1 control circuitry as
well as the gate drivers and boost rail charging diodes.
The VCC pin is not intended to supply current to other
external circuitry.
Inductor Selection
The choice of inductor used in LTC3129-1 application cir-
cuits influences the maximum deliverable output current,
the converter bandwidth, the magnitude of the inductor
current ripple and the overall converter efficiency. The
inductor must have a low DC series resistance, when
compared to the internal switch resistance, or output
current capability and efficiency will be compromised.
Larger inductor values reduce inductor current ripple
but may not increase output current capability as is the
case with peak current mode control as described in the
Maximum Output Current section. Larger value inductors
also tend to have a higher DC series resistance for a given
case size, which will have a negative impact on efficiency.
Larger values of inductance will also lower the right half
plane (RHP) zero frequency when operating in boost mode,
which can compromise loop stability. Nearly all LTC3129-1
application circuits deliver the best performance with
an inductor value between 3.3µH and 10µH. Buck mode
only applications can use the larger inductor values as
they are unaffected by the RHP zero, while mostly boost
applications generally require inductance on the low end
of this range depending on how large the step-up ratio is.
Regardless of inductor value, the saturation current rating
should be selected such that it is greater than the worst
case average inductor current plus half of the ripple cur-
rent. The peak-to-peak inductor current ripple for each
operational mode can be calculated from the following
formula, where f is the switching frequency (1.2MHz), L
is the inductance in µH and tLOW is the switch pin mini-
mum low time in µs. The switch pin minimum low time
is typically 0.09µs.
ΔIL(P−P)(BUCK) =VOUT
L
V
IN – VOUT
V
IN
⎛
⎝
⎜⎞
⎠
⎟1
f– tLOW
⎛
⎝
⎜⎞
⎠
⎟A
ΔIL(P−P)(BOOST) =V
IN
L
VOUT – V
IN
VOUT
⎛
⎝
⎜⎞
⎠
⎟1
f– tLOW
⎛
⎝
⎜⎞
⎠
⎟A
It should be noted that the worst-case peak-to-peak in-
ductor ripple current occurs when the duty cycle in buck
mode is minimum (highest VIN) and in boost mode when
the duty cycle is 50% (VOUT = 2 • VIN). As an example, if
VIN (minimum) = 2.5V and VIN (maximum) = 15V, VOUT
= 5V and L = 10µH, the peak-to-peak inductor ripples at
the voltage extremes (15V VIN for buck and 2.5V VIN for
boost) are:
BUCK = 248mA peak-to-peak
BOOST = 93mA peak-to-peak
One half of this inductor ripple current must be added to
the highest expected average inductor current in order to
select the proper saturation current rating for the inductor.
To avoid the possibility of inductor saturation during load
transients, an inductor with a saturation current rating of
at least 600mA is recommended for all applications.
In addition to its influence on power conversion efficiency,
the inductor DC resistance can also impact the maximum
output current capability of the buck-boost converter
particularly at low input voltages. In buck mode, the
output current of the buck-boost converter is primarily
limited by the inductor current reaching the average cur-
rent limit threshold. However, in boost mode, especially
LTC3129-1
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For more information www.linear.com/LTC3129-1
at large step-up ratios, the output current capability can
also be limited by the total resistive losses in the power
stage. These losses include, switch resistances, inductor
DC resistance and PCB trace resistance. Avoid inductors
with a high DC resistance (DCR) as they can degrade the
maximum output current capability from what is shown
in the Typical Performance Characteristics section and
from the Typical Application circuits.
As a guideline, the inductor DCR should be significantly
less than the typical power switch resistance of 750mΩ
each. The only exceptions are applications that have a
maximum output current requirement much less than
what the LTC3129-1 is capable of delivering. Generally
speaking, inductors with a DCR in the range of 0.15Ω to
0.3Ω are recommended. Lower values of DCR will improve
the efficiency at the expense of size, while higher DCR
values will reduce efficiency (typically by a few percent)
while allowing the use of a physically smaller inductor.
Different inductor core materials and styles have an impact
on the size and price of an inductor at any given current
rating. Shielded construction is generally preferred as it
minimizes the chances of interference with other circuitry.
The choice of inductor style depends
upon the price, sizing,
and EMI requirements of a particular application. Table 2
provides a wide sampling of inductors that are well suited
to many LTC3129-1 applications.
Table 2. Recommended Inductors
VENDOR PART
Coilcraft
www.coilcraft.com
EPL2014, EPL3012, EPL3015, XFL3012
LPS3015, LPS3314
Coiltronics
www.cooperindustries.com
SDH3812, SD3814
SD3114, SD3118
Murata
www.murata.com
LQH3NP
LQH32P
LQH44P
Sumida
www.sumida.com
CDRH2D16, CDRH2D18
CDRH3D14, CDRH3D16
Taiyo-Yuden
www.t-yuden.com
NR3012T, NR3015T, NRS4012T
BRC2518
TDK
www.tdk.com
VLS3012, VLS3015
VLF302510MT, VLF302512MT
Toko
www.tokoam.com
DB3015C, DB3018C, DB3020C
DP418C, DP420C, DEM2815C,
DFE322512C, DFE252012C
Würth
www.we-online.com
WE-TPC 2813, WE-TPC 3816,
WE-TPC 2828
applicaTions inForMaTion
Recommended inductor values for different operating
voltage ranges are given in Table 3. These values were
chosen to minimize inductor size while maintaining an
acceptable amount of inductor ripple current for a given
VIN and VOUT range.
Table 3. Recommended Inductor and Output Capacitor Values
VIN AND VOUT RANGE RECOMMENDED
INDUCTOR
VALUES
MAXIMUM RECOMMENDED
TOTAL OUTPUT CAPACITOR
VALUE FOR PWM MODE
OPERATION AT LIGHT LOAD
(<15mA, PWM PIN HIGH)
VIN and VOUT Both < 4.5V 3.3µH to 4.7µH 10µF
VIN and VOUT Both < 8V 4.7µH to 6.8µH 10µF
VIN and VOUT Both < 11V 6.8µH to 8.2µH 10µF
VIN and VOUT Up to 15V 8.2µH to 10µH 10µF
Due to the fixed, internal loop compensation and feedback
divider provided by the LTC3129-1, there are limitations to
the maximum recommended total output capacitor value in
applications that must operate in PWM mode at light load
(PWM pin pulled high with minimum load currents less
than ~15mA). In these applications, a maximum output
capacitor value, shown in Table 3, is recommended. For
applications that must operate in PWM mode at light load
with higher values of output capacitance, the LTC3129 is
recommended. Its external feedback pin allows the use
of additional feedforward compensation for improved
light-load stability under these conditions.
Note that for applications where Burst Mode operation
is enabled (PWM pin grounded), the output capacitor
value can be increased without limitation regardless of
the minimum load current or inductor value.
Output Capacitor Selection
A low effective series resistance (ESR) output capacitor
of 4.7µF minimum should be connected at the output of
the buck-boost converter in order to minimize output volt-
age ripple. Multilayer ceramic capacitors are an excellent
option as they have low ESR and are available in small
footprints. The capacitor value should be chosen large
enough to reduce the output voltage ripple to acceptable
levels. Neglecting the capacitor’s ESR and ESL (effec-
tive series inductance), the peak-to-peak output voltage
ripple in PWM mode can be calculated by the following
LTC3129-1
20
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For more information www.linear.com/LTC3129-1
formula, where f is the frequency in MHz (1.2MHz), COUT
is the capacitance in µF, tLOW is the switch pin minimum
low time in µs (0.09µs typical) and ILOAD is the output
current in amperes.
ΔV
P−P(BUCK) =
I
LOAD
t
LOW
COUT
V
ΔV
P−P(BOOST) =ILOAD
fCOUT
VOUT – VIN +tLOWfVIN
VOUT
⎛
⎝
⎜⎞
⎠
⎟V
Examining the previous equations reveals that the output
voltage ripple increases with load current and is gener-
ally higher in boost mode than in buck mode. Note that
these equations only take into account the voltage ripple
that occurs from the inductor current to the output being
discontinuous. They provide a good approximation to the
ripple at any significant load current but underestimate the
output voltage ripple at very light loads where the output
voltage ripple is dominated by the inductor current ripple.
In addition to the output voltage ripple generated across
the output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportional to the series resistance of the output capacitor
and is given by the following expressions where RESR is
the series resistance of the output capacitor and all other
terms as previously defined.
ΔV
P−P(BUCK) =
I
LOAD
R
ESR
1– tLOWf≅ILOADRESR V
ΔV
P−P(BOOST) =ILOADRESRVOUT
VIN 1– tLOWf
( )
≅ILOADRESR
VOUT
V
IN
⎛
⎝
⎜⎞
⎠
⎟V
In most LTC3129-1 applications, an output capacitor be-
tween 10µF and 22µF will work well. To minimize output
ripple in Burst Mode operation, values of 22µF operation
or larger are recommended.
Input Capacitor Selection
The VIN pin carries the full inductor current and provides
power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 4.7µF
should be located as close to the VIN pin as possible. The
traces connecting this capacitor to VIN and the ground
plane should be made as short as possible.
When powered through long leads or from a power source
with significant resistance, a larger value bulk input ca-
pacitor may be required and is generally recommended.
In such applications, a 47µF to 100µF low-ESR electrolytic
capacitor in parallel with a 1µF ceramic capacitor generally
yields a high performance, low cost solution.
Note that applications using the MPPC feature should
use a minimum CIN of 22µF. Larger values can be used
without limitation.
Recommended Input and Output Capacitor Types
The capacitors used to filter the input and output of the
LTC3129-1 must
have low ESR and must be rated to handle
the AC currents generated by the switching converter.
This is important to maintain proper functioning of the
IC and to reduce output voltage ripple. There are many
capacitor types that are well suited to these applications
including multilayer ceramic, low ESR tantalum, OS-CON
and POSCAP technologies. In addition, there are certain
types of electrolytic capacitors such as solid aluminum
organic polymer capacitors that are designed for low ESR
and high AC currents and these are also well suited to
some LTC3129-1 applications. The choice of capacitor
technology is primarily dictated by a trade-off between
size, leakage current and cost. In backup power applica-
tions, the input or output capacitor might be a super or
ultra capacitor with a capacitance value measuring in the
farad range. The selection criteria in these applications
are generally similar except that voltage ripple is generally
not a concern. Some capacitors exhibit a high DC leak-
age current which may preclude their consideration for
applications that require a very low quiescent current in
Burst Mode operation. Note that ultra capacitors may have
applicaTions inForMaTion
LTC3129-1
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For more information www.linear.com/LTC3129-1
applicaTions inForMaTion
a rather high ESR, therefore a 4.7µF (minimum) ceramic
capacitor is recommended in parallel, close to the IC pins.
Ceramic capacitors are often utilized in switching con-
verter applications due to their small size, low ESR and
low leakage currents. However, many ceramic capacitors
intended for power applications experience a significant
loss in capacitance from their rated value as the DC bias
voltage on the capacitor increases. It is not uncommon for
a small surface mount capacitor to lose more than 50%
of its rated capacitance when operated at even half of its
maximum rated voltage. This effect is generally reduced
as the case size is increased for the same nominal value
capacitor. As a result, it is often necessary to use a larger
value capacitance or a higher voltage rated capacitor than
would ordinarily be required to actually realize the intended
capacitance at the operating voltage of the application. X5R
and X7R dielectric types are recommended as they exhibit
the best performance over the wide operating range and
temperature of the LTC3129-1. To verify that the intended
capacitance is achieved in the application circuit, be sure
to consult the capacitor vendor’s curve of capacitance
versus DC bias voltage.
Using the Programmable RUN Function to Operate
from Extremely Weak Input Sources
Another application of the programmable RUN pin is that
it can be used to operate the converter in a hiccup mode
from extremely low current sources. This allows opera-
tion from sources that can only generate microamps of
output current, and would be far too weak to sustain
normal steady-state operation, even with the use of the
MPPC pin. Because the LTC3129-1 draws only 1.9µA
typical from VIN until it is enabled, the RUN pin can be
programmed to keep the IC disabled until VIN reaches the
programmed voltage level. In this manner, the input source
can trickle-charge an input storage capacitor, even if it
can only supply microamps of current, until VIN reaches
the turn-on threshold set by the RUN pin divider. The
converter will then be enabled, using the stored charge
in the input capacitor, until VIN drops below the turn-off
threshold, at which point the converter will turn off and
the process will repeat.
This approach allows the converter to run from weak
sources such as thin-film solar cells using indoor lighting.
Although the converter will be operating in bursts, it is
enough to charge an output capacitor to power low duty
cycle loads, such as wireless sensor applications, or to
trickle charge a battery. In addition, note that the input
voltage will be cycling (with a small ripple as set by the
RUN hysteresis) about a fixed voltage, as determined by
the divider. This allows the high impedance source to
operate at the programmed optimal voltage for maximum
power transfer.
When using high value divider resistors (in the MΩ range)
to minimize current draw on VIN, a small noise filter ca-
pacitor may be necessary across the lower divider resis-
tor to prevent noise from erroneously tripping the RUN
comparator. The capacitor value should be minimized
so as not to introduce a time delay long enough for the
input voltage to drop significantly below the desired VIN
threshold before the converter is turned off. Note that
larger VIN decoupling capacitor values will minimize this
effect by providing more holdup time on VIN.
Programming the MPPC Voltage
As discussed in the previous section, the LTC3129-1 in-
cludes an MPPC function to optimize performance when
operating from voltage sources with relatively high source
resistance. Using an external voltage divider from VIN, the
MPPC function takes control of the average inductor current
when necessary to maintain a minimum input voltage, as
programmed by the user. Referring to Figure 3:
VIN(MPPC) = 1.175V • (1 + R5/R6)
This is useful for such applications as photovoltaic pow-
ered converters, since the maximum power transfer point
occurs when the photovoltaic panel is operated at about
75% of its open-circuit voltage. For example, when operat-
ing from a photovoltaic panel with an open-circuit voltage
of 5V, the maximum power transfer point will be when
the panel is loaded such that its output voltage is about
3.75V. Choosing values of 2MΩ for R5 and 909k for R6
will program the MPPC function to regulate the maximum
input current so as to maintain VIN at a minimum of 3.74V
(typical). Note that if the panel can provide more power
than the LTC3129-1 can draw, the input voltage will rise
above the programmed MPPC point. This is fine as long
as the input voltage doesn't exceed 15V.
LTC3129-1
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Figure 5. Example of VCC Bootstrap
For weak input sources with very high resistance (hun-
dreds of Ohms or more), the LTC3129-1 may still draw
more current than the source can provide, causing VIN to
drop below the UVLO threshold. For these applications, it
is recommended that the programmable RUN feature be
used, as described in the previous section.
MPPC Compensation and Gain
When using MPPC, there are a number of variables that
affect the gain and phase of the input voltage control
loop. Primarily these are the input capacitance, the MPPC
divider ratio and the VIN source resistance (or current). To
simplify the design of the application circuit, the MPPC
control loop in the LTC3129 is designed with a relatively
low gain, such that external MPPC loop compensation is
generally not required when using a VIN capacitor value
of at least 22µF. The gain from the MPPC pin to the in-
ternal VC control voltage is about 12, so a drop of 50mV
on the MPPC pin (below the 1.175V MPPC threshold),
corresponds to a 600mV drop on the internal VC voltage,
which reduces the average inductor current all the way
to zero. Therefore, the programmed input MPPC voltage
will be maintained within about 4% over the load range.
Note that if large-value VIN capacitors are used (which may
have a relatively high ESR) a small ceramic capacitor of
at least 4.7µF should be placed in parallel across the VIN
input, near the VIN pin of the IC.
Bootstrapping the VCC Regulator
The high and low side gate drivers are powered through
the VCC rail, which is generated from the input voltage, VIN,
through an internal linear regulator. In some applications,
especially at high input voltages, the power dissipation
in the linear regulator can become a major contributor to
thermal heating of the IC and overall efficiency. The Typical
Performance Characteristics section provides data on the
VCC current and resulting power loss versus VIN and VOUT.
A significant performance advantage can be attained in high
VIN applications where converter output voltage (VOUT) is
programmed to 5V, if VOUT is used to power the VCC rail.
Powering VCC in this manner is referred to as bootstrap-
ping. This can be done by connecting a Schottky diode
(such as a BAT54) from VOUT to VCC as shown in Figure 5.
With the bootstrap diode installed, the gate driver currents
are supplied by the buck-boost converter at high efficiency
rather than through the internal linear regulator. The in-
ternal linear regulator contains reverse blocking circuitry
that allows VCC to be driven above its nominal regulation
level with only a very slight amount of reverse current.
Please note that the bootstrapping supply (either VOUT or
a separate regulator) must be limited to less than 5.7V so
as not to exceed the maximum VCC voltage of 5.5V after
the diode drop.
By maintaining VCC above its UVLO threshold, bootstrap-
ping, even to a 3.3V output, also allows operation down
to the VIN UVLO threshold of 1.8V (typical).
applicaTions inForMaTion
31291 F05
LTC3129-1
VOUT VOUT
BAT54
COUT
VCC
2.2µF
Sources of Small Photovoltaic Panels
A list of companies that manufacture small solar panels
(sometimes referred to as modules or solar cell arrays)
suitable for use with the LTC3129-1 is provided in Table 4.
Table 4. Small Photovoltaic Panel Manufacturers
Sanyo http://panasonic.net/energy/amorton/en/
PowerFilm http://www.powerfilmsolar.com/
IXYS
Corporation
http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx
G24
Innovations
http://www.g24i.com/
LTC3129-1
23
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For more information www.linear.com/LTC3129-1
Typical applicaTions
3.3V Converter Provides Extremely Long Run Time in Low Drain Applications Using Lithium Thionyl Chloride Battery
Low Noise, Fixed Frequency, Wide VIN Range 12V Converter
BST1
VOUT VOUT
12V
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD PGOOD
GND
VCC
VIN
VIN
2.42V TO 15V
RUN
MPPC
PWM
VS1
VCC
VS2
VS3
22nF 6.8µH
1M
4.7µF
10µF
16V
2.2µF
31291 TA02
PGND
VIN < 12V, IOUT = 30mA
VIN > 12V, IOUT = 200mA
BST1
VOUT VOUT
3.3V
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD PGOOD
GND
VCC
VIN
VIN
RUN
MPPC
PWM
VS1
VCC
VS2
VS3
22nF 4.2µH
1M
Li-SoCl2
AA
SAFT LS14500
TADIRAN TL-4903
22µF47µF
2.2µF
31291 TA03
PGND
RUN TIME
> 100,000 HRS (11.4 YEARS) AT 10µA (33µW) AVERAGE LOAD
> 34,000 HRS (3.9 YEARS) AT 50µA (165µW) AVERAGE LOAD
LTC3129-1
24
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For more information www.linear.com/LTC3129-1
Typical applicaTions
15V Converter Powered from Flexible Solar Panel
Hiccup Converter Keeps Li-Ion Battery Charged with Indoor Lighting Average IOUT vs Light Level
(Indoors)
IOUT vs Light Level (Daylight)
LIGHT LEVEL (Lx)
IOUT (mA)
31291 TA04b
100
10
1
10000 100000 1000000
LIGHT LEVEL (Lx)
IOUT (µA)
31291 TA05b
1000
100
10
100 1000 10000
BST1
VOUT VOUT
15V
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD
GND
1M
VCC
VCC
VIN
VIN VMPPC = 6V
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 10µH
243k
47µF
11.4cm × 15cm
10µF
2.2µF
31291 TA04a
PGND
PowerFilm
MPT6-150
SOLAR
MODULE
IOUT = 32mA IN FULL SUN
BST1
VOUT
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD
GND
4.42M
VCC
VIN
VIN UVLO = 3.5V
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 3.3µH
VCC
2.37M
10pF
4.9cm × 5.8cm
470µF
6.3V
4.7µF
4.7µF
2.2µF
31291 TA05a
PGND
PV PANEL
SANYO
AM-1815
VOUT
4.1V
Li-Ion
+
LTC3129-1
25
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For more information www.linear.com/LTC3129-1
5V Converter Operates from Tw o to Eight AA or AAA Cells Using Bootstrap Diode to Increase Efficiency
at High VIN and Extend Operation at Low VIN
Typical applicaTions
3.3V Converter Uses MPPC Function to Work with High Resistance Battery Pack
BST1
VOUT VOUT
3.3V
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD
GND
10Ω
VCC
VCC
VIN
VIN VMPPC = 2.9V
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 3.3µH
1.5M
RC
150k
1M
10µF
CC
33pF
10µF
2.2µF
31291 TA07
PGND
IOUT = 100mA
1.5V
1.5V
1.5V
NOTE: RC AND CC HAVE BEEN ADDED FOR IMPROVED MPPC LOOP STABILITY WHEN USING AN INPUT
CAPACITOR VALUE LESS THAN THE RECOMMENDED MINIMUM OF 22µF
VCC
BST1
VOUT VOUT
5V
SW1 SW2
LTC3129-1
22nF
BST2
PGOOD
GND
VCC
VIN
VIN
1.92V TO 15V
RUN
22nF 8.2µH
BAT54
22µF
2.2µF
31291 TA06
PGND
VIN < 5V, IOUT = 100mA
VIN > 5V, IOUT = 200mA
TWO TO EIGHT
AA OR AAA
BATTERIES
MPPC
PWM
VS1
VS2
VS3
10µF
AFTER STARTUP
LTC3129-1
26
31291fc
For more information www.linear.com/LTC3129-1
Typical applicaTions
Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications
Percentage of Added Battery Life vs Light Level and Load
(PowerFilm SP4.2-37, 30sq cm Panel)
LIGHT LEVEL (Lx)
100
ADDED BATTERY LIFE (%)
1000
100
10
1
31291 TA09b
10,0001,000
AVERAGE LOAD = 165µW
AVERAGE LOAD = 330µW
AVERAGE LOAD = 660µW
AVERAGE LOAD = 1650µW
AVERAGE LOAD = 3300µW
BST1
VOUT
SW1 SW2
LTC3129-1
22nF
3.30V
BST2
PGOOD
GND
VCC
VCC
VIN
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 3.3µH
10pF
2.2µF
VOUT
BAT54
470µF
6.3V
74LVC2G04
31291 TA09
PGND
FDC6312P
DUAL PMOS
PV PANEL
SANYO AM-1815
OR
PowerFilm SP4.2-37
4.7µF
VIN UVLO = 3.7V
4.99M 2.43M
D1 D2
S2S1
G2
CR2032
3V COIN CELL
VOUT
3V TO 3.3V
G1
2.43M
22µF
2.2µF
+
LTC3129-1
27
31291fc
For more information www.linear.com/LTC3129-1
package DescripTion
Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings.
3.00 ±0.10
(4 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
1.65 ±0.05
(4 SIDES)
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
PIN 1
TOP MARK
(NOTE 6)
0.40 ±0.10
BOTTOM VIEW—EXPOSED PAD
1.65 ±0.10
(4-SIDES)
0.75 ±0.05 R = 0.115
TYP
0.25 ±0.05
1
PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER
15 16
2
0.50 BSC
0.200 REF
2.10 ±0.05
3.50 ±0.05
0.70 ±0.05
0.00 – 0.05
(UD16 VAR A) QFN 1207 REV A
0.25 ±0.05
0.50 BSC
PACKAGE OUTLINE
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1700 Rev A)
Exposed Pad Variation AA
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1700 Rev A)
Exposed Pad Variation AA
LTC3129-1
28
31291fc
For more information www.linear.com/LTC3129-1
package DescripTion
Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings.
MSOP (MSE16) 0213 REV F
0.53 ±0.152
(.021 ±.006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
16
16151413121110
12345678
9
9
18
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.10
(.201)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ±0.127
(.035 ±.005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ±0.038
(.0120 ±.0015)
TYP
0.50
(.0197)
BSC
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102
(.065 ±.004)
0.1016 ±0.0508
(.004 ±.002)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0.280 ±0.076
(.011 ±.003)
REF
4.90 ±0.152
(.193 ±.006)
DETAIL “B”
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
LTC3129-1
29
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For more information www.linear.com/LTC3129-1
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision hisTory
REV DATE DESCRIPTION PAGE NUMBER
A 5/14 Clarified VCC Leakage to VIN if VCC > VIN: from –7µA to –27µA 4
B 10/14 Clarified PGOOD Pin Description
Clarified Operation Paragraph
9
16
C 10/15 Changed MAX VCC Current Limit
Modified MPPC section
Modified Table 4
4
16
22
LTC3129-1
30
31291fc
For more information www.linear.com/LTC3129-1
LINEAR TECHNOLOGY CORPORATION 2013
LT 1015 REV C • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/3129-1
Typical applicaTion
TEG Powered Converter Operates from a 10°C Temperature Differential and Provides 3.3V at 25mA
for 50ms Every 15 Seconds for a Wireless Sensor
relaTeD parTs
PART NUMBER DESCRIPTION COMMENTS
LTC3103 15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current
VIN(MIN) = 2.2V, VIN(MAX) = 15V, VOUT(MIN) = 0.8V, IQ = 1.8µA,
ISD <1µA, 3mm × 3mm DFN-10, MSOP-10 Packages
LTC3104 15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current and 10mA LDO
VIN(MIN) = 2.2V, VIN(MAX) = 15V, VOUT(MIN) = 0.8V, IQ = 2.8µA,
ISD <1µA, 4mm × 3mm DFN-14, MSOP-16 Packages
LTC3105 400mA Step-up Converter with MPPC and 250mV Start-Up VIN(MIN) = 0.2V, VIN(MAX) = 5V, VOUT(MIN) = 0 5.25VMAX, IQ = 22µA,
ISD <1µA, 3mm × 3mm DFN-10/MSOP-12 Packages
LTC3112 15V, 2.5A, 750kHz Monolithic Synch Buck/Boost VIN(MIN) = 2.7V, VIN(MAX) = 15V, VOUT(MIN) = 2.7V to 14V, IQ = 50µA,
ISD <1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages
LTC3115-1 40V, 2A, 2MHz Monolithic Synch Buck/Boost VIN(MIN) = 2.7V, VIN(MAX) = 40V, VOUT(MIN) = 2.7V to 40V, IQ = 50µA,
ISD <1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages
LTC3531 5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MIN) = 2V to 5V, IQ = 16µA,
ISD <1µA, 3mm × 3mm DFN-8 and ThinSOT Packages
LTC3388-1/
LTC3388-3
20V, 50mA High Efficiency Nano Power Step-Down Regulator VIN(MIN) = 2.7V, VIN(MAX) =20V, VOUT(MIN) = Fixed 1.1V to 5.5V,
IQ = 720nA, ISD = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages
LTC3108/
LTC3108-1
Ultralow Voltage Step-Up Converter and Power Manager VIN(MIN) = 0.02V, VIN(MAX) = 1V, VOUT(MIN) = Fixed 2.35V to 5V,
IQ = 6µA, ISD <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages
LTC3109 Auto-Polarity, Ultralow Voltage Step-Up Converter and Power
Manager
VIN(MIN) = 0.03V, VIN(MAX) = 1V, VOUT(MIN) = Fixed 2.35V to 5V,
IQ = 7µA, ISD <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages
LTC3588-1 Piezo Electric Energy Harvesting Power Supply VIN(MIN) = 2.7V, VIN(MAX) = 20V, VOUT(MIN) = Fixed 1.8V to 3.6V,
IQ = 950nA, ISD 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages
LTC4070 Li-Ion/Polymer Low Current Shunt Battery Charger System VIN(MIN) = 450nA to 50mA, VFLOAT + 4.0V, 4.1V, 4.2V, IQ = 300nA,
2mm × 3mm DFN-8, MSOP-8 Packages
BST1
VOUT VOUT
3.3V
SW1 SW2
LTC3129-1
BAT54
22nF
BST2
PGOOD PGOOD
GND
VCC
VCC
VIN
RUN
MPPC
PWM
VS1
VS2
VS3
22nF 4.7µH
3.01M
1M
1M
1N4148
1M
10µF
31291 TA08
PGND
2.2µF
1µF
VOUT VAUX
VAUX
VOUT2_EN
VOUT2
VSTORE
LTC3109
PGOOD
VLDO
C2B
330k
1nF
C2A
C1B
C1A
VS2
VINB
VINA
SWB
SWA
VS1
VAUX
1µF
33nF
220µF
MARLOW NL1025T
TEG MOUNTED TO
A HEAT SINK WITH
LESS THAN 15°C/W
THERMAL RESISTANCE
COILCRAFT
LPR6235-123QML
1:50
•• 470µF
6.3V
+
+
10pF
