TDA2030L-TB5-T - Unisonic Technologies Co., Ltd.

UNISONIC TECHNOLOGIES CO., LTD

TDA2030 LINEAR INTEGRATED CIRCUIT

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14W HI-FI AUDIO AMPLIFIER

 DESCRIPTION

The UTC TDA2030 is a monolithic audio power amplifier

integrated circuit.

 FEATURES

* Very low external component required.

* High current output and high operating voltage.

* Low harmonic and crossover distortion.

* Built-in Over temperature protection.

* Short circuit protection between all pins.

* Safety Operating Area for output transistors.

*Pb-free plating product number: TDA2030L

 ORDERING INFORMATION

Ordering Number

Normal Lead Free Plating Package Packing

TDA2030-TA5-T TDA2030L-TA5-T TO-220-5 Tube

TDA2030-TB5-T TDA2030L-TB5-T TO-220B Tube

 PIN CONFIGURATION

PIN NO. PIN NAME

1 Non inverting input

2 Inverting input

3 -VS

4 Output

5 +VS

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 ABSOLUTE MAXIMUM RATINGS (Ta=25°C)

PARAMETER SYMBOL RATINGS UNIT

Supply Voltage Vs ±18 V

Input Voltage VIN Vs V

Differential Input Voltage VI(DIFF) ±15 V

Peak Output Current(internally lim ited) IOUT 3.5 A

Total Power Dissipation at Tc=90°C PD 20 W

Junction Temperature TJ -40~+150 °C

Storage Temperature TSTG -40~+150 °C

Note: Absolute maximum ratings are thos e values beyond which the device coul d be permanently damage d.

Absolute maximum ratings are stress ratings only and functional device operation is not implied.

 ELECTRICAL CHARACTERISTICS (Refer to the test circuit, Vs =±16V,Ta=25°C)

PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNIT

Supply Voltage Vs ±6 ±18 V

Quiescent Drain Current IQ 40 60 mA

Input Bias Current II(BIAS) 0.2 2 µA

Input Offset Voltage VI(OFF)

±2 ±20 MV

Input Offset Current II(OFF) Vs=±18v ±20 ±200 NA

Power Bandwidth BW P

OUT=12W, RL=4Ω, Gv=30dB 10~140,000 Hz

RL=4Ω 12 14 W d=0.5%, Gv=30dB

f=40Hz to 15KHz RL=8Ω 8 9 W

RL=4Ω 18 W

Output Power POUT d=10%, Gv=30dB

f=1KHz RL=8Ω 11 W

Open Loop Voltage Gain Gvo 90 dB

Closed Loop Voltage Gain Gvc f=1kHz 29.5 30 30 .5 dB

POUT=0.1 to 12W, RL=4Ω

f=40Hz to 15KHz, Gv=30dB 0.2 0.5 %

Distortion THD

POUT=0.1 to 8W, RL=8Ω

f=40Hz to 15KHz, Gv=30dB 0.1 0.5 %

Input Noise Voltage eN B= 22Hz to 22kHz 3 10 µV

Input Noise Current iN B= 22Hz to 22kHz 80 200 pA

Input Resistance(pin 1) RIN 0.5 5 MΩ

Supply Voltage Rejection SVR RL=4Ω, Gv=30dB

Rg=22kΩ, fripple=100Hz,

Vripple=0.5Veff 40 50 dB

Thermal Shut-Down Junction

Temperature TJ 145 °C

TDA2030 LINEAR INTEGRATED CIRCUIT

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 TEST CIRCUIT

 APPLICATION CIRCUIT

UTC

TDA 2030

1

23

5

4

Vi

+Vs

-Vs

C1

1 F

C2

22 FC6

100 FC4

100nF C7

220nF

C3

100nF

C5

220 FD1

1N4001

D1

1N4001

R3

22k

R1

13k

R4

1RL

R3

680

TDA2030 LINEAR INTEGRATED CIRCUIT

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 TYPICAL CHARACTERISTICS

102103104105106107

101

-60

-20

20

60

100

140

Phase

Gain

Gv(dB)

180

90

0

Phase

Fig.2 Open loop frequency response

24 28 32 36 40 44

24

4

8

12

16

20

RL=4

RL=8

Gv=26dB

d=0.5%

f=40 to 15kHz

Fig.3 Output power vs. Supply voltage

Fig.4 Total harmonic distortion

vs. output power Fig.5 Two tone CCIF intermodulation

distortion

10

-1 100101102

10

-2

10

-2

10

-1

100

101

102

Vs=38V

RL=8

Vs=32V

RL=4

f=15kHz

f=1kHz

Gv=26dB

101102

10

-2

10

-1

100

101

102

103104105

Order (2f1-f2)

Order (2f2-f1)

Vs=32V

PoUT=4W

RL=4

Gv=26dB

Fig.6 Large signal frequ ency response Fig.7 Maximum allowable power

dissipation vs. ambient temperture

101102103104

30

5

10

15

20

25

Vs=+-15V

RL=4

Vs=+-15V

RL=8

-50 0 50 100 150 200

30

5

10

15

20

25

infinite heatsink

heatsink having

Rth=25

°

C/W

heatsink having

Rth=4

°

C/W

heatsink having

Rth=8

°

C/W

Ta (°C)

PD (W)

Frequency (kHz)

Vo(Vp-p)

Po (W) Frequency (Hz)

Vs (V)Frequency (Hz)

PoUT (W)

d( % )

TDA2030 LINEAR INTEGRATED CIRCUIT

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UTC

TDA2030

1

23

54

Vi

+Vs

C3

0.22 F

R3

56k

RL=4

R4

3.3k

1N4001

C4

10 F

R1

56k

C1

2.2 F

R2

56k

C2

22 F

C5

220 F

/40V

C8

2200 F

R6

1.5

C6

0.22 F

R5

30k

R7

1.5

1N4001

R8

1

C7

0.22 F

Fig. 1 Single supply hig h power amplifier

 TYPICAL PERFORMANCE OF THE CIRCUIT OF FIG. 1

PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNIT

Supply Voltage Vs 36 44 V

Quiescent Drain Current IQ Vs=36V 50 mA

d=0.5%,RL=4Ω

f=40Hz to 15kHz,Vs=39V 35

d=0.5%,RL=4Ω

f=40Hz to 15kHz,Vs=36V 28

d=10%,f=1kHz,

RL=4Ω,Vs=39V 44

Output Power POUT

d=10%,RL=4Ω

f=1kHz,Vs=36V 35

W

Voltage Gain Gv f=1kHz 19.5 20 20.5 dB

Slew Rate SR 8 V/µsec

POUT=20W,f=1kHz 0.02 %

Total Harmonic Distortion d POUT=20W,f=40Hz to 15kHz 0.05 %

Input Sensitivity VIN Gv=20dB,POUT=20W,

f=1kHz,RL=4Ω 890 mV

RL=4Ω,Rg=10kΩ

B=curve A,POUT=25W

108 dB

Signal to Noise Ratio S/N RL=4Ω,Rg=10kΩ

B=curve A,POUT=4W 100

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 TYPICAL PERFORMA NCE CHAR ACTER ISTICS

24 28 32 34 36 40

5

15

25

35

45

Output Power vs. Supply Voltage

Vs (V) 10-1 100101

10-2

10-1

100

f=15kHz

f=1kHz

Vs=36V

RL=4

Gv=20dB

PoUT (W)

Total Harmonic Distorti on vs. Output Power

100 250 400 550 700

0

5

10

15

20 Gv=26dB

Gv=20dB

VIN (mV)

Output Power vs. Input Level

0

5

10

15

20

0 8 16 24 32

PoUT (W)

Complete

Amplifier

UTC

TDA2030

Power Dissipation vs. Output Power

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TYPICAL AMPLIFIER WITH SPLIT POWER SUPPLY

1

23

54

Vi

+Vs

-Vs

C1

1  F

C2

22  F C6

100  F C4

100nF C7

220nF

C3

100nF

C5

100  F D1

1N4001

D2

1N4001

R3

22k

R1

22k

R5 C8 R4

1 RL

R3

680

BRIDGE AMPLIFIER WITH SPLIT POWER SUPPLY(POUT=34W,VS=16V, VS=-16V)

UTC TDA2030

C1

2.2 F

C6

100 FC7

100nF

1

23

5

4

1

23

4

5

C8

0.22

F

C4

22 F

C9

0. 22 F

C5

22 F

C3

100nF

C2

100 F

R2

22k

R5

22k

R6

680

R9

1

R8

1

R4

680

R3

22k

R7

22k

R1

22k

Vs+

Vs-

IN

RL

8

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 MULTIWAY SPEAKER SYSTEMS AND AC TIVE BOXES

Multiway loudspeaker systems provide the best possible acoustic performance since each loudspeaker is

specially designed and optimized to handle a limited range of frequencies. Commonly, these loudspeaker systems

divide the audio spectrum two or three bands.

To maintain a flat frequency response over the Hi-Fi audio range the bands cobered by each loudspeaker must

overlap slightly. Imbalance between the loudspeakers produces unacceptable results therefore it is important to

ensure that each unit generates the correct amount of aco ustic ener gy for its s egments of the audio sp ectrum. In this

respect it is also important to know the energy distribution of the music spectrum to determine the cutoff frequencies

of the crossover filters(see Fig. 2).As an example, a 100W three-way system with crossover frequencies of 400Hz

and 3KHz would require 50W for the woofer,35W for the midrange unit an d 15W for the tweeter.

Both active and passive filters can be used for crossovers but active filters cost significantly less than a good

passive filter using aircored inductors and non-electrolytic capacitors. In addition active filters do not suffer from the

typical defects of passive filters:

--Power less;

--Increased impedance seen by the loudspeaker(lower damping)

--Difficulty of precise design due to variable loudspeaker impedance.

Obviously, active crossovers can only be used if a power amplifier is provide for each drive unit. This makes it

particularly interesting and economically sound to use monolithic power amplifiers.

In some applications complex filters are not relay necessary and simple RC low-pass and high-pass

networks(6dB/octave) can be recommended.

T he result obtained are excellent because this is the best type of audio filter and the only one free from phase and

transient distortion.

The rather poor out of ban d attenuation of single RC filters means that the loudsp eaker must operate linearly well

beyond the crossover frequency to avoid distortion.

A more effective soluti on is shown in Fig. 3.

The proposed circuit can realize combined power amplifiers and 1 2dB/octave or high-pass or low-pass filters.

In proactive, at the input pins amplifier two equal and in-phase voltages are available, as required for the active

filter operations.

The impedance at the Pin(-) is of the order of 100Ω,while that of the Pin (+) is very high, which is also what was

wanted.

The components values calcul ated for fc=900Hz using a Bessel 3rd Sa llen and Key structure are:

C1=C2=C3=22nF,R1=8.2KΩ,R2=5.6KΩ,R3=33KΩ.

Using this type of crossover filter, a complete 3-way 60W active loudspeaker system is shown in Fig. 20.

It employs 2nd order Buttherworth filter with the crossover frequencies equal to 300Hz and 3kHz.

The midrange section consistors of two filters a high pass circuit followed by a low pass network. With Vs=36V the

output power delivered to the woofer is 25W at d=0.06%( 30W at d= 0.5%).The power delivered to the m idrange and

the tweeter can be optimized in the design phase taking in account the loudspeaker efficiency and impedance

(RL=4Ω to 8Ω).

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It is quite common that midrange and tweeter speakers have an efficie ncy 3dB higher than woofers.

1

2

54

3

UTC

TDA2030

1

2

54

3

UTC

TDA2030

1

2

54

3

UTC

TDA2030

0.22 F2200 F

18nF

33nF

100 F

0.22 F

1N4001

1 F

0.1 F0.1 F

0.22 FVs+

18nF

3.3nF

100 F

0.22 F

3.3 nF3.3 nF

47 F

0.22 F

100 F

0.22 F

220 F

0.22 F

2200 F

1N4001

BD908

BD907

22k

1

4

1.5

3.3k

22k

680

100

1

22k22k

6.8k

3.3k

100

2.2k

Vs+

1N4001

8

1

2.2k

12k

100

22k

8

22k

Vs+

100 F

Vs+

IN

Woofer

Midrange

Tweeter

High-pass

3KHz

High-pass

3KHz

Band-pass

300Hz to 3KHz

Low-pass

300Hz

1N4001

 MUSICAL INSTRUMENTS AMPLIFIERS

Another important field of appl ication for active system is music.

In this area the use of several medium power amplifiers is more convenie nt than a single h igh po wer amplifier, and

it is also more reliable. A typical example (see Fig. 4) consist of four amplifiers each driving a low-cost, 12 inch

loudspeaker. This applic ation can supply 80 to 160W rms.

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 TRANSIENT INTER-MODULATION DISTORTION(TIM)

Transient inter-modulation distortion is an unfortunate phenomena associated with negative-feedback amplifiers.

When a feedback amplifier receives an input signal which rises very steeply, i.e. contains high-frequency

components, the feedback can arrive too late so that the amplifiers overloads and a burst of inter-modulation

distortion will be produced as in Fig.5. Since transients occur frequently in music this obviously a problem for the

designed of audio amplifi ers. Unfortunately, heav y negativ e feedback is frequ ency used to reduc e the total harmonic

distortion of an amplifier, which tends to aggravate the transient inter-modulation (TIM situation.)

20 to 40W

Amplifier

20 to 40W

Amplifier

20 to 40W

Amplifier

20 to 40W

Amplifier

PRE AMPLIFIER POWER

AMPLIFIER

FEEDBACK

PATH

INPUT

V1 V2 V3 V4

¦ÂV4

OUTPUT

V1

V2

V3

V4

Fig.4 High power active box for musical instrument Fig.5 Overshoot phenomenon in feedback amplifiers

The best known method for the measurement of TIM consists of feeding sine waves superimposed onto square

wavers, into the amplifier under test. The output spectrum is then examined using a spectrum analyzer and

compared to the input. This method suffers from serious disadvantages: the accurac y is limited, the measurement is

a tatter delicate operation and an expensive spectrum analyzer is ess ential.

The "inverting-sa wtooth" method of measurement is based on the response of an amplifier to a 20KHz sa w-tooth

wave-form. The amplifier has no difficulty following the slow ramp but it cannot follow the fast edge. The output will

follow the upper line in Fig.6 cutting of the shade area and thus increasing the mean level. If this output signal is

filtered to remove the saw-tooth, direct voltage remains which indicates the amount of TIM distortion, although it is

difficult to measure because it is indist inguishable from the DC offset of the amplifier. This problem is neatly avoide d

in the IS-TIM method by periodically inverting the saw-tooth wave-form at a low audio frequency as sho wn in

Fig.7. In the case of the saw-tooth in Fig. 8 the mean level was increased by the TIM distortion, for a saw-tooth in the

other direction the opposite is true.

m2

m1

SR(V/ s) Input

Signal

Filtered

Output

Siganal

Fig.6 20kHz sawtooth waveform Fig.7 Inverting sawtooth waveform

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The result is an AC signal at the output whole peak-to-peak value is the TIM voltage, which can be measured

easily with an oscilloscop e. If the peak-topeak value of the signal and the peak-to-peak of the inverting sawtooth are

measured, the TIM can be found very simply from:

TIM VOUT

Vsawtooth * 100=

TIM(%)

TIM=0.1%

TIM=0.01%

TIM=1%

SR(V/¦Ìs)

In Fig.8 The experimental results are shown for the 30W amplifier using the UTC TDA2030 as a driver and a

low-cost complementary pair. A simple RC fi lter on th e input of the amplifi er to limit the m aximum signal slope(SS) is

an effective way to reduce TIM.

The Diagram of Fig.9 can be used to find the Slew-Rate(SR) required for a given output power or voltage and a

TIM design target.

For example if an anti-TIM filter with a cutoff at 30kHz is used and the max. peak to peak output voltage is 20V

then, referring to the diagram, a Slew-Rate of 6V/µs is necessary for 0.1% TIM.

As shown Slew-Rates of above 10V/µs do not contribute to a further reduction in TIM.

Slew-Rates of 100V/µs are not only useless but also a disadvantage in hi-fi audio amplif iers because they tend to

turn the amplifier into a radio receiver.

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 POWER SUPPLY

Using monolithic audio amplifier with non regulated supply correctly. In any working case it must provide a supply

voltage less than the maximu m value fixed by the IC breakdown voltage.

It is essential to take into account all the working conditions, in particular mains fluctuations and supply voltage

variations with and without load. The UTC TDA2030 (Vsmax=44V) is particularly suitable for substitution of the

standard IC power amplifiers (with Vsmax=36V) for more reliable applications.

An example, using a simple full-wave rectifier follo wed by a capacitor filter, is shown in the table and in the diagram

of Fig.10.

A regulated supply is not usually used for the power output stages because of its dimensioning must be done

taking into account the power to supply in signal peaks. They are not only a small percentage of the total music

signal, with consequently large overdimensioning of the circuit.

Even if with a regulated supply higher output power can be obtained(Vs is constant in all working conditions),the

additional cost and power dissipation do not usually justify its use. using non-regulated supplies, there are fewer

designee restriction. In fact, when signal peaks are present, the capacitor filter acts as a flywheel supplying the

required energy.

In average c onditions, the continuous power supplied is l ower. The music power/continuous power ratio is greater

in case than for the case of regulated supplied, with space saving and cost reduction.

0 0.4 0.8 1.2 1.6 2.0

28

30

32

34

36

VOUT(V)

IOUT(A)

Fig.10 DC characteristics of 50W non-regulated supply

Vo

3300 F

220V

0

2

4

Ripple (Vp-p)

Ripple

Vout

DC Output Voltage(VOUT)

Mains(220V) Secondary Voltage IOUT =0 IOUT =0.1A IOUT =1A

+20% 28.8V 43.2V 42V 37.5V

+15% 27.6V 41.4V 40.3V 35.8V

+10% 26.4V 39.6V 38.5V 34.2V

— 24V 36.2V 35V 31V

-10% 21.6V 32.4V 31.5V 27.8V

-15% 20.4V 30.6V 29.8V 26V

-20% 19.2V 28.8V 28V 24.3V

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 SHORT CIRCUIT PROTECTION

The UTC TDA2030 has an original circuit which limits the current of the output transistors. This function can be

considered as being peak p ower limiting rather than simple current lim iting. It reduces the possibility that the dev ice

gets damaged during an accid ental short circuit from AC output to Ground.

 THERMAL SHUT- DOW N

The presence of a thermal limiting circuit offers the following advantages:

1).An overload on the output (even if it is permanent),or an above limit ambient temperature can be easily

supported since the Tj can not be high er than 150°C

2).The heatsink can have a smaller factor of safety compared with that of a congenital circuit, There is no

possibility of device damage due to high junction temperature increase up to 150°C, the thermal shut-down

simply reduces the power dissipatio n and the current consumption.

 APPLICATION SUGGESTION

The recommended values of the components are those shown on application circuit of Fig.14. Different values can

be used. The following tabl e can help the designer.

COMPONENT RECOMMENDED

VALUE PURPOSE LARGER T HAN

RECOMMENDED VALUE

SMALLER THAN

RECOMMENDED

VALUE

R1 22KΩ Closed loop gaon

setting. Increase of Gain Decrease of Gain

R2 680Ω Closed loop gaon

setting. Decrease of Gain Increase of Gain

R3 22KΩ Non inverting input

biasing Increase of input impedance Decrease of input

impedance

R4 1Ω Frequency stability

Danger of oscillation at high

frequencies with inductive

loads.

R5 ≈3R2 Upper frequency cutoff Poor high frequencies

attenuation Danger of oscillation

C1 1µF Input DC decoupling Increase of low

frequencies cutoff

C2 22µF Inverting DC

decoupling Increase of low

frequencies cutoff

C3,C4 0.1µF Supply voltage bypass Danger of oscillation

C5,C6 100µF Supply voltage bypass Danger of oscillation

C7 0.22µF Frequency stability Larger bandwidth

C8 ≈1/(2π*B*R1) Upper freque ncy cutoff smaller bandwidth Larger bandwidth

D1,D2 1N4001

To protect the device

against output voltage

spikes.

UTC assumes no responsibility for equipment failures that result from using products at values that

exceed, even momentarily, rated values (such as maximum ratings, operating condition ranges, or

other param eters) l isted in products specif icat ions of any and all UTC products descri bed or contained

herein. UTC products are not designed for use in life support appliances, devices or systems where

m al funct ion of these pr oducts can be reasonably ex pected to result i n personal i njury. Reproduction in

whole or in part is prohibited without the prior written consent of the copyright owner. The inf ormation

presented in this docum ent does not f orm part of any quotat ion or cont ract, i s bel iev ed to be accurate

and rel iable and m ay be c hanged without not ice.