LM2716
Dual (Step-Up and Step-Down) PWM DC/DC Converter
General Description
The LM2716 is composed of two PWM DC/DC converters. A
buck (step-down) converter is used to generate a fixed out-
put voltage of 3.3V. A boost (step-up) converter is used to
generate an adjustable output voltage. Both converters fea-
ture low R
DSON
(0.16Ωand 0.12Ω) internal switches for
maximum efficiency. Operating frequency can be adjusted
anywhere between 300kHz and 600kHz allowing the use of
small external components. External soft-start pins for each
enables the user to tailor the soft-start times to a specific
application. Each converter may also be shut down indepen-
dently with its own shutdown pin. The LM2716 is available in
a low profile 24-lead TSSOP package.
Features
nFixed 3.3V buck converter with a 1.8A, 0.16Ω, internal
switch
nAdjustable boost converter with a 3.6A, 0.12Ω, internal
switch
nAdjustable boost output voltage up to 20V
nOperating input voltage range of 4V to 20V
nInput undervoltage protection
n300kHz to 600kHz pin adjustable operating frequency
nOver temperature protection
nSmall 24-Lead TSSOP package
nPatented current limit circuitry
Applications
nTFT-LCD Displays
nHandheld Devices
nPortable Applications
nCellular Phones/Digital Camers
Typical Application Circuit
20071201
November 2005
LM2716 Dual (Step-Up and Step-Down) PWM DC/DC Converter
© 2005 National Semiconductor Corporation DS200712 www.national.com
Connection Diagram
Top View
20071204
24-Lead TSSOP
Ordering Information
Order Number Package Type NSC Package Drawing Supplied As
LM2716MT TSSOP-24 MTC24 61 Units, Rail
LM2716MTX TSSOP-24 MTC24 2500 Units, Tape and Reel
LM2716
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Pin Descriptions
Pin Name Function
1 PGND Power ground. AGND and PGND pins must be connected together directly at the part.
2 FB1 Buck output voltage feedback input.
3V
C1
Buck compensation network connection. Connected to the output of the voltage error
amplifier.
4V
BG
Bandgap connection.
5 SS2 Boost soft start pin.
6V
C2
Boost compensation network connection. Connected to the output of the voltage error
amplifier.
7 FB2 Boost output voltage feedback input.
8 AGND Analog ground. AGND and PGND pins must be connected together directly at the part.
9 AGND Analog ground. AGND and PGND pins must be connected together directly at the part.
10 PGND Power ground. AGND and PGND pins must be connected together directly at the part.
11 PGND Power ground. AGND and PGND pins must be connected together directly at the part.
12 PGND Power ground. AGND and PGND pins must be connected together directly at the part.
13 SW2 Boost power switch input. Switch connected between SW2 pins and PGND pins. SW2
pins should be connected directly together at the device.
14 SW2 Boost power switch input. Switch connected between SW2 pins and PGND pins. SW2
pins should be connected directly together at the device.
15 SW2 Boost power switch input. Switch connected between SW2 pins and PGND pins. SW2
pins should be connected directly together at the device.
16 V
IN
Analog power input. V
IN
pins must be connected together directly at the DUT.
17 V
IN
Analog power input. V
IN
pins must be connected together directly at the DUT.
18 SHDN2 Shutdown pin for Boost converter. Active low.
19 FSLCT Switching frequency select input. Use a resistor to set the frequency anywhere between
300kHz and 600kHz.
20 SS1 Buck soft start pin.
21 SHDN1 Shutdown pin for Buck converter. Active low.
22 CB1 Buck converter bootstrap capacitor connection.
23 V
IN
Analog power input. V
IN
pins must be connected together directly at the DUT.
24 SW1 Buck power switch input. Switch connected between V
IN
pins and SW1 pin.
LM2716
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Block Diagram
20071203
LM2716
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Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
V
IN
−0.3V to 22V
SW1 Voltage −0.3V to 22V
SW2 Voltage −0.3V to 22V
FB1 Voltage −0.3V to 7V
FB2 Voltage −0.3V to 7V
V
C1
Voltage 1.75V ≤V
C1
≤2.25V
V
C2
Voltage 0.965V ≤V
C2
≤1.565V
SHDN1 Voltage −0.3V to 7.5V
SHDN2 Voltage −0.3V to 7.5V
SS1 Voltage −0.3V to 2.1V
SS2 Voltage −0.3V to 0.6V
FSLCT Voltage AGND to 5V
CB1 Voltage V
IN
+7V(V
IN
=V
SW
)
Maximum Junction Temperature 150˚C
Power Dissipation(Note 2) Internally Limited
Lead Temperature 300˚C
Vapor Phase (60 sec.) 215˚C
Infrared (15 sec.) 220˚C
ESD Susceptibility (Note 3)
Human Body Model 2kV
Machine Model 200V
Operating Conditions
Operating Junction
Temperature Range
(Note 4) −40˚C to +125˚C
Storage Temperature −65˚C to +150˚C
Supply Voltage 4V to 20V
SW1 Voltage 20V
SW2 Voltage 20V
Electrical Characteristics
Specifications in standard type face are for T
J
= 25˚C and those with boldface type apply over the full Operating Tempera-
ture Range (T
J
= −40˚C to +125˚C) Unless otherwise specified. V
IN
= 5V and I
L
= 0A, unless otherwise specified.
Symbol Parameter Conditions Min
(Note 4)
Typ
(Note 5)
Max
(Note 4) Units
I
Q
Total Quiescent Current (both
switchers)
Not Switching 2.8 3.5 mA
Switching, switch open 4 4.5 mA
V
SHDN
=0V 915 µA
V
BG
Bandgap Voltage 1.235 1.26 1.285 V
I
CL1
(Note 6) Buck Switch Current Limit 95% Duty Cycle (Note 7) 1.8 A
I
CL2
(Note 6) Boost Switch Current Limit 95% Duty Cycle (Note 7) 3.6 A
I
FB1
Buck FB Pin Bias Current
(Note 8)
V
FB1
= 3.3V 65 75 µA
I
FB2
Boost FB Pin Bias Current
(Note 8)
V
FB2
= 1.265V 27 55 nA
V
IN
Input Voltage Range 420V
g
m1
Buck Error Amp
Transconductance
∆I = 20µA 1200 µmho
g
m2
Boost Error Amp
Transconductance
∆I = 5µA 175 µmho
A
V1
Buck Error Amp Voltage Gain 100 V/V
A
V2
Boost Error Amp Voltage
Gain 135 V/V
D
MAX
Maximum Duty Cycle 90 95 98 %
F
SW
Switching Frequency R
F
= 47.5kΩ250 300 350 kHz
R
F
= 22.6kΩ500 600 700 kHz
I
SHDN1
Buck Shutdown Pin Current 0V <V
SHDN1
<7.5V −5 5 µA
I
SHDN2
Boost Shutdown Pin Current 0V <V
SHDN2
<7.5V −5 5 µA
I
L1
Buck Switch Leakage Current V
DS
= 20V 0.2 5 µA
I
L2
Boost Switch Leakage
Current
V
DS
= 20V 0.2 3 µA
R
DSON1
Buck Switch R
DSON
160 mΩ
R
DSON2
Boost Switch R
DSON
120 mΩ
LM2716
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Electrical Characteristics (Continued)
Specifications in standard type face are for T
J
= 25˚C and those with boldface type apply over the full Operating Tempera-
ture Range (T
J
= −40˚C to +125˚C) Unless otherwise specified. V
IN
= 5V and I
L
= 0A, unless otherwise specified.
Symbol Parameter Conditions Min
(Note 4)
Typ
(Note 5)
Max
(Note 4) Units
Th
SHDN1
Buck SHDN Threshold Output High 1.37 2V
Output Low 0.8 1.35
Th
SHDN2
Boost SHDN Threshold Output High 1.37 2V
Output Low 0.8 1.35
I
SS1
Buck Soft Start Pin Current 69.5 12 µA
I
SS2
Boost Soft Start Pin Current 15 19 22 µA
UVP On Threshold 3.35 3.8 4.0 V
Off Threshold 3.10 3.6 3.9
θ
JA
Thermal Resistance
(Note 9)
TSSOP, package only 115 ˚C/W
Note 1: Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the device is intended to
be functional, but device parameter specifications may not be guaranteed. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 2: The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance, θJA,
and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance. The maximum allowable power dissipation at any ambient
temperature is calculated using: PD(MAX) = (TJ(MAX) −T
A)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the
regulator will go into thermal shutdown.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5kΩresistor into each pin. The machine model is a 200pF capacitor discharged
directly into each pin.
Note 4: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% tested
or guaranteed through statistical analysis. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods.
All limits are used to calculate Average Outgoing Quality Level (AOQL).
Note 5: Typical numbers are at 25˚C and represent the most likely norm.
Note 6: Duty cycle affects current limit due to ramp generator.
Note 7: Current limit at 95% duty cycle.
Note 8: Bias current flows into FB pin.
Note 9: Refer to National’s packaging website for more detailed thermal information and mounting techniques for the TSSOP package.
LM2716
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Typical Performance Characteristics
Switching Frequency vs. R
F
Resistor
Switching Frequency vs. Input Voltage
(F
SW
= 300kHz)
20071223
20071224
Switching Frequency vs. Input Voltage
(F
SW
= 600kHz)
Buck Efficiency vs. Load Current
(F
SW
= 300kHz)
20071225
20071226
Buck Efficiency vs. Load Current
(F
SW
= 600kHz)
Boost Efficiency vs. Load Current
(F
SW
= 300kHz)
20071227 20071231
LM2716
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Typical Performance Characteristics (Continued)
Boost Efficiency vs. Load Current
(F
SW
= 600kHz) Boost Switch R
DSON
vs. Input Voltage
20071232
20071235
LM2716
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Buck Operation
PROTECTION (BOTH REGULATORS)
The LM2716 has dedicated protection circuitry running dur-
ing normal operation to protect the IC. The Thermal Shut-
down circuitry turns off the power devices when the die
temperature reaches excessive levels. The UVP comparator
protects the power devices during supply power startup and
shutdown to prevent operation at voltages less than the
minimum input voltage. The OVP comparator is used to
prevent the output voltage from rising at no loads allowing
full PWM operation over all load conditions. The LM2716
also features a shutdown mode for each converter decreas-
ing the supply current to 9µA (both in shutdown mode).
CONTINUOUS CONDUCTION MODE
The LM2716 contains a current-mode, PWM buck regulator.
A buck regulator steps the input voltage down to a lower
output voltage. In continuous conduction mode (when the
inductor current never reaches zero at steady state), the
buck regulator operates in two cycles. The power switch is
connected between V
IN
and SW1.
In the first cycle of operation the transistor is closed and the
diode is reverse biased. Energy is collected in the inductor
and the load current is supplied by C
OUT
and the rising
current through the inductor.
During the second cycle the transistor is open and the diode
is forward biased due to the fact that the inductor current
cannot instantaneously change direction. The energy stored
in the inductor is transferred to the load and output capacitor.
The ratio of these two cycles determines the output voltage.
The output voltage is defined approximately as:
where D is the duty cycle of the switch, D and D' will be
required for design calculations.
DESIGN PROCEDURE
This section presents guidelines for selecting external com-
ponents.
INPUT CAPACITOR
A low ESR aluminum, tantalum, or ceramic capacitor is
needed betwen the input pin and power ground. This capaci-
tor prevents large voltage transients from appearing at the
input. The capacitor is selected based on the RMS current
and voltage requirements. The RMS current is given by:
The RMS current reaches its maximum (I
OUT
/2) when
V
IN
equals 2V
OUT
. This value should be increased by 50% to
account for the ripple current increase due to the boost
regulator. For an aluminum or ceramic capacitor, the voltage
rating should be at least 25% higher than the maximum input
voltage. If a tantalum capacitor is used, the voltage rating
required is about twice the maximum input voltage. The
tantalum capacitor should be surge current tested by the
manufacturer to prevent being shorted by the inrush current.
The minimum capacitor value should be 47µF for lower
output load current applications and less dynamic (quickly
changing) load conditions. For higher output current applica-
tions or dynamic load conditions a 68µF to 100µF low ESR
capacitor is recommended. It is also recommended to put a
small ceramic capacitor (0.1 µF) between the input pin and
ground pin to reduce high frequency spikes.
INDUCTOR SELECTION
The most critical parameters for the inductor are the induc-
tance, peak current and the DC resistance. The inductance
is related to the peak-to-peak inductor ripple current, the
input and the output voltages:
A higher value of ripple current reduces inductance, but
increases the conductance loss, core loss, current stress for
the inductor and switch devices. It also requires a bigger
output capacitor for the same output voltage ripple require-
ment. A reasonable value is setting the ripple current to be
30% of the DC output current. Since the ripple current in-
creases with the input voltage, the maximum input voltage is
always used to determine the inductance. The DC resistance
of the inductor is a key parameter for the efficiency. Lower
DC resistance is available with a bigger winding area. A good
tradeoff between the efficiency and the core size is letting the
inductor copper loss equal 2% of the output power.
OUTPUT CAPACITOR
The selection of C
OUT
is driven by the maximum allowable
output voltage ripple. The output ripple in the constant fre-
quency, PWM mode is approximated by:
The ESR term usually plays the dominant role in determining
the voltage ripple. A low ESR aluminum electrolytic or tanta-
lum capacitor (such as Nichicon PL series, Sanyo OS-CON,
Sprague 593D, 594D, AVX TPS, and CDE polymer alumi-
num) is recommended. An electrolytic capacitor is not rec-
ommended for temperatures below −25˚C since its ESR
rises dramatically at cold temperature. A tantalum capacitor
has a much better ESR specification at cold temperature and
is preferred for low temperature applications.
BOOT CAPACITOR
A 3.3 nF or larger ceramic capacitor is recommended for the
bootstrap capacitor.
SOFT-START CAPACITOR (BOTH REGULATORS)
The SS pins are used to tailor the soft-start for a specific
application. A current source charges the external soft-start
capacitor, C
SS
. The soft-start time can be estimated as:
T
SS
=C
SS
*0.6V/I
SS
Soft-start times may be implemented using the SS pin and a
capacitor C
SS
.
When programming the softstart time, simply use the equa-
tion given in the Soft-Start Capacitor section above. This
equation uses the typical room temperature value of the soft
start current to set the soft start time.
LM2716
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Buck Operation (Continued)
COMPENSATION COMPONENTS
In the control to output transfer function, the first pole F
P1
can be estimated as 1/(2πR
OUT
C
OUT
); The ESR zero F
Z1
of
the output capacitor is 1/(2πESRC
OUT
); Also, there is a high
frequency pole F
P2
in the range of 45kHz to 150kHz:
F
P2
=F
SW
/(πn(1−D))
whereD=V
OUT
/V
IN
, n = 1+0.348L/(V
IN
−V
OUT
)(LisinµHs
and V
IN
and V
OUT
in volts).
The total loop gain G is approximately 500/I
OUT
where I
OUT
is in amperes.
A Gm amplifier is used inside the LM2716. The output resis-
tor R
o
of the Gm amplifier is about 85kΩ.C
C1
and R
C1
together with R
o
give a lag compensation to roll off the gain:
F
PC1
= 1/(2πC
C1
(R
o
+R
C1
)), F
ZC1
= 1/2πC
C1
R
C1
.
In some applications, the ESR zero F
Z1
can not be cancelled
by F
P2
. Then, C
C3
is needed to introduce F
PC2
to cancel the
ESR zero, F
P2
= 1/(2πC
C3
R
o
\R
C1
).
The rule of thumb is to have more than 45˚ phase margin at
the crossover frequency (G=1).
SCHOTTKY DIODE
The breakdown voltage rating of D
1
is preferred to be 25%
higher than the maximum input voltage. Since D1 is only on
for a short period of time, the average current rating for D1
only requires being higher than 30% of the maximum output
current.
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Boost Operation
CONTINUOUS CONDUCTION MODE
The LM2716 contains a current-mode, PWM boost regulator.
A boost regulator steps the input voltage up to a higher
output voltage. In continuous conduction mode (when the
inductor current never reaches zero at steady state), the
boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 1 (a), the
transistor is closed and the diode is reverse biased. Energy
is collected in the inductor and the load current is supplied by
C
OUT
.
The second cycle is shown in Figure 1 (b). During this cycle,
the transistor is open and the diode is forward biased. The
energy stored in the inductor is transferred to the load and
output capacitor.
The ratio of these two cycles determines the output voltage.
The output voltage is defined approximately as:
where D is the duty cycle of the switch, D and D' will be
required for design calculations.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a
resistor divider connected to the output as shown in Figure 3.
The feedback pin voltage is 1.26V, so the ratio of the feed-
back resistors sets the output voltage according to the fol-
lowing equation:
INTRODUCTION TO COMPENSATION
20071202
FIGURE 1. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
20071205
FIGURE 2. (a) Inductor current. (b) Diode current.
LM2716
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Boost Operation (Continued)
The LM2716 has a current mode PWM boost converter. The
signal flow of this control scheme has two feedback loops,
one that senses switch current and one that senses output
voltage.
To keep a current programmed control converter stable
above duty cycles of 50%, the inductor must meet certain
criteria. The inductor, along with input and output voltage,
will determine the slope of the current through the inductor
(see Figure 2 (a)). If the slope of the inductor current is too
great, the circuit will be unstable above duty cycles of 50%.
If the duty cycle is approaching near 85% up to the maximum
of 95%, it may be necessary to increase the inductance by
as much as 2X. See Inductor and Diode Selection for more
detailed inductor sizing.
The LM2716 provides a compensation pin (V
C2
) to custom-
ize the voltage loop feedback. It is recommended that a
series combination of R
C2
and C
C2
be used for the compen-
sation network, as shown in Figure 3. For any given appli-
cation, there exists a unique combination of R
C2
and C
C2
that will optimize the performance of the LM2716 circuit in
terms of its transient response. The series combination of
R
C2
and C
C2
introduces a pole-zero pair according to the
following equations:
where R
O
is the output impedance of the error amplifier,
approximately 850kΩ. For most applications, performance
can be optimized by choosing values within the range 5kΩ≤
R
C2
≤20kΩ(R
C2
can be up to 200kΩif C
C4
is used, see
High Output Capacitor ESR Compensation) and 680pF ≤
C
C2
≤4.7nF. Refer to the Applications Information section for
recommended values for specific circuits and conditions.
Refer to the Compensation section for other design require-
ment.
COMPENSATION
This section will present a general design procedure to help
insure a stable and operational circuit. The designs in this
datasheet are optimized for particular requirements. If differ-
ent conversions are required, some of the components may
need to be changed to ensure stability. Below is a set of
general guidelines in designing a stable circuit for continu-
ous conduction operation (loads greater than approximately
100mA), in most all cases this will provide for stability during
discontinuous operation as well. The power components and
their effects will be determined first, then the compensation
components will be chosen to produce stability.
INDUCTOR AND DIODE SELECTION
Although the inductor sizes mentioned earlier are fine for
most applications, a more exact value can be calculated. To
ensure stability at duty cycles above 50%, the inductor must
have some minimum value determined by the minimum
input voltage and the maximum output voltage. This equa-
tion is:
where F
SW
is the switching frequency, D is the duty cycle,
and R
DSON
is the ON resistance of the internal switch taken
from the graph "Boost Switch R
DSON
vs. Input Voltage" in the
Typical Performance Characteristics section. This equation
is only good for duty cycles greater than 50% (D>0.5), for
duty cycles less than 50% the recommended values may be
used. The corresponding inductor current ripple as shown in
Figure 2 (a) is given by:
The inductor ripple current is important for a few reasons.
One reason is because the peak switch current will be the
average inductor current (input current or I
LOAD
/D’) plus ∆i
L
.
As a side note, discontinuous operation occurs when the
inductor current falls to zero during a switching cycle, or ∆i
L
is greater than the average inductor current. Therefore, con-
tinuous conduction mode occurs when ∆i
L
is less than the
average inductor current. Care must be taken to make sure
that the switch will not reach its current limit during normal
operation. The inductor must also be sized accordingly. It
should have a saturation current rating higher than the peak
inductor current expected. The output voltage ripple is also
affected by the total ripple current.
The output diode for a boost regulator must be chosen
correctly depending on the output voltage and the output
current. The typical current waveform for the diode in con-
tinuous conduction mode is shown in Figure 2 (b). The diode
must be rated for a reverse voltage equal to or greater than
the output voltage used. The average current rating must be
greater than the maximum load current expected, and the
peak current rating must be greater than the peak inductor
current. During short circuit testing, or if short circuit condi-
tions are possible in the application, the diode current rating
must exceed the switch current limit. Using Schottky diodes
with lower forward voltage drop will decrease power dissipa-
tion and increase efficiency.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete
feedback loop with the power components, it forms a closed-
loop system that must be stabilized to avoid positive feed-
back and instability. A value for open-loop DC gain will be
required, from which you can calculate, or place, poles and
zeros to determine the crossover frequency and the phase
margin. A high phase margin (greater than 45˚) is desired for
the best stability and transient response. For the purpose of
stabilizing the LM2716, choosing a crossover point well be-
low where the right half plane zero is located will ensure
sufficient phase margin. A discussion of the right half plane
zero and checking the crossover using the DC gain will
follow.
OUTPUT CAPACITOR SELECTION
The choice of output capacitors is somewhat arbitrary and
depends on the design requirements for output voltage
ripple. It is recommended that low ESR (Equivalent Series
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Boost Operation (Continued)
Resistance, denoted R
ESR
) capacitors be used such as
ceramic, polymer electrolytic, or low ESR tantalum. Higher
ESR capacitors may be used but will require more compen-
sation which will be explained later on in the section. The
ESR is also important because it determines the peak to
peak output voltage ripple according to the approximate
equation:
∆V
OUT
)2∆i
L
R
ESR
(in Volts)
A minimum value of 10µF is recommended and may be
increased to a larger value. After choosing the output capaci-
tor you can determine a pole-zero pair introduced into the
control loop by the following equations:
Where R
L
is the minimum load resistance corresponding to
the maximum load current. The zero created by the ESR of
the output capacitor is generally very high frequency if the
ESR is small. If low ESR capacitors are used it can be
neglected. If higher ESR capacitors are used see the High
Output Capacitor ESR Compensation section.
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right
half plane zero (RHP zero). This zero has the effect of a zero
in the gain plot, causing an imposed +20dB/decade on the
rolloff, but has the effect of a pole in the phase, subtracting
another 90˚ in the phase plot. This can cause undesirable
effects if the control loop is influenced by this zero. To ensure
the RHP zero does not cause instability issues, the control
loop should be designed to have a bandwidth of less than
1
⁄
2
the frequency of the RHP zero. This zero occurs at a fre-
quency of:
where I
LOAD
is the maximum load current and D’ corre-
sponds to the minimum input voltage.
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components
R
C2
and C
C2
is to set a dominant low frequency pole in the
control loop. Simply choose values for R
C2
and C
C2
within
the ranges given in the Introduction to Compensation section
to set this pole in the area of 10Hz to 500Hz. The frequency
of the pole created is determined by the equation:
where R
O
is the output impedance of the error amplifier,
approximately 850kΩ. Since R
C2
is generally much less than
R
O
, it does not have much effect on the above equation and
can be neglected until a value is chosen to set the zero f
ZC
.
f
ZC
is created to cancel out the pole created by the output
capacitor, f
P1
. The output capacitor pole will shift with differ-
ent load currents as shown by the equation, so setting the
zero is not exact. Determine the range of f
P1
over the ex-
pected loads and then set the zero f
ZC
to a point approxi-
mately in the middle. The frequency of this zero is deter-
mined by:
Now R
C2
can be chosen with the selected value for C
C2
.
Check to make sure that the pole f
PC
is still in the 10Hz to
500Hz range, change each value slightly if needed to ensure
both component values are in the recommended range. After
checking the design at the end of this section, these values
can be changed a little more to optimize performance if
desired. This is best done in the lab on a bench, checking the
load step response with different values until the ringing and
overshoot on the output voltage at the edge of the load steps
is minimal. This should produce a stable, high performance
circuit. For improved transient response, higher values of
R
C2
should be chosen. This will improve the overall band-
width which makes the regulator respond more quickly to
transients. If more detail is required, or the most optimal
performance is desired, refer to a more in depth discussion
of compensating current mode DC/DC switching regulators.
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or
just to improve the overall phase margin of the control loop,
another pole may be introduced to cancel the zero created
by the ESR. This is accomplished by adding another capaci-
tor, C
C4
, directly from the compensation pin V
C2
to ground, in
parallel with the series combination of R
C2
and C
C2
. The
pole should be placed at the same frequency as f
Z1
, the ESR
zero. The equation for this pole follows:
To ensure this equation is valid, and that C
C4
can be used
without negatively impacting the effects of R
C2
and C
C2
,f
PC4
must be greater than 10f
ZC
.
CHECKING THE DESIGN
The final step is to check the design. This is to ensure a
bandwidth of
1
⁄
2
or less of the frequency of the RHP zero.
This is done by calculating the open-loop DC gain, A
DC
. After
this value is known, you can calculate the crossover visually
by placing a −20dB/decade slope at each pole, and a +20dB/
decade slope for each zero. The point at which the gain plot
crosses unity gain, or 0dB, is the crossover frequency. If the
crossover frequency is less than
1
⁄
2
the RHP zero, the phase
margin should be high enough for stability. The phase mar-
gin can also be improved by adding C
C4
as discussed earlier
in the section. The equation for A
DC
is given below with
additional equations required for the calculation:
LM2716
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Boost Operation (Continued)
mc )0.072F
SW
(in V/s)
where R
L
is the minimum load resistance, V
IN
is the mini-
mum input voltage, g
m
is the error amplifier transconduc-
tance found in the Electrical Characteristics table, and R
D-
SON
is the value chosen from the graph "R
DSON2
vs. V
IN
"in
the Typical Performance Characteristics section.
LAYOUT CONSIDERATIONS
The LM2716 uses two separate ground connections, PGND
for the drivers and boost NMOS power device and AGND for
the sensitive analog control circuitry. The AGND and PGND
pins should be tied directly together at the package. The
feedback and compensation networks should be connected
directly to a dedicated analog ground plane and this ground
plane must connect to the AGND pin. If no analog ground
plane is available then the ground connections of the feed-
back and compensation networks must tie directly to the
AGND pin. Connecting these networks to the PGND can
inject noise into the system and effect performance.
The input bypass capacitor C
IN
, as shown in Figure 3, must
be placed close to the IC. This will reduce copper trace
resistance which effects input voltage ripple of the IC. For
additional input voltage filtering, a 100nF bypass capacitor
can be placed in parallel with C
IN
, close to the V
IN
pin, to
shunt any high frequency noise to ground. The output ca-
pacitors, C
OUT1
and C
OUT2
, should also be placed close to
the IC. Any copper trace connections for the C
OUTX
capaci-
tors can increase the series resistance, which directly effects
output voltage ripple. The feedback network, resistors R
FB1
and R
FB2
, should be kept close to the FB pin, and away from
the inductor, to minimize copper trace connections that can
inject noise into the system. Trace connections made to the
inductors and schottky diodes should be minimized to re-
duce power dissipation and increase overall efficiency. See
Figure 3,Figure 4, and Figure 5 for a good example of
proper layout. For more detail on switching power supply
layout considerations see Application Note AN-1149: Layout
Guidelines for Switching Power Supplies.
Application Information
Some Recommended Inductors (others may be used)
Manufacturer Inductor Contact Information
Coilcraft DO3316 and DO5022 series www.coilcraft.com
Coiltronics DRQ73 and CD1 series www.cooperet.com
Pulse P0751 and P0762 series www.pulseeng.com
Sumida CDRH8D28 and CDRH8D43 series www.sumida.com
Some Recommended Input and Output Capacitors (others may be used)
Manufacturer Capacitor Contact Information
Vishay Sprague 293D, 592D, and 595D series tantalum www.vishay.com
Taiyo Yuden High capacitance MLCC ceramic www.t-yuden.com
Cornell Dubilier ESRD seriec Polymer Aluminum Electrolytic
SPV and AFK series V-chip series www.cde.com
Panasonic High capacitance MLCC ceramic
EEJ-L series tantalum www.panasonic.com
LM2716
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Application Information (Continued)
20071257
FIGURE 3. 15V, 3.3V Output Application
LM2716
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Application Information (Continued)
20071258
FIGURE 4. PCB Layout, Top
LM2716
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Application Information (Continued)
20071259
FIGURE 5. PCB Layout, Bottom
LM2716
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Physical Dimensions inches (millimeters) unless otherwise noted
TSSOP-24 Pin Package (MTC)
For Ordering, Refer to Ordering Information Table
NS Package Number MTC24
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body, or
(b) support or sustain life, and whose failure to perform when
properly used in accordance with instructions for use
provided in the labeling, can be reasonably expected to result
in a significant injury to the user.
2. A critical component is any component of a life support
device or system whose failure to perform can be reasonably
expected to cause the failure of the life support device or
system, or to affect its safety or effectiveness.
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LM2716 Dual (Step-Up and Step-Down) PWM DC/DC Converter
