LM4960SQX - National Semiconductor

LM4960

Piezoelectric Speaker Driver

General Description

The LM4960 utilizes a switching regulator to drive a dual audio

power amplifier. It delivers 24VP-P mono-BTL to a ceramic

speaker with less than 1.0% THD+N while operating on a 3.0V

power supply.

The LM4960's switching regulator is a current-mode boost

converter operating at a fixed frequency of 1.6MHz.

Boomer audio power amplifiers were designed specifically to

provide high quality output power with a minimal amount of

external components. The LM4960 does not require output

coupling capacitors or bootstrap capacitors, and therefore is

ideally suited for mobile phone and other low voltage appli-

cations where minimal power consumption is a primary re-

quirement.

The LM4960 features a low-power consumption externally

controlled micropower shutdown mode. Additionally, the

LM4960 features and internal thermal shutdown protection

mechanism along with a short circuit protection.

The LM4960 is unity-gain stable and can be configured by

external gain-setting resistors.

Key Specifications

●VOUT @ VDD = 3.0 THD+N ≤ 1% 24VP-P (typ)

●Power supply range 3.0 to 7V

●Switching Frequency 1.6MHz (typ)

Features

●Low current shutdown mode

●"Click and pop" suppression circuitry

●Low Quiescent current

●Unity-gain stable audio amplifiers

●External gain configuration capability

●Thermal shutdown protection circuitry

●Wide input voltage range (3.0V - 7V)

●1.6MHz switching frequency

Applications

●Mobile phone

●PDA's

Connection Diagram

LM4960SQ

20076582

Top View

Order Number LM4960SQ

See NS Package Number

Boomer® is a registered trademark of National Semiconductor Corporation.

PRODUCTION DATA information is current as of

publication date. Products conform to specifications per

the terms of the Texas Instruments standard warranty.

Production processing does not necessarily include

testing of all parameters.

Typical Application

20076581

FIGURE 1. Typical Audio Amplifier Application Circuit

LM4960

Absolute Maximum Ratings (Note 1, Note 2)

If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for

availability and specifications.

Supply Voltage (VDD)8.5V

Supply Voltage (V1)

(Pin 27 referred to GND) 18V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJA (LLP) °C/W

See AN-1187 'Leadless Leadframe Packaging (LLP).'

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 3.0V ≤ VDD ≤ 7V

Supply Voltage (V1) 9.6V ≤ V1 ≤ 16V

Electrical Characteristics VDD = 3.0V (Note 1, Note 2)

The following specifications apply for VDD = 3V, AV = 10, RL = 800nF+20Ω, V1 = 12V unless otherwise specified. Limits apply for

TA = 25°C.

Symbol Parameter Conditions

LM4960

Units

(Limits)

Typical

(Note 6)

Limit

(Note 7,

Note 8)

IDD Quiescent Power Supply Current VIN = GND, No Load 85 150 mA (max)

ISD Shutdown Current VSHUTDOWN = GND (Note 9) 30 100 µA (max)

VOS Output Offset Voltage 5 40 mV (max)

VSDIH Shutdown Voltage Input High 2 V (max)

VSDIL Shutdown Voltage Input Low 0.4 V (min)

TWU Wake-up Time CB = 0.22µF 50 ms

TSD Thermal Shutdown Temperature 170 150

190

°C (min)

°C (max)

VOOutput Voltage THD = 1% (max); f = 1kHz 24 20 VP-P (min)

THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz 0.04 %

εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV

PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 55 50 dB (min)

VFB Feedback Pin Reference Voltage 1.23 V (max)

Electrical Characteristics VDD = 5.0V (Note 1, Note 2)

The following specifications apply for VDD = 5V, AV = 10, RL = 800nF+20Ω unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions LM4960 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7,

Note 8)

IDD Quiescent Power Supply Current VIN = GND, No Load 45 mA (max)

ISD Shutdown Current VSHUTDOWN = GND (Note 9) 55 100 µA (max)

VSDIH Shutdown Voltage Input High 2 V (max)

LM4960

Symbol Parameter Conditions LM4960 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7,

Note 8)

VSDIL Shutdown Voltage Input Low 0.4 V (min)

TWU Wake-up Time CB = 0.22µF 50 ms

TSD Thermal Shutdown Temperature 170 150

190

°C (min)

°C (max)

VOOutput Voltage THD = 1% (max); f = 1kHz

RL = Ceramic Speaker 24 20 VP-P (min)

THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz 0.04 %

εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV

PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 60 dB (min)

VFB Feedback Pin Reference Voltage 1.23 V (max)

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions

which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters

where no limit is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4960 typical application (shown

in Figure 1) with VDD = 12V, RL = 4Ω stereo operation the total power dissipation is 3.65W. θJA = 35°C/W.

Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25°C and represent the parametric norm.

Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for minimum shutdown

current.

LM4960

Typical Performance Characteristics

THD+N vs Frequency

VDD = 3V, V1 = 9.6V, V0 = 3Vrms

20076514

THD+N vs Frequency

VDD = 3V, V1 = 12V, V0 = 3Vrms

20076515

THD+N vs Frequency

VDD = 3V, V1 = 15V, V0 = 3Vrms

20076516

THD+N vs Frequency

VDD = 5V, V1 = 9.6V, V0 = 3Vrms

20076517

THD+N vs Frequency

VDD = 5V, V1 =12V, V0 = 3Vrms

20076518

THD+N vs Frequency

VDD = 5V, V1 =15V, V0 = 3Vrms

20076519

LM4960

THD+N vs Output Power

VDD = 3V, V1 = 9.6V,

f = 100Hz, 1kHz, 10kHz

20076520

THD+N vs Output Power

VDD = 3V, V1 = 12V,

f = 100Hz, 1kHz, 10kHz

20076521

THD+N vs Output Power

VDD = 3V, V1 = 15V,

f = 100Hz, 1kHz, 10kHz

20076522

THD+N vs Output Power

VDD = 5V, V1 = 9.6V,

f = 100Hz, 1kHz, 10kHz

20076523

THD+N vs Output Power

VDD = 5V, V1 = 12V,

f = 100Hz, 1kHz, 10kHz

20076524

THD+N vs Output Power

VDD = 5V, V1 = 15V,

f = 100Hz, 1kHz, 10kHz

20076525

LM4960

Power Dissipation vs Output Voltage

VDD = 3V, from top to bottom:

V1 = 15V, V1 = 12V, V1 = 9.6V

20076509

Power Dissipation vs Output Voltage

VDD = 5V, from top to bottom:

V1 = 15V, V1 = 12V, V1 = 9.6V

20076510

Supply Current vs Supply Voltage

from top to bottom:

VDD = 15V, VDD = 12V, VDD = 9.6V

20076513

Power Supply Rejection Ratio

VDD = 3V

20076511

Power Supply Rejection Ratio

VDD = 5V

20076512

LM4960

Application Information

BRIDGE CONFIGURATION EXPLANATION

The Audio Amplifier portion of the LM4960 has two internal amplifiers allowing different amplifier configurations. The first amplifier’s

gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closed-

loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier’s gain is fixed by the two internal

20kΩ resistors. Figure 1 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers

producing signals identical in magnitude, but out of phase by 180°. Consequently, the differential gain for the Audio Amplifier is

AVD = 2 *(Rf/Ri)

By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as “bridged mode”

is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load

is connected to ground.

A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive to the load,

thus doubling the output swing for a specified supply voltage. Four times the output power is possible as compared to a single-

ended amplifier under the same conditions. This increase in attainable output power assumes that the amplifier is not current limited

or clipped. In order to choose an amplifier’s closed-loop gain without causing excessive clipping, please refer to the Audio Power

Amplifier Design section.

The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential outputs, Vo1 and

Vo2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the need for an output coupling capacitor

which is required in a single supply, single-ended amplifier configuration. Without an output coupling capacitor, the half-supply bias

across the load would result in both increased internal IC power dissipation and also possible loudspeaker damage.

AMPLIFIER POWER DISSIPATION

Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A

direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation.

Since the amplifier portion of the LM4960 has two operational amplifiers, the maximum internal power dissipation is 4 times that

of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation 1.

PDMAX(AMP) = 4(VDD)2 / (2π2ZL) (1)

where

ZL = Ro1 + Ro2 +1/2πfc

BOOST CONVERTER POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power

dissipation within the LM2731 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation 2.

PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (2)

where DC is the duty cycle.

There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation.

TOTAL POWER DISSIPATION

The total power dissipation for the LM4960 can be calculated by adding Equation 1 and Equation 2 together to establish Equation

PDMAX(TOTAL) = [4*(VDD)2/2π2ZL] + [DC x IIND(AVE)2 xRDS(ON)] (3)

The result from Equation 3 must not be greater than the power dissipation that results from Equation 4:

PDMAX = (TJMAX - TA) / θJA (4)

For the LQA28A, θJA = 59°C/W. TJMAX = 125°C for the LM4960. Depending on the ambient temperature, TA, of the system sur-

roundings, Equation 4 can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of

Equation 3 is greater than that of Equation 4, then either the supply voltage must be increased, the load impedance increased or

TA reduced. For the typical application of a 3V power supply, with V1 set to 12V and a 800nF + 20Ω load, the maximum ambient

temperature possible without violating the maximum junction temperature is approximately 118°C provided that device operation

LM4960

is around the maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power dissipation

is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient tem-

perature may be increased accordingly. Refer to the Typical Performance Characteristics curves for power dissipation information

for lower output levels.

EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS

The LM4960’s exposed-DAP (die attach paddle) package (LD) provides a low thermal resistance between the die and the PCB to

which the part is mounted and soldered. The low thermal resistance allows rapid heat transfer from the die to the surrounding PCB

copper traces, ground plane, and surrounding air. The LD package should have its DAP soldered to a copper pad on the PCB. The

DAP’s PCB copper pad may be connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat

sink, and radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting an LD (LLP)

package is found in National Semiconductor’s Package Engineering Group under application note AN1187.

SHUTDOWN FUNCTION

In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth

transition into shutdown. Another solution is to use a single-pole, single-throw switch, and a pull-up resistor. One terminal of the

switch is connected to GND. The other side is connected to the two shutdown pins and the terminal of the pull-up resistor. The

remaining resistance terminal is connected to VDD. If the switch is open, then the external pull-up resistor connected to VDD will

enable the LM4960. This scheme guarantees that the shutdown pins will not float thus preventing unwanted state changes.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC converters, is

critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system

quality.

The best capacitors for use with the switching converter portion of the LM4960 are multi-layer ceramic capacitors. They have the

lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency

switching converters.

When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U and Y5F have

such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of

rated capacitance in many typical applications. Always consult capacitor manufacturer’s data curves before selecting a capacitor.

High-quality ceramic capacitors can be obtained from Taiyo-Yuden, AVX, and Murata.

POWER SUPPLY BYPASSING

As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor

location on both V1 and VDD pins should be as close to the device as possible.

SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER

One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the

choice of external components shown in Figure 1. The input coupling capacitor, Ci, forms a first order high pass filter which limits

low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons.

High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed

to couple in low frequencies without severe attenuation. But ceramic speakers used in portable systems, whether internal or ex-

ternal, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase

actual system performance.

In addition to system cost and size, click and pop performance is affected by the value of the input coupling capacitor, Ci. A high

value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes

from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on

desired low frequency response, turn-on pops can be minimized.

SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER

Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value. Bypass capacitor,

CB, is the most critical component to minimize turn-on pops since it determines how fast the amplifer turns on. The slower the

amplifier’s outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the smaller the turn-on pop. Choosing CB equal to 1.0µF

along with a small value of Ci (in the range of 0.039µF to 0.39µF), should produce a virtually clickless and popless shutdown function.

Although the device will function properly, (no oscillations or motorboating), with CB equal to 0.1µF, the device will be much more

susceptible to turn-on clicks and pops. Thus, a value of CB equal to 1.0µF is recommended in all but the most cost sensitive designs.

SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER

The LM4960 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic speakers, a

typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) will be needed as shown in

Figure 2 to bandwidth limit the amplifier.

This feedback capacitor creates a low pass filter that eliminates possible high frequency oscillations. Care should be taken when

calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency

LM4960

SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER

A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of

capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics

with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors

are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating

from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability.

In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching

losses, and output voltage ripple.

SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER

An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns

ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7µF, but

larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the

amount of EMI passed back along that line to other circuitry.

SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER

The output voltage is set using the external resistors R1 and R2 (see Figure 1). A value of approximately 13.3kΩ is recommended

for R2 to establish a divider current of approximately 92µA. R1 is calculated using the formula:

R1 = R2 X (V2/1.23 − 1) (5)

FEED-FORWARD COMPENSATION FOR BOOST CONVERTER

Although the LM4960's internal Boost converter is internally compensated, the external feed-forward capacitor Cf is required for

stability (see Figure 1). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for

the zero fz should be approximately 6kHz. Cf1 can be calculated using the formula:

Cf1 = 1 / (2 X R1 X fz) (6)

SELECTING DIODES

The external diode used in Figure 1 should be a Schottky diode. A 20V diode such as the MBR0520 is recommended.

The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A

average but less than 1A, a Microsemi UPS5817 can be used.

DUTY CYCLE

The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter

can attain in continuous mode of operation. The duty cycle for a given boost application is defined as:

Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW

This applies for continuous mode operation.

INDUCTANCE VALUE

The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest sized component

and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor

ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean

more load power can be delivered because the energy stored during each switching cycle is:

E = L/2 X (lp)2

LM4960

Where “lp” is the peak inductor current. An important point to observe is that the LM4960 will limit its switch current based on peak

current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.

Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.

Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest,

typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the inductor current

to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load

is reduced far enough, but a larger inductor stays “continuous” over a wider load current range.

To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10µH inductor) will be analyzed. We will

assume:

VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V

Since the frequency is 1.6MHz (nominal), the period is approximately 0.625µs. The duty cycle will be 62.5%, which means the ON-

time of the switch is 0.390µs. It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V.

Using the equation:

V = L (di/dt)

We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON-time. Using these facts, we can

then show what the inductor current will look like during operation:

20076583

FIGURE 2. 10μH Inductor Current

5V - 12V Boost (LM4960)

During the 0.390µs ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFF-time. This

is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33mA, the inductor current will

begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost

converter, to make sure the ripple current is reasonable and continuous operation will be maintained at the typical load current

values.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is

illustrated in a graph in the typical performance characterization section which shows typical values of switch current as a function

of effective (actual) duty cycle.

CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP)

As shown in Figure 2 which depicts inductor current, the load current is related to the average inductor current by the relation:

ILOAD = IIND(AVG) x (1 - DC) (7)

Where "DC" is the duty cycle of the application. The switch current can be found by:

ISW = IIND(AVG) + 1/2 (IRIPPLE) (8)

Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:

IRIPPLE = DC x (VIN-VSW) / (f x L) (9)

LM4960

Combining all terms, we can develop an expression which allows the maximum available load current to be calculated:

ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (10)

The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses

of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET "ON" voltage (referred to as VSW in equations 7 thru 10) is dependent on load current. A good approximation

can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current.

FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see

Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V.

The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical Perfor-

mance Characteristics curves.

INDUCTOR SUPPLIERS

Recommended suppliers of inductors for the LM4960 include, but are not limited to Taiyo-Yuden, Sumida, Coilcraft, Panasonic,

TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation

at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered

when selecting the current rating.

PCB LAYOUT GUIDELINES

High frequency boost converters require very careful layout of components in order to get stable operation and low noise. All

components must be as close as possible to the LM4802 device. It is recommended that a 4-layer PCB be used so that internal

ground planes are available.

Some additional guidelines to be observed:

1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co will increase noise

and ringing.

2. The feedback components R1, R2 and Cf 1 must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace.

3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative

sides of capacitors Cs1 and Co.

GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION

This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power and ground traces.

Designers should note that these are only "rule-of-thumb" recommendations and the actual results will depend heavily on the final

layout.

Power and Ground Circuits

For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the analog power and ground

trace paths. Star trace routing techniques (bringing individual traces back to a central point rather than daisy chaining traces together

in a serial manner) can have a major impact on low level signal performance. Star trace routing refers to using individual traces to

feed power and ground to each circuit or even device. This technique will take require a greater amount of design time but will not

increase the final price of the board. The only extra parts required may be some jumpers.

Single-Point Power / Ground Connection

The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can be helpful in

minimizing high frequency noise coupling between the analog and digital sections. It is further recommended to place digital and

analog power traces over the corresponding digital and analog ground traces to minimize noise coupling.

Placement of Digital and Analog Components

All digital components and high-speed digital signals traces should be located as far away as possible from analog components

and circuit traces.

Avoiding Typical Design / Layout Problems

Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB layer. When traces

must cross over each other do it at 90 degrees. Running digital and analog traces at 90 degrees to each other from the top to the

bottom side as much as possible will minimize capacitive noise coupling and crosstalk.

LM4960

Physical Dimensions inches (millimeters) unless otherwise noted

LLP, Plastic, Quad

Order Number LM4960SQ

NS Package Number SQA28A

LM4960

Notes

Incorporated

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