LM4960
Piezoelectric Speaker Driver
General Description
The LM4960 utilizes a switching regulator to drive a dual audio
power amplifier. It delivers 24VP-P mono-BTL to a ceramic
speaker with less than 1.0% THD+N while operating on a 3.0V
power supply.
The LM4960's switching regulator is a current-mode boost
converter operating at a fixed frequency of 1.6MHz.
Boomer audio power amplifiers were designed specifically to
provide high quality output power with a minimal amount of
external components. The LM4960 does not require output
coupling capacitors or bootstrap capacitors, and therefore is
ideally suited for mobile phone and other low voltage appli-
cations where minimal power consumption is a primary re-
quirement.
The LM4960 features a low-power consumption externally
controlled micropower shutdown mode. Additionally, the
LM4960 features and internal thermal shutdown protection
mechanism along with a short circuit protection.
The LM4960 is unity-gain stable and can be configured by
external gain-setting resistors.
Key Specifications
VOUT @ VDD = 3.0 THD+N 1% 24VP-P (typ)
Power supply range 3.0 to 7V
Switching Frequency 1.6MHz (typ)
Features
Low current shutdown mode
"Click and pop" suppression circuitry
Low Quiescent current
Unity-gain stable audio amplifiers
External gain configuration capability
Thermal shutdown protection circuitry
Wide input voltage range (3.0V - 7V)
1.6MHz switching frequency
Applications
Mobile phone
PDA's
Connection Diagram
LM4960SQ
20076582
Top View
Order Number LM4960SQ
See NS Package Number
Boomer® is a registered trademark of National Semiconductor Corporation.
PRODUCTION DATA information is current as of
publication date. Products conform to specifications per
the terms of the Texas Instruments standard warranty.
Production processing does not necessarily include
testing of all parameters.
200765 SNAS221B Copyright © 1999-2012, Texas Instruments Incorporated
Typical Application
20076581
FIGURE 1. Typical Audio Amplifier Application Circuit
LM4960
2 Copyright © 1999-2012, Texas Instruments Incorporated
Absolute Maximum Ratings (Note 1, Note 2)
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for
availability and specifications.
Supply Voltage (VDD)8.5V
Supply Voltage (V1)
(Pin 27 referred to GND) 18V
Storage Temperature −65°C to +150°C
Input Voltage −0.3V to VDD + 0.3V
Power Dissipation (Note 3) Internally limited
ESD Susceptibility (Note 4) 2000V
ESD Susceptibility (Note 5) 200V
Junction Temperature 150°C
Thermal Resistance
 θJA (LLP) °C/W
See AN-1187 'Leadless Leadframe Packaging (LLP).'
Operating Ratings
Temperature Range
TMIN TA TMAX −40°C TA +85°C
Supply Voltage (VDD) 3.0V VDD 7V
Supply Voltage (V1) 9.6V V1 16V
Electrical Characteristics VDD = 3.0V (Note 1, Note 2)
The following specifications apply for VDD = 3V, AV = 10, RL = 800nF+20, V1 = 12V unless otherwise specified. Limits apply for
TA = 25°C.
Symbol Parameter Conditions
LM4960
Units
(Limits)
Typical
(Note 6)
Limit
(Note 7,
Note 8)
IDD Quiescent Power Supply Current VIN = GND, No Load 85 150 mA (max)
ISD Shutdown Current VSHUTDOWN = GND (Note 9) 30 100 µA (max)
VOS Output Offset Voltage 5 40 mV (max)
VSDIH Shutdown Voltage Input High 2 V (max)
VSDIL Shutdown Voltage Input Low 0.4 V (min)
TWU Wake-up Time CB = 0.22µF 50 ms
TSD Thermal Shutdown Temperature 170 150
190
°C (min)
°C (max)
VOOutput Voltage THD = 1% (max); f = 1kHz 24 20 VP-P (min)
THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz 0.04 %
εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV
PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 55 50 dB (min)
VFB Feedback Pin Reference Voltage 1.23 V (max)
Electrical Characteristics VDD = 5.0V (Note 1, Note 2)
The following specifications apply for VDD = 5V, AV = 10, RL = 800nF+20 unless otherwise specified. Limits apply for TA = 25°C.
Symbol Parameter Conditions LM4960 Units
(Limits)
Typical
(Note 6)
Limit
(Note 7,
Note 8)
IDD Quiescent Power Supply Current VIN = GND, No Load 45 mA (max)
ISD Shutdown Current VSHUTDOWN = GND (Note 9) 55 100 µA (max)
VSDIH Shutdown Voltage Input High 2 V (max)
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 3
Symbol Parameter Conditions LM4960 Units
(Limits)
Typical
(Note 6)
Limit
(Note 7,
Note 8)
VSDIL Shutdown Voltage Input Low 0.4 V (min)
TWU Wake-up Time CB = 0.22µF 50 ms
TSD Thermal Shutdown Temperature 170 150
190
°C (min)
°C (max)
VOOutput Voltage THD = 1% (max); f = 1kHz
RL = Ceramic Speaker 24 20 VP-P (min)
THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz 0.04 %
εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV
PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 60 dB (min)
VFB Feedback Pin Reference Voltage 1.23 V (max)
Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.
Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions
which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters
where no limit is given, however, the typical value is a good indication of device performance.
Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum
allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4960 typical application (shown
in Figure 1) with VDD = 12V, RL = 4Ω stereo operation the total power dissipation is 3.65W. θJA = 35°C/W.
Note 4: Human body model, 100pF discharged through a 1.5k resistor.
Note 5: Machine Model, 220pF–240pF discharged through all pins.
Note 6: Typicals are measured at 25°C and represent the parametric norm.
Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).
Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for minimum shutdown
current.
LM4960
4 Copyright © 1999-2012, Texas Instruments Incorporated
Typical Performance Characteristics
THD+N vs Frequency
VDD = 3V, V1 = 9.6V, V0 = 3Vrms
20076514
THD+N vs Frequency
VDD = 3V, V1 = 12V, V0 = 3Vrms
20076515
THD+N vs Frequency
VDD = 3V, V1 = 15V, V0 = 3Vrms
20076516
THD+N vs Frequency
VDD = 5V, V1 = 9.6V, V0 = 3Vrms
20076517
THD+N vs Frequency
VDD = 5V, V1 =12V, V0 = 3Vrms
20076518
THD+N vs Frequency
VDD = 5V, V1 =15V, V0 = 3Vrms
20076519
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 5
THD+N vs Output Power
VDD = 3V, V1 = 9.6V,
f = 100Hz, 1kHz, 10kHz
20076520
THD+N vs Output Power
VDD = 3V, V1 = 12V,
f = 100Hz, 1kHz, 10kHz
20076521
THD+N vs Output Power
VDD = 3V, V1 = 15V,
f = 100Hz, 1kHz, 10kHz
20076522
THD+N vs Output Power
VDD = 5V, V1 = 9.6V,
f = 100Hz, 1kHz, 10kHz
20076523
THD+N vs Output Power
VDD = 5V, V1 = 12V,
f = 100Hz, 1kHz, 10kHz
20076524
THD+N vs Output Power
VDD = 5V, V1 = 15V,
f = 100Hz, 1kHz, 10kHz
20076525
LM4960
6 Copyright © 1999-2012, Texas Instruments Incorporated
Power Dissipation vs Output Voltage
VDD = 3V, from top to bottom:
V1 = 15V, V1 = 12V, V1 = 9.6V
20076509
Power Dissipation vs Output Voltage
VDD = 5V, from top to bottom:
V1 = 15V, V1 = 12V, V1 = 9.6V
20076510
Supply Current vs Supply Voltage
from top to bottom:
VDD = 15V, VDD = 12V, VDD = 9.6V
20076513
Power Supply Rejection Ratio
VDD = 3V
20076511
Power Supply Rejection Ratio
VDD = 5V
20076512
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 7
Application Information
BRIDGE CONFIGURATION EXPLANATION
The Audio Amplifier portion of the LM4960 has two internal amplifiers allowing different amplifier configurations. The first amplifier’s
gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closed-
loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier’s gain is fixed by the two internal
20k resistors. Figure 1 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers
producing signals identical in magnitude, but out of phase by 180°. Consequently, the differential gain for the Audio Amplifier is
AVD = 2 *(Rf/Ri)
By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as “bridged mode”
is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load
is connected to ground.
A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive to the load,
thus doubling the output swing for a specified supply voltage. Four times the output power is possible as compared to a single-
ended amplifier under the same conditions. This increase in attainable output power assumes that the amplifier is not current limited
or clipped. In order to choose an amplifier’s closed-loop gain without causing excessive clipping, please refer to the Audio Power
Amplifier Design section.
The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential outputs, Vo1 and
Vo2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the need for an output coupling capacitor
which is required in a single supply, single-ended amplifier configuration. Without an output coupling capacitor, the half-supply bias
across the load would result in both increased internal IC power dissipation and also possible loudspeaker damage.
AMPLIFIER POWER DISSIPATION
Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A
direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation.
Since the amplifier portion of the LM4960 has two operational amplifiers, the maximum internal power dissipation is 4 times that
of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation 1.
PDMAX(AMP) = 4(VDD)2 / (2π2ZL) (1)
where
ZL = Ro1 + Ro2 +1/2πfc
BOOST CONVERTER POWER DISSIPATION
At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power
dissipation within the LM2731 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation 2.
PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (2)
where DC is the duty cycle.
There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation.
TOTAL POWER DISSIPATION
The total power dissipation for the LM4960 can be calculated by adding Equation 1 and Equation 2 together to establish Equation
3:
PDMAX(TOTAL) = [4*(VDD)2/2π2ZL] + [DC x IIND(AVE)2 xRDS(ON)] (3)
The result from Equation 3 must not be greater than the power dissipation that results from Equation 4:
PDMAX = (TJMAX - TA) / θJA (4)
For the LQA28A, θJA = 59°C/W. TJMAX = 125°C for the LM4960. Depending on the ambient temperature, TA, of the system sur-
roundings, Equation 4 can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of
Equation 3 is greater than that of Equation 4, then either the supply voltage must be increased, the load impedance increased or
TA reduced. For the typical application of a 3V power supply, with V1 set to 12V and a 800nF + 20 load, the maximum ambient
temperature possible without violating the maximum junction temperature is approximately 118°C provided that device operation
LM4960
8 Copyright © 1999-2012, Texas Instruments Incorporated
is around the maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power dissipation
is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient tem-
perature may be increased accordingly. Refer to the Typical Performance Characteristics curves for power dissipation information
for lower output levels.
EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS
The LM4960’s exposed-DAP (die attach paddle) package (LD) provides a low thermal resistance between the die and the PCB to
which the part is mounted and soldered. The low thermal resistance allows rapid heat transfer from the die to the surrounding PCB
copper traces, ground plane, and surrounding air. The LD package should have its DAP soldered to a copper pad on the PCB. The
DAP’s PCB copper pad may be connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat
sink, and radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting an LD (LLP)
package is found in National Semiconductor’s Package Engineering Group under application note AN1187.
SHUTDOWN FUNCTION
In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth
transition into shutdown. Another solution is to use a single-pole, single-throw switch, and a pull-up resistor. One terminal of the
switch is connected to GND. The other side is connected to the two shutdown pins and the terminal of the pull-up resistor. The
remaining resistance terminal is connected to VDD. If the switch is open, then the external pull-up resistor connected to VDD will
enable the LM4960. This scheme guarantees that the shutdown pins will not float thus preventing unwanted state changes.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC converters, is
critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system
quality.
The best capacitors for use with the switching converter portion of the LM4960 are multi-layer ceramic capacitors. They have the
lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency
switching converters.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U and Y5F have
such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of
rated capacitance in many typical applications. Always consult capacitor manufacturer’s data curves before selecting a capacitor.
High-quality ceramic capacitors can be obtained from Taiyo-Yuden, AVX, and Murata.
POWER SUPPLY BYPASSING
As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor
location on both V1 and VDD pins should be as close to the device as possible.
SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER
One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the
choice of external components shown in Figure 1. The input coupling capacitor, Ci, forms a first order high pass filter which limits
low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons.
High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed
to couple in low frequencies without severe attenuation. But ceramic speakers used in portable systems, whether internal or ex-
ternal, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase
actual system performance.
In addition to system cost and size, click and pop performance is affected by the value of the input coupling capacitor, Ci. A high
value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes
from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on
desired low frequency response, turn-on pops can be minimized.
SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER
Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value. Bypass capacitor,
CB, is the most critical component to minimize turn-on pops since it determines how fast the amplifer turns on. The slower the
amplifier’s outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the smaller the turn-on pop. Choosing CB equal to 1.0µF
along with a small value of Ci (in the range of 0.039µF to 0.39µF), should produce a virtually clickless and popless shutdown function.
Although the device will function properly, (no oscillations or motorboating), with CB equal to 0.1µF, the device will be much more
susceptible to turn-on clicks and pops. Thus, a value of CB equal to 1.0µF is recommended in all but the most cost sensitive designs.
SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER
The LM4960 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic speakers, a
typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) will be needed as shown in
Figure 2 to bandwidth limit the amplifier.
This feedback capacitor creates a low pass filter that eliminates possible high frequency oscillations. Care should be taken when
calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 9
SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER
A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of
capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics
with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors
are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating
from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability.
In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching
losses, and output voltage ripple.
SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER
An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns
ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7µF, but
larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the
amount of EMI passed back along that line to other circuitry.
SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER
The output voltage is set using the external resistors R1 and R2 (see Figure 1). A value of approximately 13.3k is recommended
for R2 to establish a divider current of approximately 92µA. R1 is calculated using the formula:
R1 = R2 X (V2/1.23 − 1) (5)
FEED-FORWARD COMPENSATION FOR BOOST CONVERTER
Although the LM4960's internal Boost converter is internally compensated, the external feed-forward capacitor Cf is required for
stability (see Figure 1). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for
the zero fz should be approximately 6kHz. Cf1 can be calculated using the formula:
Cf1 = 1 / (2 X R1 X fz) (6)
SELECTING DIODES
The external diode used in Figure 1 should be a Schottky diode. A 20V diode such as the MBR0520 is recommended.
The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A
average but less than 1A, a Microsemi UPS5817 can be used.
DUTY CYCLE
The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter
can attain in continuous mode of operation. The duty cycle for a given boost application is defined as:
Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest sized component
and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor
ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean
more load power can be delivered because the energy stored during each switching cycle is:
E = L/2 X (lp)2
LM4960
10 Copyright © 1999-2012, Texas Instruments Incorporated
Where “lp” is the peak inductor current. An important point to observe is that the LM4960 will limit its switch current based on peak
current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.
Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest,
typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the inductor current
to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load
is reduced far enough, but a larger inductor stays “continuous” over a wider load current range.
To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10µH inductor) will be analyzed. We will
assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Since the frequency is 1.6MHz (nominal), the period is approximately 0.625µs. The duty cycle will be 62.5%, which means the ON-
time of the switch is 0.390µs. It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V.
Using the equation:
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON-time. Using these facts, we can
then show what the inductor current will look like during operation:
20076583
FIGURE 2. 10μH Inductor Current
5V - 12V Boost (LM4960)
During the 0.390µs ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFF-time. This
is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33mA, the inductor current will
begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost
converter, to make sure the ripple current is reasonable and continuous operation will be maintained at the typical load current
values.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is
illustrated in a graph in the typical performance characterization section which shows typical values of switch current as a function
of effective (actual) duty cycle.
CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP)
As shown in Figure 2 which depicts inductor current, the load current is related to the average inductor current by the relation:
ILOAD = IIND(AVG) x (1 - DC) (7)
Where "DC" is the duty cycle of the application. The switch current can be found by:
ISW = IIND(AVG) + 1/2 (IRIPPLE) (8)
Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L) (9)
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 11
Combining all terms, we can develop an expression which allows the maximum available load current to be calculated:
ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (10)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses
of the FET and diode.
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in equations 7 thru 10) is dependent on load current. A good approximation
can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current.
FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see
Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V.
The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical Perfor-
mance Characteristics curves.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for the LM4960 include, but are not limited to Taiyo-Yuden, Sumida, Coilcraft, Panasonic,
TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation
at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered
when selecting the current rating.
PCB LAYOUT GUIDELINES
High frequency boost converters require very careful layout of components in order to get stable operation and low noise. All
components must be as close as possible to the LM4802 device. It is recommended that a 4-layer PCB be used so that internal
ground planes are available.
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co will increase noise
and ringing.
2. The feedback components R1, R2 and Cf 1 must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace.
3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative
sides of capacitors Cs1 and Co.
GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION
This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power and ground traces.
Designers should note that these are only "rule-of-thumb" recommendations and the actual results will depend heavily on the final
layout.
Power and Ground Circuits
For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the analog power and ground
trace paths. Star trace routing techniques (bringing individual traces back to a central point rather than daisy chaining traces together
in a serial manner) can have a major impact on low level signal performance. Star trace routing refers to using individual traces to
feed power and ground to each circuit or even device. This technique will take require a greater amount of design time but will not
increase the final price of the board. The only extra parts required may be some jumpers.
Single-Point Power / Ground Connection
The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can be helpful in
minimizing high frequency noise coupling between the analog and digital sections. It is further recommended to place digital and
analog power traces over the corresponding digital and analog ground traces to minimize noise coupling.
Placement of Digital and Analog Components
All digital components and high-speed digital signals traces should be located as far away as possible from analog components
and circuit traces.
Avoiding Typical Design / Layout Problems
Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB layer. When traces
must cross over each other do it at 90 degrees. Running digital and analog traces at 90 degrees to each other from the top to the
bottom side as much as possible will minimize capacitive noise coupling and crosstalk.
LM4960
12 Copyright © 1999-2012, Texas Instruments Incorporated
Physical Dimensions inches (millimeters) unless otherwise noted
LLP, Plastic, Quad
Order Number LM4960SQ
NS Package Number SQA28A
LM4960
Copyright © 1999-2012, Texas Instruments Incorporated 13
Notes
Copyright © 1999-2012, Texas Instruments
Incorporated
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