General Description
MAX1846/MAX1847 high-efficiency PWM inverting con-
trollers allow designers to implement compact, low-
noise, negative-output DC-DC converters for telecom
and networking applications. Both devices operate
from +3V to +16.5V input and generate -500mV to
-200V output. To minimize switching noise, both
devices feature a current-mode, constant-frequency
PWM control scheme. The operating frequency can be
set from 100kHz to 500kHz through a resistor.
The MAX1846 is available in an ultra-compact 10-pin
µMAX®package. Operation at high frequency, compat-
ibility with ceramic capacitors, and inverting topology
without transformers allow for a compact design.
Compatibility with electrolytic capacitors and flexibility
to operate down to 100kHz allow users to minimize the
cost of external components. The high-current output
drivers are designed to drive a P-channel MOSFET and
allow the converter to deliver up to 30W.
The MAX1847 features clock synchronization and shut-
down functions. The MAX1847 can also be configured
to operate as an inverting flyback controller with an N-
channel MOSFET and a transformer to deliver up to
70W. The MAX1847 is available in a 16-pin QSOP
package.
Current-mode control simplifies compensation and pro-
vides good transient response. Accurate current-mode
control and over current protection are achieved
through low-side current sensing.
Applications
Cellular Base Stations
Networking Equipment
Optical Networking Equipment
SLIC Supplies
CO DSL Line Driver Supplies
Industrial Power Supplies
Automotive Electronic Power Supplies
Servers
VOIP Supplies
Features
o90% Efficiency
o+3.0V to +16.5V Input Range
o-500mV to -200V Output
oDrives High-Side P-Channel MOSFET
o100kHz to 500kHz Switching Frequency
oCurrent-Mode, PWM Control
oInternal Soft-Start
oElectrolytic or Ceramic Output Capacitor
oThe MAX1847 also offers:
Synchronization to External Clock
Shutdown
N-Channel Inverting Flyback Option
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
________________________________________________________________
Maxim Integrated Products
1
MAX1846
MAX1847
VL IN
COMP
FREQ
REF
GND FB
PGND
CS
EXT
P
POSITIVE
VIN
NEGATIVE
VOUT
Typical Operating Circuit
19-2091; Rev 2; 9/10
EVALUATION KIT
AVAILABLE
Ordering Information
PART TEMP RANGE PIN-PACKAGE
MAX1846EUB -40°C to +85°C 10 µMAX
MAX1846EUB+ -40°C to +85°C 10 µMAX
MAX1847EEE -40°C to +85°C 16 QSOP
MAX1847EEE+ -40°C to +85°C 16 QSOP
Pin Configurations appear at end of data sheet.
+
Denotes a lead(Pb)-free/RoHS-compliant package.
µMAX is a registered trademark of Maxim Integrated Products, Inc.
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim's website at www.maxim-ic.com.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ±1%, CVL = 0.47µF, CREF = 0.1µF, TA= 0°C to +85°C, unless
otherwise noted.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
IN, SHDN to GND ...................................................-0.3V to +20V
PGND to GND .......................................................-0.3V to +0.3V
VL to PGND for VIN ≤5.7V...........................-0.3V to (VIN + 0.3V)
VL to PGND for VIN > 5.7V .......................................-0.3V to +6V
EXT to PGND ...............................................-0.3V to (VIN + 0.3V)
REF, COMP to GND......................................-0.3V to (VL + 0.3V)
CS, FB, FREQ, POL, SYNC to GND .........................-0.3V to +6V
Continuous Power Dissipation (TA = +70°C)
10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW
16-Pin QSOP (derate 8.3mW/°C above +70°C)...........696mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow)
Lead(Pb)-free...............................................................+260°C
Containing lead(Pb) .....................................................+240°C
PARAMETER CONDITIONS MIN TYP MAX UNITS
PWM CONTROLLER
Operating Input Voltage Range 3.0 16.5 V
VIN rising 2.8 2.95
UVLO Threshold
VIN falling 2.6 2.74
V
UVLO Hysteresis 60 mV
FB Threshold No load -12 0 12 mV
FB Input Current VFB = -0.1V -50 -6 50 nA
Load Regulation CCOMP = 0.068µF, VOUT = -48V,
IOUT = 20mA to 200mA (Note 1) -1 0 %
Line Regulation CCOMP = 0.068µF, VOUT = -48V,
VIN = +8V to +16.5V, IOUT = 100mA 0.04 %
Current-Limit Threshold 85 100 115 mV
CS Input Current CS = GND 10 20 µA
Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V 0.75 1.2 mA
Shutdown Supply Current SHDN = GND, VIN = +3.0V to +16.5V 10 25 µA
REFERENCE AND VL REGULATOR
REF Output Voltage IREF = 50µA 1.236 1.25 1.264 V
REF Load Regulation IREF = 0 to 500µA -2 -15 mV
VL Output Voltage IVL = 100µA 3.85 4.25 4.65 V
VL Load Regulation IVL = 0.1mA to 2.0mA -20 -60 mV
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ±1%, CVL = 0.47µF, CREF = 0.1µF, TA= 0°C to +85°C, unless
otherwise noted.)
OSCILLATOR
RFREQ = 500kΩ ±1% 90 100 110
RFREQ = 147kΩ ±1% 255 300 345 Oscillator Frequency
RFREQ = 76.8kΩ ±1% 500
kHz
RFREQ = 500kΩ ±1% 93 96 97
RFREQ = 147kΩ ±1% 85 88 90
Maximum Duty Cycle
RFREQ = 76.8kΩ ±1% 80
%
SYNC Input Signal Duty-Cycle
Range 7 93 %
Minimum SYNC Input Logic-Low
Pulse Width 50 200 ns
SYNC Input Rise/Fall Time (Note 2) 200 ns
SYNC Input Frequency Range 100 550 kHz
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage 2.0 V
POL, SYNC, SHDN Input Low
Voltage 0.45 V
POL, SYNC Input Current POL, SYNC = GND or VL 20 40 µA
VSHDN = +5V or GND -12 -4 0
SHDN Input Current VSHDN = +16.5V 1.5 6
µA
SOFT-START
Soft-Start Clock Cycles 1024
Cycles
Soft-Start Levels 64
EXT OUTPUT
EXT Sink/Source Current VIN = +5V, VEXT forced to +2.5V 1 A
EXT high or low, tested with 100mA load, VIN = +5V 2 5
EXT On-Resistance EXT high or low, tested with 100mA load, VIN = +3V 5 10
Ω
Note 1: Production test correlates to operating conditions.
Note 2: Guaranteed by design and characterization.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
4 _______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ±1%, CVL = 0.47µF, CREF = 0.1µF, TA= -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER CONDITIONS MIN MAX UNITS
PWM CONTROLLER
Operating Input Voltage Range 3.0 16.5 V
VIN rising 2.95
UVLO Threshold VIN falling 2.6
V
FB Threshold No load -20 +20 mV
FB Input Current VFB = -0.1V -50 +50 nA
Load Regulation CCOMP = 0.068µF, VOUT = -48V,
IOUT= 20mA to 200mA (Note 1) -2 0 %
Current Limit Threshold 85 115 mV
CS Input Current CS = GND 20 µA
Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V 1.2 mA
Shutdown Supply Current SHDN = GND, VIN = +3.0V to +16.5V 25 µA
REFERENCE AND VL REGULATOR
REF Output Voltage IREF = 50µA 1.225 1.275 V
REF Load Regulation IREF = 0 to 500µA -15 mV
VL Output Voltage IVL = 100µA 3.85 4.65 V
VL Load Regulation IVL = 0.1mA to 2.0mA -60 mV
OSCILLATOR
RFREQ = 500kΩ ±1% 84 116
Oscillator Frequency RFREQ = 147kΩ ±1% 255 345
kHz
RFREQ = 500kΩ ±1% 93 98
Maximum Duty Cycle RFREQ = 147kΩ ±1% 84 93
%
SYNC Input Signal Duty-Cycle
Range 7 93 %
Minimum SYNC Input Logic Low
Pulse Width 200 ns
SYNC Input Rise/Fall Time (Note 2) 200 ns
SYNC Input Frequency Range 100 550 kHz
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage 2.0 V
POL, SYNC, SHDN Input Low
Voltage 0.45 V
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
_______________________________________________________________________________________
5
100
0
1 10 100 1000 10,000
EFFICIENCY vs. LOAD CURRENT
20
MAX1846/7 toc01
LOAD CURRENT (mA)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 5V
VIN = 16.5V
VOUT = -5V
APPLICATION CIRCUIT A
100
0
1 10 100 1000 10,000
EFFICIENCY vs. LOAD CURRENT
20
MAX1846/7 toc02
LOAD CURRENT (mA)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 5V
VIN = 3.3V
VOUT = -12V
APPLICATION CIRCUIT B
VIN = 3V
90
0
1 100010010
EFFICIENCY vs. LOAD CURRENT
30
10
70
50
100
40
20
80
60
MAX1846/7 toc03
LOAD CURRENT (mA)
EFFICIENCY (%)
VIN = 12V
VIN = 16.5V
APPLICATION CIRCUIT C VOUT = -48V
-12.10
-12.04
-12.06
-12.08
-12.02
-12.00
-11.98
-11.96
-11.94
-11.92
-11.90
0 200100 300 400 500 600
OUTPUT VOLTAGE LOAD REGULATION
MAX1846/7 toc04
LOAD CURRENT (mA)
OUTPUT VOLTAGE (V)
APPLICATION CIRCUIT B VIN = 5V
0
0.4
0.2
0.8
0.6
1.0
1.2
1.4
1.6
0462 8 10 12 14 16
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
MAX1846/7 toc05
VIN (V)
IIN (mA)
VFB = -0.1V
1.238
1.246
1.242
1.254
1.250
1.258
1.262
-40 20 40-20 0 60 80 100
REFERENCE VOLTAGE
vs. TEMPERATURE
MAX1846/7 toc06
TEMPERATURE (°C)
VREF (V)
Typical Operating Characteristics
(Circuit references are from Table 1 in the
Main Application Circuits
section, CVL = 0.47µF, CREF= 0.1µF, TA = +25°C, unless otherwise
noted.)
ELECTRICAL CHARACTERISTICS (continued)
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ±1%, CVL = 0.47µF, CREF = 0.1µF, TA= -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER CONDITIONS MIN MAX UNITS
POL, SYNC Input Current POL, SYNC = GND or VL 40 µA
V SHDN = +5V or GND -12 0
SHDN Input Current V SHDN = +16.5V 6
µA
EXT OUTPUT
EXT high or low, IEXT = 100mA, VIN = +5V 7.5
EXT On-Resistance EXT high or low, IEXT = 100mA, VIN = +3V 12
Ω
Note 3: Parameters to -40°C are guaranteed by design and characterization.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
6 _______________________________________________________________________________________
SHUTDOWN SUPPLY CURRENT
vs. TEMPERATURE
MAX1846/7 toc10
0
2
6
4
12
14
10
8
16
SHUTDOWN SUPPLY CURRENT (µA)
-40 0 20-20 40 60 80 100
TEMPERATURE (°C)
VIN = 10V
VIN = 16.5V
VIN = 3V
0
4
2
8
6
12
10
14
-40 0 20-20 40 60 80 100
OPERATING CURRENT
vs. TEMPERATURE
MAX1846/7 toc11
TEMPERATURE (°C)
OPERATING CURRENT (mA)
A
B
C
A: VIN = 3V, VOUT = -12V
B: VIN = 5V, VOUT = -5V
C: VIN = 16.5V, VOUT = -5V
APPLICATION CIRCUIT A
0
100
300
200
400
500
0200100 300 400 500 600
SWITCHING FREQUENCY
vs. RFREQ
MAX1846/7 toc12
RFREQ (kΩ)
fOSC (kHz)
SWITCHING FREQUENCY
vs. TEMPERATURE
MAX1846/7 toc13
294
295
297
296
300
301
299
298
302
FREQUENCY (kHz)
-40 0 20-20 40 60 80 100
TEMPERATURE (°C)
RFREQ = 147kΩ ±1%
0
40
20
60
120
140
100
80
160
0 2000 4000 6000 8000 10,000
EXT RISE/FALL TIME
vs. CAPACITANCE
MAX1846/7 toc14
CAPACITANCE (pF)
TIME (ns)
RISE TIME
FALL TIME
VIN = 12V
IL
0
5V/div
1A/div
5V/div
VOUT
EXITING SHUTDOWN
MAX1846/7 toc15
APPLICATION CIRCUIT B
1ms/div
SHDN
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the
Main Application Circuits
section, CVL = 0.47µF, CREF= 0.1µF, TA = +25°C, unless otherwise
noted.)
1.240
1.245
1.255
1.250
1.260
0 100 200 300 400 500
REFERENCE LOAD REGULATION
MAX1846/7 toc07
IREF (µA)
VREF (V)
4.100
4.180
4.140
4.260
4.220
4.300
4.340
-40 20 40-20 0 60 80 100
VL VOLTAGE
vs. TEMPERATURE
MAX1846/7 toc08
TEMPERATURE (°C)
VL (V)
IVL = 0
4.22
4.23
4.24
4.25
4.26
4.27
0 0.8 1.00.4 0.60.2 1.2 1.4 1.6 1.8 2.0
VL LOAD REGULATION
MAX1846/7 toc09
IVL (mA)
VL (V)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
_______________________________________________________________________________________ 7
IL
0
5V/div
1A/div
5V/div
VOUT
ENTERING SHUTDOWN
MAX1846/7 toc16
APPLICATION CIRCUIT B
1ms/div
SHDN
IL
LX
1A/div
10V/div
100mV/div
VOUT
HEAVY-LOAD SWITCHING
WAVEFORM
MAX1846/7 toc17
APPLICATION CIRCUIT B
1µs/div
ILOAD = 600mA
IL
LX
1A/div
10V/div
100mV/div
VOUT
LIGHT-LOAD SWITCHING
WAVEFORM
MAX1846/7 toc18
APPLICATION CIRCUIT B
1µs/div
ILOAD = 50mA
IL1A/div
500mV/div
ILOAD
VOUT
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc19
APPLICATION CIRCUIT B
2ms/div
ILOAD = 10mA to 400mA
IL500mA/div
200mV/div
ILOAD
VOUT
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc20
APPLICATION CIRCUIT C
400µs/div
ILOAD = 4mA to 100mA
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the
Main Application Circuits
section, CVL = 0.47µF, CREF= 0.1µF, TA = +25°C, unless otherwise
noted.)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
8 _______________________________________________________________________________________
Pin Description
PIN
MAX1846 MAX1847 NAME FUNCTION
— 1 POL
Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external
PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS low-
side FET in transformer-based applications.
1 2 VL VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND.
2 3 FREQ
Oscillator Frequency Set Input. Connect a resistor (RFREQ) from FREQ to GND to set the
internal oscillator frequency from 100kHz (RFREQ = 500kΩ) to 500kHz (RFREQ = 76.8kΩ).
RFREQ is still required if an external clock is used at SYNC. See Setting the Operating
Frequency section.
3 4 COMP Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network
from COMP to GND for loop compensation. See Design Procedure.
4 5 REF 1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic
capacitor from REF to GND.
56FB
Feedback Input. Connect FB to the center of a resistor-divider connected between the
output and REF. The FB threshold is 0.
— 7, 9 N.C. No Connection
—8SHDN Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect
to IN for normal operation.
6 10, 11 GND Analog Ground. Connect to PGND.
7 12 PGND Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND.
813CS
P osi ti ve C ur r ent- S ense Inp ut. C onnect a cur r ent- sense r esi stor ( RC S
) b etw een C S and
9 14 EXT External MOSFET Gate-Driver Output. EXT swings from IN to PGND.
10 15 IN Power-Supply Input
— 16 SYNC
Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set
the internal oscillator frequency with RFREQ. Drive SYNC with a logic-level clock input
signal to externally set the converter’s operating frequency. DC-DC conversion cycles
initiate on the rising edge of the input clock signal. Note that when driving SYNC with an
external signal, RFREQ must still be connected to FREQ.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
_______________________________________________________________________________________ 9
MAX1847
REF FB
GND
CS
EXT
PGND
FREQ
COMP
VL IN
0.22µF
150kΩ
0.1µF
R2
10.0kΩ
1%
R1
95.3kΩ
1%
0.02Ω
1W
10µH
DO5022P-103
CMSH5-40
47µF
16V
FDS6375
3 x 22µF
10V
POL
SYNC
SHDN
10kΩ
47µF
16V
22kΩ
1
2
16
8
3
4
5
10, 11
6
12
13
14
15
7, 9
N.C.
VIN
+3V to +5.5V
0.47µFVOUT
-12V AT 400mA
SANYO
16TPB47M
1200pF
220pF
Typical Application Circuit
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
10 ______________________________________________________________________________________
MAX1846
MAX1847
STARTUP
CIRCUITRY
CONTROL
CIRCUITRY
VL
REGULATOR
OSCILLATOR
REFERENCE
SLOPE
COMP
UNDER-
VOLTAGE
LOCK OUT
SOFT-START
IN
MAX1847 ONLY
MAX1847 ONLY
POL
FREQ
COMP
FB
REF
SYNC
SHDN
ERROR
AMPLIFIER
CURRENT-
SENSE
AMPLIFIER
GND
ERROR
COMPARATOR
EXT DRIVER
EXT
VL
CS
PGND
PGND
GM
X3.3
Functional Diagram
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
______________________________________________________________________________________ 11
Detailed Description
The MAX1846/MAX1847 current-mode PWM controllers
use an inverting topology that is ideal for generating
output voltages from -500mV to -200V. Features include
shutdown, adjustable internal operating frequency or
synchronization to an external clock, soft-start,
adjustable current limit, and a wide (+3V to +16.5V)
input range.
PWM Controller
The architecture of the MAX1846/MAX1847 current-
mode PWM controller is a BiCMOS multi-input system
that simultaneously processes the output-error signal,
the current-sense signal, and a slope-compensation
ramp (
Functional Diagram
). Slope compensation pre-
vents subharmonic oscillation, a potential result in cur-
rent-mode regulators operating at greater than 50%
duty cycle. The controller uses fixed-frequency, cur-
rent-mode operation where the duty ratio is set by the
input-to-output voltage ratio. The current-mode feed-
back loop regulates peak inductor current as a function
of the output error signal.
Internal Regulator
The MAX1846/MAX1847 incorporate an internal low-
dropout regulator (LDO). This LDO has a 4.25V output
and powers all MAX1846/MAX1847 functions (exclud-
ing EXT) for the primary purpose of stabilizing the per-
formance of the IC over a wide input voltage range
(+3V to +16.5V). The input to this regulator is connect-
ed to IN, and the dropout voltage is typically 100mV, so
that when VIN is less than 4.35V, VL is typically VIN
minus 100mV. When the LDO is in dropout, the
MAX1846/MAX1847 still operate with VIN as low as 3V.
For best performance, it is recommended to connect
VL to IN when the input supply is less than 4.5V.
Undervoltage Lockout
The MAX1846/MAX1847 have an undervoltage lockout
circuit that monitors the voltage at VL. If VL falls below
the UVLO threshold (2.8V typ), the control logic turns the
P-channel FET off (EXT high impedance). The rest of the
IC circuitry is still powered and operating. When VL
increases to 60mV above the UVLO threshold, the IC
resumes operation from a start up condition (soft-start).
Soft-Start
The MAX1846/MAX1847 feature a “digital” soft-start
that is preset and requires no external capacitor. Upon
startup, the FB threshold decrements from the refer-
ence voltage to 0 in 64 steps over 1024 cycles of fOSC
or fSYNC. See the
Typical Operating Characteristics
for
a scope picture of the soft-start operation. Soft-start is
implemented: 1) when power is first applied to the IC,
2) when exiting shutdown with power already applied,
and 3) when exiting undervoltage lockout.
Shutdown (MAX1847 only)
The MAX1847 shuts down to reduce the supply current
to 10µA when SHDN is low. In this mode, the internal ref-
erence, error amplifier, comparators, and biasing circuit-
ry turn off. The EXT output becomes high impedance
and the external pullup resistor connected to EXT pulls
VEXT to VIN, turning off the P-channel MOSFET. When in
shutdown mode, the converter’s output goes to 0.
Frequency Synchronization
(MAX1847 only)
The MAX1847 is capable of synchronizing its switching
frequency with an external clock source. Drive SYNC
with a logic-level clock input signal to synchronize the
MAX1847. A switching cycle starts on the rising edge
of the signal applied to SYNC. Note that the frequency
of the signal applied to SYNC must be higher than the
default frequency set by RFREQ. This frequency is
required so that the internal clock does not start a
switching cycle prematurely. If SYNC is inactive for an
entire clock cycle of the internal oscillator, the internal
oscillator takes over the switching operation. Choose
RFREQ such that fOSC = 0.9 fSYNC.
EXT Polarity (MAX1847 only)
The MAX1847 features an option to utilize an N-channel
MOSFET configuration, rather than the typical p-chan-
nel MOSFET configuration (Figure 1). In order to drive
the different polarities of these MOSFETs, the MAX1847
is capable of reversing the phase of EXT by 180
degrees. When driving a P-channel MOSFET, connect
POL to GND. When driving an n-channel MOSFET, con-
nect POL to VL. These POL connections ensure the
proper polarity for EXT. For design guidance in regard
to this application, refer to the MAX1856 data sheet.
Design Procedure
Initial Specifications
In order to start the design procedure, a few parameters
must be identified: the minimum input voltage expected
(VIN(MIN)), the maximum input voltage expected
(VIN(MAX)), the desired output voltage (VOUT), and the
expected maximum load current (ILOAD).
Calculate the Equivalent Load Resistance
This is a simple calculation used to shorten the verifica-
tion equations:
RLOAD = VOUT / ILOAD
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
12 ______________________________________________________________________________________
Calculate the Duty Cycle
The duty cycle is the ratio of the on-time of the MOSFET
switch to the oscillator period. It is determined by the
ratio of the input voltage to the output voltage. Since
the input voltage typically has a range of operation, a
minimum (DMIN) and maximum (DMAX) duty cycle is
calculated by:
where VDis the forward drop across the output diode,
VSW is the drop across the external FET when on, and
VLIM is the current-limit threshold. To begin with,
assume VD= 0.5V for a Schottky diode, VSW = 100mV,
and VLIM = 100mV. Remember that VOUT is negative
when using this formula.
Setting the Output Voltage
The output voltage is set using two external resistors to
form a resistive-divider to FB between the output and
REF (refer to R1 and R2 in Figure 1). VREF is nominally
1.25V and the regulation voltage for FB is nominally 0.
The load presented to the reference by the feedback
resistors must be less than 500µA to guarantee that
VREF is in regulation (see
Electrical Characteristics
Table
). Conversely, the current through the feedback
resistors must be large enough so that the leakage cur-
rent of the FB input (50nA) is insignificant. Therefore,
select R2 so that IR2 is between 50µA and 250µA.
IR2 = VREF / R2
where VREF = 1.25V. A typical value for R2 is 10kΩ.
Once R2 is selected, calculate R1 with the following
equation:
R1 = R2 x (-VOUT / VREF)
Setting the Operating Frequency
The MAX1846/MAX1847 are capable of operating at
switching frequencies from 100kHz to 500kHz. Choice
of operating frequency depends on a number of fac-
tors:
1) Noise considerations may dictate setting (or syn-
chronizing) fOSC above or below a certain fre-
quency or band of frequencies, particularly in RF
applications.
DVV
VVVVV
MAX OUT D
IN MIN SW LIM OUT D
=+
+
−
−−−
()
DVV
VVVVV
MIN OUT D
IN MAX SW LIM OUT D
=+
−+
−
−−
()
COMP
0.033µF270kΩ
SYNC
150kΩ
GND
FREQ
VL
VIN
+12V
VOUT
-48V AT 100mA
12µF
100V
12µF
25V
2
14
13
12
6
5
3
4
0.05Ω
0.5W
1800pF
15
0.47µF
0.1µF
10, 11
EXT
PGND
REF
IN
CS
FB
N.C.
MAX1847
10.0kΩ
1%
383kΩ
1%
7, 9
POL
1
8
16
VP1-0190
12.2µH1:4
CMR1U-02
470Ω
100pF
100V
SHDN
IRLL2705
Figure 1. Using an N-Channel MOSFET (MAX1847 only)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
______________________________________________________________________________________ 13
2) Higher frequencies allow the use of smaller value
(hence smaller size) inductors and capacitors.
3) Higher frequencies consume more operating
power both to operate the IC and to charge and
discharge the gate at the external FET, which
tends to reduce the efficiency at light loads.
4) Higher frequencies may exhibit lower overall effi-
ciency due to more transition losses in the FET;
however, this shortcoming can often be nullified
by trading some of the inductor and capacitor size
benefits for lower-resistance components.
5) High-duty-cycle applications may require lower
frequencies to accommodate the controller mini-
mum off-time of 0.4µs. Calculate the maximum
oscillator frequency with the following formula:
Remember that VOUT is negative when using this formula.
When running at the maximum oscillator frequency
(fOSCILLATOR) and maximum duty cycle (DMAX), do not
exceed the minimum value of DMAX stated in the
Electrical Characteristics
table. For designs that
exceed the DMAX and fOSC(MAX), an autotransformer
can reduce the duty cycle and allow higher operating
frequencies.
The oscillator frequency is set by a resistor, RFREQ,
which is connected from FREQ to GND. The relation-
ship between fOSC (in Hz) and RFREQ (in Ω) is slightly
nonlinear, as illustrated in the Typical Operating
Characteristics. Choose the resistor value from the
graph and check the oscillator frequency using the fol-
lowing formula:
External Synchronization (MAX1847 only)
The SYNC input provides external-clock synchroniza-
tion (if desired). When SYNC is driven with an external
clock, the frequency of the clock directly sets the
MAX1847’s switching frequency. A rising clock edge
on SYNC is interpreted as a synchronization input. If
the sync signal is lost, the internal oscillator takes over
at the end of the last cycle, and the frequency is
returned to the rate set by RFREQ. Choose RFREQ such
that fOSC = 0.9 x fSYNC.
Choosing Inductance Value
The inductance value determines the operation of the
current-mode regulator. Except for low-current applica-
tions, most circuits are more efficient and economical
operating in continuous mode, which refers to continu-
ous current in the inductor. In continuous mode there is
a trade-off between efficiency and transient response.
Higher inductance means lower inductor ripple current,
lower peak current, lower switching losses, and, there-
fore, higher efficiency. Lower inductance means higher
inductor ripple current and faster transient response. A
reasonable compromise is to choose the ratio of induc-
tor ripple current to average continuous current at mini-
mum duty cycle to be 0.4. Calculate the inductor ripple
with the following formula:
Then calculate an inductance value:
L = (VIN(MAX) / IRIPPLE) x (DMIN / fOSC)
Choose the closest standard value. Once again, remem-
ber that VOUT is negative when using this formula.
Determining Peak Inductor Current
The peak inductor current required for a particular out-
put is:
ILPEAK = ILDC + (ILPP / 2)
where ILDC is the average DC inductor current and ILPP
is the inductor peak-to-peak ripple current. The ILDC
and ILPP terms are determined as follows:
where L is the selected inductance value. The satura-
tion rating of the selected inductor should meet or
exceed the calculated value for ILPEAK, although most
coil types can be operated up to 20% over their satura-
tion rating without difficulty. In addition to the saturation
criteria, the inductor should have as low a series resis-
II
D
I
VVVxD
Lxf
LDC LOAD
MAX
LPP
IN MIN SW LIM MAX
OSC
=−
()
=−−
()
()
1
I
IVVVVV
VVV
RIPPLE
LOAD MAX IN MAX SW LIM OUT D
IN MAX SW LIM
=
×× +
()
()
−−−
−−
04. () ()
()
f
RR
OSC
FREQ FREQ
=
×
()
+×
()
××
()
×
()
⎡
⎣
⎢⎤
⎦
⎥
−− −
−
1
5 21 10 1 92 10 4 86 10
711 19
2
. . .
fVVV
VVVVV
t
OSC MAX
IN MIN SW LIM
IN MIN SW LIM OUT D
OFF MIN
()
()
()
()
=+
×
−−
−−−
1
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
14 ______________________________________________________________________________________
tance as possible. For continuous inductor current, the
power loss in the inductor resistance (PLR) is approxi-
mated by:
where RLis the inductor series resistance.
Once the peak inductor current is calculated, the cur-
rent sense resistor, RCS, is determined by:
RCS = 85mV / ILPEAK
For high peak inductor currents (>1A), Kelvin-sensing
connections should be used to connect CS and PGND
to RCS. Connect PGND and GND together at the
ground side of RCS. A lowpass filter between RCS and
CS may be required to prevent switching noise from
tripping the current-sense comparator at heavy loads.
Connect a 100Ωresistor between CS and the high side
of RCS, and connect a 1000pF capacitor between CS
and GND.
Checking Slope-Compensation Stability
In a current-mode regulator, the cycle-by-cycle stability
is dependent on slope compensation to prevent sub-
harmonic oscillation at duty cycles greater than 50%.
For the MAX1846/MAX1847, the internal slope compen-
sation is optimized for a minimum inductor value (LMIN)
with respect to duty cycle. For duty cycles greater then
50%, check stability by calculating LMIN using the fol-
lowing equation:
where VIN(MIN) is the minimum expected input voltage,
Msis the Slope Compensation Ramp (41 mV/µs) and
DMAX is the maximum expected duty cycle. If LMIN is
larger than L, increase the value of L to the next stan-
dard value that is larger than LMIN to ensure slope
compensation stability.
Choosing the Inductor Core
Choosing the most cost-effective inductor usually
requires optimizing the field and flux with size. With
higher output voltages the inductor may require many
turns, and this can drive the cost up. Choosing an
inductor value at LMIN can provide a good solution if
discontinuous inductor current can be tolerated.
Powdered iron cores can provide the most economical
solution but are larger in size than ferrite.
Power MOSFET Selection
The MAX1846/MAX1847 drive a wide variety of P-chan-
nel power MOSFETs (PFETs). The best performance,
especially with input voltages below 5V, is achieved
with low-threshold PFETs that specify on-resistance
with a gate-to-source voltage (VGS) of 2.7V or less.
When selecting a PFET, key parameters include:
•Total gate charge (QG)
•Reverse transfer capacitance (CRSS)
•On-resistance (RDS(ON))
•Maximum drain-to-source voltage (VDS(MAX))
•Minimum threshold voltage (VTH(MIN))
At high-switching rates, dynamic characteristics (para-
meters 1 and 2 above) that predict switching losses
may have more impact on efficiency than RDS(ON),
which predicts DC losses. QGincludes all capacitance
associated with charging the gate. In addition, this
parameter helps predict the current needed to drive the
gate at the selected operating frequency. The power
MOSFET in an inverting converter must have a high
enough voltage rating to handle the input voltage plus
the magnitude of the output voltage and any spikes
induced by leakage inductance and ringing.
An RC snubber circuit across the drain to ground might
be required to reduce the peak ringing and noise.
Choose RDS(ON)(MAX) specified at VGS < VIN(MIN) to be
one to two times RCS. Verify that VIN(MAX) < VGS(MAX)
and VDS(MAX) > VIN(MAX) - VOUT + VD. Choose the rise-
and fall-times (tR, tF) to be less than 50ns.
Output Capacitor Selection
The output capacitor (COUT) does all the filtering in an
inverting converter. The output ripple is created by the
variations in the charge stored in the output capacitor
with each pulse and the voltage drop across the
capacitor’s equivalent series resistance (ESR) caused
by the current into and out of the capacitor. There are
two properties of the output capacitor that affect ripple
voltage: the capacitance value, and the capacitor’s
ESR. The output ripple due to the output capacitor’s
value is given by:
VRIPPLE-C = (ILOAD DMAX TOSC ) / COUT
The output ripple due to the output capacitor’s ESR is
given by:
VRIPPLE-R = ILPP RESR
These two ripple voltages are additive and the total out-
put ripple is:
VRIPPLE-T = VRIPPLE-C + VRIPPLE-R
LVxRM
xxD D
MIN IN MIN CS S
MAX MAX
=
()
⎡
⎣
⎢⎤
⎦
⎥
()()
⎡
⎣
⎢⎤
⎦
⎥
−−
() /
/211
PRx
I
ID
LR L LOAD
MAX
~
−
⎛
⎝
⎜⎞
⎠
⎟
2
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
______________________________________________________________________________________ 15
The ESR-induced ripple usually dominates this last
equation, so typically output capacitor selection is
based mostly upon the capacitor’s ESR, voltage rating,
and ripple current rating. Use the following formula to
determine the maximum ESR for a desired output ripple
voltage (VRIPPLE-D):
RESR = VRIPPLE-D / ILPP
Select a capacitor with ESR rating less than RESR. The
value of this capacitor is highly dependent on dielectric
type, package size, and voltage rating. In general, when
choosing a capacitor, it is recommended to use low-ESR
capacitor types such as ceramic, organic, or tantalum
capacitors. Ensure that the selected capacitor has suffi-
cient margin to safely handle the maximum RMS ripple
current.
For continuous inductor current the maximum RMS ripple
current in the output filter capacitor is:
Choosing Compensation Components
The MAX1846/MAX1847 are externally loop-compen-
sated devices. This feature provides flexibility in
designs to accommodate a variety of applications.
Proper compensation of the control loop is important to
prevent excessive output ripple and poor efficiency
caused by instability. The goal of compensation is to
cancel unwanted poles and zeros in the DC-DC con-
verter’s transfer function created by the power-switch-
ing and filter elements. More precisely, the objective of
compensation is to ensure stability by ensuring that the
DC-DC converter’s phase shift is less than 180° by a
safe margin, at the frequency where the loop gain falls
below unity. One method for ensuring adequate phase
margin is to introduce corresponding zeros and poles
in the feedback network to approximate a single-pole
response with a -20dB/decade slope all the way to
unity-gain crossover.
Calculating Poles and Zeros
The MAX1846/MAX1847 current-mode architecture
takes the double pole caused by the inductor and out-
put capacitor and shifts one of these poles to a much
higher frequency to make loop compensation easier.
To compensate these devices, we must know the cen-
ter frequencies of the right-half plane zero (zRHP) and
the higher frequency pole (pOUT2). Calculate the zRHP
frequency with the following formula:
The calculations for pOUT2 are very complex. For most
applications where VOUT does not exceed -48V (in a
negative sense), the pOUT2 will not be lower than 1/8th
of the oscillator frequency and is generally at a higher
frequency than zRHP. Therefore:
pOUT2 ≥0.125 fOSC
A pole is created by the output capacitor and the load
resistance. This pole must also be compensated and
its center frequency is given by the formula:
pOUT1 = 1 / (2πRLOAD COUT)
Finally, there is a zero introduced by the ESR of the out-
put capacitor. This zero is determined from the follow-
ing equation:
zESR = 1 / (2πCOUT RESR)
Calculating the Required Pole Frequency
To ensure stability of the MAX1846/MAX1847, the gain
of the error amplifier must roll-off the total loop gain to 1
before ZRHP or POUT2 occurs. First, calculate the DC
open-loop gain, ADC:
where:
ACS is the current sense amplifier gain = 3.3
B is the feedback-divider attenuation factor =
GMis the error-amplifier transconductance =
400 µA/V
ROis the error-amplifier output resistance = 3 MΩ
RCS is the selected current-sense resistor
Determining the Compensation Component Values
Select a unity-gain crossover frequency (fCROS), which
is lower than zRHP and pOUT2 and higher than pOUT1.
Using fCROS, calculate the compensation resistor
(RCOMP).
RfxR
AxP f
COMP CROS O
DC OUT CROS
=−
1
R
RR
2
12+
ABxG R D R
AxR
DC M O MAX LOAD
CS CS
xx
=−()1
Z
DxV VxR
xV L
RHP
MAX IN MIN OUT LOAD
OUT
=
−−
()
−
()
⎡
⎣
⎢
⎢
⎤
⎦
⎥
⎥
×
()
1
2
2
()
π
II
ID
xD D
RMS LOAD
MAX
MAX MAX
=−−2
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
16 ______________________________________________________________________________________
Select the next smaller standard value of resistor and
then calculate the compensation capacitor required to
cancel out the output-capacitor-induced pole (POUT1)
determined previously.
Choose the next larger standard value of capacitor.
In order for pCOMP to compensate the loop, the open-
loop gain must reach unity at a lower frequency than
the right-half-plane zero or the second output pole,
whichever is lower in frequency. If the second output
pole and the right-half-plane zero are close together in
frequency, the higher resulting phase shift at unity gain
may require a lower crossover frequency. For duty
cycles greater than 50%, slope compensation reduces
ADC, reducing the actual crossover frequency from
fCROS. It is also a good practice to reduce noise on
COMP with a capacitor (CCOMP2) to ground. To avoid
adding extra phase margin at the crossover, the
capacitor (CCOMP2) should roll-off noise at five times
the crossover frequency. The value for CCOMP2 can be
found using:
It might require a couple iterations to obtain a suitable
combination of compensation components.
Finally, the zero introduced by the output capacitor’s
ESR must be compensated. This compensation is
accomplished by placing a capacitor between REF
and FB creating a pole directly in the feedback loop.
Calculate the value of this capacitor using the frequen-
cy of zESR and the selected feedback resistor values
with the formula:
When using low-ESR, ceramic chip capacitors (MLCCs)
at the output, calculate the value of CFB as follows:
Applications Information
Maximum Output Power
The maximum output power that the MAX1846/MAX1847
can provide depends on the maximum input power
available and the circuit’s efficiency:
POUT(MAX) = Efficiency PIN(MAX)
Furthermore, the efficiency and input power are both
functions of component selection. Efficiency losses can
be divided into three categories: 1) resistive losses
across the inductor, MOSFET on-resistance, current-
sense resistor, rectification diode, and the ESR of the
input and output capacitors; 2) switching losses due to
the MOSFET’s transition region, and charging the MOS-
FET’s gate capacitance; and 3) inductor core losses.
Typically, 80% efficiency can be assumed for initial cal-
culations. The required input power depends on the
inductor current limit, input voltage, output voltage, out-
put current, inductor value, and the switching frequen-
cy. The maximum output power is approximated by the
following formula:
PMAX = [VIN - (VLIM + ILIM x RDS(ON))] x ILIM x
[1 - (LIR / 2)] x [(-VOUT + VD) / (VIN - VSW - VLIM
- VOUT + VD)]
where ILIM is the peak current limit and LIR is the
inductor current-ripple ratio and is calculated by:
LIR = ILPP / ILDC
Again, remember that VOUT for the MAX1846/
MAX1847 is negative.
Diode Selection
The MAX1846/MAX1847’s high-switching frequency
demands a high-speed rectifier. Schottky diodes are
recommended for most applications because of their fast
recovery time and low forward voltage. Ensure that the
diode’s average current rating exceeds the peak inductor
current by using the diode manufacturer’s data.
Additionally, the diode’s reverse breakdown voltage must
exceed the potential difference between VOUT and the
input voltage plus the leakage-inductance spikes. For
high output voltages (-50V or more), Schottky diodes may
not be practical because of this voltage requirement. In
these cases, use an ultrafast recovery diode with ade-
quate reverse-breakdown voltage.
Input Filter Capacitor
The input capacitor (CIN) must provide the peak current
into the inverter. This capacitor is selected the same way
CRR
fRR
FB OSC
.
=+
×× ××
12
12
2314
CRxCx
RR
RxR
FB ESR OUT
=+
12
12
CRR
xxfxRxR
COMP O COMP
CROS O COMP
25628
.
=+
CxP xR
COMP OUT COMP
.
=1
628 1
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
______________________________________________________________________________________ 17
as the output capacitor (COUT). Under ideal conditions,
the RMS current for the input capacitor is the same as the
output capacitor. The capacitor value and ESR must be
selected to reduce noise to an acceptable value and also
handle the ripple current (INRMS) where:
Bypass Capacitor
In addition to CIN and COUT, other ceramic bypass
capacitors are required with the MAX1846/MAX1847.
Bypass REF to GND with a 0.1µF or larger capacitor.
Bypass VLto GND with a 0.47µF or larger capacitor. All
bypass capacitors should be located as close to their
respective pins as possible.
PC Board Layout Guidelines
Good PC board layout and routing are required in high-
frequency-switching power supplies to achieve good reg-
ulation, high efficiency, and stability. It is strongly
recommended that the evaluation kit PC board layouts be
followed as closely as possible. Place power components
as close together as possible, keeping their traces short,
direct, and wide. Avoid interconnecting the ground pins
of the power components using vias through an internal
ground plane. Instead, keep the power components
close together and route them in a “star” ground configu-
ration using component-side copper, then connect the
star ground to internal ground using multiple vias.
Main Application Circuits
The MAX1846/MAX1847 are extremely versatile devices.
Figure 2 shows a generic schematic of the MAX1846.
Table 1 lists component values for several typical appli-
cations. These component values also apply to the
MAX1847. The first two applications are featured in the
MAX1846/MAX1847 EV kit.
IxI
ID
xD D
NRMS
MAX
MAX MAX
O
=−
−
12 2
.
( )
COMP
CCOMP RCOMP
RFREQ
GND
FREQ
VL
VIN
VOUT
COUT
CIN
1
9
8
7
5
4
2
3
P
L1
RCS
CFB
D1
10
22k
0.47µF
0.1µF
6
EXT
PGND
REF
IN
CS
FB
MAX1846
R2
R1
APPLICATION B
ONLY
NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.
CCOMP2
Figure 2. MAX1846 Main Application Circuit
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
18 ______________________________________________________________________________________
SUPPLIER COMPONENT PHONE WEBSITE
AVX Capacitors 803-946-0690 www.avxcorp.com
Central Semiconductor Diodes 516-435-1110 www.centralsemi.com
Coilcraft Inductors 847-639-6400 www.coilcraft.com
Dale Resistors 402-564-3131 www.vishay.com/company/brands/dale/
Fairchild MOSFETs 408-721-2181 www.fairchildsemi.com
International
Rectifier MOSFETs 310-322-3331 www.irf.com
IRC Resistors 512-992-7900 www.irctt.com
Kemet Capacitors 864-963-6300 www.kemet.com
On Semiconductor MOSFETs, Diodes 602-303-5454 www.onsemi.com
Panasonic Capacitors, resistors 201-348-7522 www.panasonic.com
Sanyo Capacitors 619-661-6835 www.secc.co.jp
Siliconix MOSFETs 408-988-8000 www.siliconix.com
Sprague Capacitors 603-224-1961 www.vishay.com/company/brands/sprague/
Sumida Inductors 847-956-0666 www.remtechcorp.com
Vitramon Resistors 203-268-6261 www.vishay.com/company/brands/vitramon/
Component Suppliers
Note: Indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.
CIRCUIT ID ABCD
Input (V) 12 3 to 5.5 12 12
Output (V) -5 -12 -48 -72
Output (A) 2 0.4 0.1 0.1
CCOMP (µF) 0.047 0.22 0.1 0.068
CIN (µF) 3 x 10 3 x 22 10 10
COUT (µF) 2 x 100 2 x 47 39 39
CFB (pF) 390 1200 1000 1000
R1 (kΩ) (1%) 40.2 95.3 383 576
R2 (kΩ) (1%) 10 10 10 10
RCOMP (kΩ) 8.2 10 220 470
RCS (Ω) 0.02 0.02 0.05 0.05
RFREQ (kΩ) 150 150 150 150
D1 CMSH5-40 CMSH5-40 CMR1U-02 CMR1U-02
L1 (µH) 10 10 47 82
P1 FDS6685 FDS6375 IRFR5410 IRFR5410
CCOMP2 (pF) 220 220 22 12
Table 1. Component List for Main Application Circuits
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
______________________________________________________________________________________ 19
Chip Information
PROCESS: BiCMOS
QSOP
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
POL SYNC
IN
EXT
CS
PGND
GND
GND
N.C.
MAX1847
VL
FREQ
FB
COMP
REF
N.C.
1
++
2
3
4
5
10
9
8
7
6
IN
EXT
CS
PGNDREF
COMP
FREQ
VL
MAX1846
µMAX
TOP VIEW
GNDFB
SHDN
Package Information
For the latest package outline information and land patterns, go
to www.maxim-ic.com/packages. Note that a “+”, “#”, or “-” in
the package code indicates RoHS status only. Package draw-
ings may show a different suffix character, but the drawing per-
tains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE OUTLINE NO. LAND
PATTERN NO.
10 µMAX U10+2 21-0061 90-0330
16 QSOP E16+1 21-0055 90-0167
Pin Configurations
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
20
____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2010 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.
Revision History
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
2 9/10 Added equation in the Determining the Compensation Component Values section 16
