LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 LMV793/LMV794 88 MHz, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifiers Check for Samples: LMV793, LMV794 FEATURES DESCRIPTION 1 (Typical 5V Supply, Unless Otherwise Noted) 2 * * * * * * * * Input Referred Voltage Noise 5.8 nV/Hz Input Bias Current 100 fA Gain Bandwidth Product 88 MHz Supply Current per Channel - LMV793 1.15 mA - LMV794 1.30 mA Rail-to-Rail Output Swing - @ 10 k Load 25 mV from Rail - @ 2 k Load 45 mV from Rail Ensured 2.5V and 5.0V Performance Total Harmonic Distortion 0.04% @1 kHz, 600 Temperature Range -40C to 125C APPLICATIONS * * * * * * The LMV793 (single) and the LMV794 (dual) CMOS input operational amplifiers offer a low input voltage noise density of 5.8 nV/Hz while consuming only 1.15 mA (LMV793) of quiescent current. The LMV793/LMV794 are stable at a gain of 10 and have a gain bandwidth product (GBW) of 88 MHz. The LMV793/LMV794 have a supply voltage range of 1.8V to 5.5V and can operate from a single supply. The LMV793/LMV794 each feature a rail-to-rail output stage capable of driving a 600 load and sourcing as much as 60 mA of current. The LMV793/LMV794 provide optimal performance in low voltage and low noise systems. A CMOS input stage, with typical input bias currents in the range of a few femto-Amperes, and an input common mode voltage range, which includes ground, make the LMV793/LMV794 ideal for low power sensor applications where high speeds are needed. The LMV793/LMV794 are manufactured using TI's advanced VIP50 process. The LMV793 is offered in either a 5-Pin SOT23 or an 8-Pin SOIC package. The LMV794 is offered in either the 8-Pin SOIC or the 8Pin VSSOP. ADC Interface Photodiode Amplifiers Active Filters and Buffers Low Noise Signal Processing Medical Instrumentation Sensor Interface Applications Typical Application CF RF IIN CCM CD VB + + VOUT CIN = CD + CCM VOUT = - RF IIN VOLTAGE NOISE (nV/ Hz) 1000 5V 100 2.5V 10 1 0.1 1 10 100 1k 10k FREQUENCY (Hz) Figure 1. Photodiode Transimpedance Amplifier Figure 2. Input Referred Voltage Noise vs. Frequency 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright (c) 2007-2013, Texas Instruments Incorporated LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1) (2) Human Body Model ESD Tolerance (3) 2000V Machine Model 200V Charge-Device Model 1000V VIN Differential 0.3V Supply Voltage (V+ - V-) 6.0V Input/Output Pin Voltage V+ +0.3V, V- -0.3V -65C to 150C Storage Temperature Range Junction Temperature (4) +150C Soldering Information (1) (2) (3) (4) Infrared or Convection (20 sec) 235C Wave Soldering Lead Temp (10 sec) 260C Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test conditions, see the Electrical Characteristics Tables. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and specifications. Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC). The maximum power dissipation is a function of TJ(MAX), JA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/JA. All numbers apply for packages soldered directly onto a PC Board. Operating Ratings (1) Temperature Range (2) -40C to 125C Supply Voltage (V+ - V-) Package Thermal Resistance (JA (2)) (1) (2) -40C TA 125C 2.0V to 5.5V 0C TA 125C 1.8V to 5.5V 5-Pin SOT-23 180C/W 8-Pin SOIC 190C/W 8-Pin VSSOP 236C/W Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test conditions, see the Electrical Characteristics Tables. The maximum power dissipation is a function of TJ(MAX), JA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/JA. All numbers apply for packages soldered directly onto a PC Board. 2.5V Electrical Characteristics (1) Unless otherwise specified, all limits are specified for TA = 25C, V+ = 2.5V, V- = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol Parameter Conditions VOS Input Offset Voltage TC VOS Input Offset Voltage Temperature Drift (4) (1) (2) (3) (4) 2 Min (2) Typ Max Units 0.1 1.35 1.65 mV (3) LMV793 -1.0 LMV794 -1.8 (2) V/C Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. Limits are 100% production tested at 25C. Limits over the operating temperature range are specified through correlations using the statistical quality control (SQC) method. Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will also depend on the application and configuration. The typical values are not tested and are not ensured on shipped production material. Offset voltage average drift is determined by dividing the change in VOS by temperature change. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 2.5V Electrical Characteristics(1) (continued) Unless otherwise specified, all limits are specified for TA = 25C, V+ = 2.5V, V- = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol IB Parameter Input Bias Current Conditions VCM = 1.0V (5) (6) Min Typ Max -40C TA 85C 0.05 1 25 -40C TA 125C 0.05 1 100 (2) (3) IOS Input Offset Current VCM = 1.0V (6) CMRR Common Mode Rejection Ratio 0V VCM 1.4V 80 75 94 PSRR Power Supply Rejection Ratio 2.0V V+ 5.5V, VCM = 0V 80 75 100 1.8V V+ 5.5V, VCM = 0V 80 98 CMVR Common Mode Voltage Range CMRR 60 dB CMRR 55 dB AVOL Open Loop Voltage Gain VOUT = 0.15V to 2.2V, RL = 2 k to V+/2 Output Voltage Swing High Output Voltage Swing Low IOUT Output Current IS Supply Current Per Amplifier SR Slew Rate Units pA 10 fA dB dB -0.3 -0.3 1.5 1.5 LMV793 85 80 98 LMV794 82 78 92 88 84 110 VOUT = 0.15V to 2.2V, RL = 10 k to V+/2 VOUT (2) dB RL = 2 k to V+/2 25 75 82 RL = 10 k to V+/2 20 65 71 RL = 2 k to V+/2 30 75 78 RL = 10 k to V+/2 15 65 67 Sourcing to V- VIN = 200 mV (7) 35 28 47 Sinking to V+ VIN = -200 mV (7) 7 5 15 V mV from either rail mA LMV793 0.95 1.30 1.65 LMV794 1.1 1.50 1.85 AV = +10, Rising (10% to 90%) 32 AV = +10, Falling (90% to 10%) 24 mA V/s GBW Gain Bandwidth AV = +10, RL = 10 k 88 MHz en Input Referred Voltage Noise Density f = 1 kHz 6.2 nV/Hz in Input Referred Current Noise Density f = 1 kHz 0.01 pA/Hz THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 600 0.01 % (5) (6) (7) Positive current corresponds to current flowing into the device. This parameter is specified by design and/or characterization and is not tested in production. The short circuit test is a momentary test, the short circuit duration is 1.5 ms. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 3 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com 5V Electrical Characteristics (1) Unless otherwise specified, all limits are specified for TA = 25C, V+ = 5V, V- = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol VOS Parameter Conditions Min (2) Input Offset Voltage TC VOS IB Input Offset Voltage Temperature Drift Input Bias Current (4) Typ Max Units 0.1 1.35 1.65 mV (3) LMV793 -1.0 LMV794 -1.8 VCM = 2.0V (5) (6) 0.1 1 25 -40C TA 125C 0.1 1 100 Input Offset Current VCM = 2.0V (6) CMRR Common Mode Rejection Ratio 0V VCM 3.7V 80 75 100 PSRR Power Supply Rejection Ratio 2.0V V+ 5.5V, VCM = 0V 80 75 100 1.8V V+ 5.5V, VCM = 0V 80 98 10 Common Mode Voltage Range CMRR 60 dB CMRR 55 dB -0.3 -0.3 AVOL Open Loop Voltage Gain VOUT = 0.3V to 4.7V, LMV793 RL = 2 k to V+/2 85 80 97 LMV794 82 78 89 88 84 110 VOUT = 0.3V to 4.7V, RL = 10 k to V+/2 RL = 2 k to V+/2 Output Voltage Swing Low RL = 2 k to V+/2 IS (1) (2) (3) (4) (5) (6) (7) 4 Output Current Supply Current per Amplifier dB 4 4 75 82 LMV794 35 75 82 25 65 71 LMV793 42 75 78 LMV794 45 80 83 20 65 67 Sourcing to V- VIN = 200 mV (7) 45 37 60 Sinking to V+ VIN = -200 mV (7) 10 6 21 V dB 35 RL = 10 k to V+/2 IOUT fA LMV793 RL = 10 k to V+/2 pA dB CMVR Output Voltage Swing High V/C -40C TA 85C IOS VOUT (2) mV from either rail mA LMV793 1.15 1.40 1.75 LMV794 per Channel 1.30 1.70 2.05 mA Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. Limits are 100% production tested at 25C. Limits over the operating temperature range are specified through correlations using the statistical quality control (SQC) method. Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will also depend on the application and configuration. The typical values are not tested and are not ensured on shipped production material. Offset voltage average drift is determined by dividing the change in VOS by temperature change. Positive current corresponds to current flowing into the device. This parameter is specified by design and/or characterization and is not tested in production. The short circuit test is a momentary test, the short circuit duration is 1.5 ms. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 5V Electrical Characteristics(1) (continued) Unless otherwise specified, all limits are specified for TA = 25C, V+ = 5V, V- = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol Parameter SR Conditions Slew Rate Min Typ (2) (3) AV = +10, Rising (10% to 90%) 35 AV = +10, Falling (90% to 10%) 28 Max Units (2) V/s GBW Gain Bandwidth AV = +10, RL = 10 k 88 MHz en Input Referred Voltage Noise Density f = 1 kHz 5.8 nV/Hz in Input Referred Current Noise Density f = 1 kHz 0.01 pA/Hz THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 600 0.01 % Connection Diagram V - 5 1 + V N/C + 3 N/C 2 7 - + V -IN A +IN A +IN +IN 8 OUT A -IN 2 1 4 3 + 6 OUTPUT V -IN V Figure 3. 5-Pin SOT-23 (LMV793) Top View - 4 5 - 8 1 2 3 4 - OUTPUT 7 + +- 6 5 + V OUT B -IN B +IN B N/C Figure 4. 8-Pin SOIC (LMV793) Top View Figure 5. 8-Pin SOIC/VSSOP (LMV794) Top View Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 5 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com Typical Performance Characteristics Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. Supply Current vs. Supply Voltage (LMV793) Supply Current vs. Supply Voltage (LMV794) 2 2 1.6 1.6 SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) 125C 125C 25C 1.2 0.8 -40C 0.4 0 1.5 2.5 3.5 4.5 25C 1.2 -40C 0.8 0.4 0 1.5 5.5 6.0 2.5 3.5 + 4.5 5.5 6 + V (V) V (V) Figure 6. Figure 7. VOS vs. VCM 0.5 VOS vs. VCM 0.6 + V = 1.8V 0.55 0.45 + V = 2.5V 125C 0.5 0.4 0.45 0.3 VOS (mV) VOS (mV) -40C 0.35 25C 0.25 0.4 0.35 25C 0.3 0.25 0.2 0.15 0.1 -0.3 -40C 0.2 125C 0.15 0 0.3 0.6 0.9 0.1 -0.3 1.2 Figure 8. Figure 9. VOS vs. VCM VOS vs. Supply Voltage + 0.45 0.4 0.4 -40C 0.35 0.35 0.3 0.3 VOS (mV) VOS (mV) 1.8 0.5 V = 5V 0.45 25C 0.25 0.2 -40C 0.25 25C 0.2 0.15 0.15 125C 0.1 125C 0.1 0.05 0.05 0 0.6 1.5 2.4 3.3 4.2 1.5 2.5 3.5 4.5 5.5 6.0 + VCM (V) V (V) Figure 10. 6 1.1 VCM (V) 0.5 0 -0.3 0.4 VCM (V) Figure 11. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Typical Performance Characteristics (continued) Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. Slew Rate vs. Supply Voltage Input Bias Current vs. VCM 36 1.5 34 1 + V = 5V -40C 0.5 0 IBIAS (pA) SLEW RATE (V/Ps) RISING EDGE 32 30 28 -0.5 25C -1 -1.5 26 FALLING EDGE -2 24 -2.5 22 1.5 -3 2.5 3.5 4.5 5.5 0 1 2 + V (V) Figure 12. 4 Figure 13. Input Bias Current vs. VCM 50 Sourcing Current vs. Supply Voltage 80 + V = 5V 40 70 30 125C 60 20 125C ISOURCE (mA) IBIAS (pA) 3 VCM (V) 10 0 85C -10 -40C 50 25C 40 30 -20 20 -30 10 -40 0 -50 0 1 2 3 1 4 2 3 4 5 6 + VCM (V) V (V) Figure 14. Figure 15. Sinking Current vs. Supply Voltage Sourcing Current vs. Output Voltage 35 70 30 60 125C 125C 50 ISOURCE (mA) ISINK (mA) 25 25C 20 15 -40C 10 -40C 40 25C 30 20 5 10 0 0 1 2 3 4 5 6 0 + 1 2 3 V (V) VOUT (V) Figure 16. Figure 17. 4 5 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 7 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. Sinking Current vs. Output Voltage Positive Output Swing vs. Supply Voltage 40 30 RLOAD = 10 k: 35 VOUT FROM RAIL (mV) 125C 25 ISINK (mA) 20 25C 15 10 -40C 5 125C 25 25C 20 -40C 15 10 5 0 1.8 0 0 1 2 3 4 5 2.5 3.2 VOUT (V) V (V) Figure 18. Figure 19. 5.3 6 Positive Output Swing vs. Supply Voltage 25 50 45 VOUT FROM RAIL (mV) 20 15 125C 10 5 125C 40 25C 35 30 25 20 -40C 15 10 RLOAD = 10 k: 0 1.8 2.5 3.2 5 3.9 4.6 5.3 RLOAD = 2 k: 0 1.8 2.5 3.2 6 + V (V) 4.6 5.3 6 Figure 21. Negative Output Swing vs. Supply Voltage Positive Output Swing vs. Supply Voltage 100 50 -40C 45 90 25C VOUT FROM RAIL (mV) 40 35 125C 30 25 20 15 RLOAD = 600: 80 125C 70 60 25C 50 40 -40C 30 20 10 5 3.9 + V (V) Figure 20. VOUT FROM RAIL (mV) 4.6 Negative Output Swing vs. Supply Voltage -40C 10 RLOAD = 2 k: 0 1.8 2.5 3.2 3.9 4.6 5.3 6 0 1.8 + 2.5 3.2 3.9 4.6 5.3 6 + V (V) V (V) Figure 22. 8 3.9 + 25C VOUT FROM RAIL (mV) 30 Figure 23. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Typical Performance Characteristics (continued) Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. Negative Output Swing vs. Supply Voltage Time Domain Voltage Noise 120 VS = 2.5V 125C RLOAD = 600: VCM = 0.0V 25C 80 400 nV/DIV VOUT FROM RAIL (mV) 100 -40C 60 40 20 0 1.8 2.5 3.2 3.9 4.6 5.3 1S/DIV 6 + V (V) Figure 24. Figure 25. Input Referred Voltage Noise vs. Frequency Overshoot and Undershoot vs. CLOAD 70 OVERSHOOT AND UNDERSHOOT % VOLTAGE NOISE (nV/ Hz) 1000 5V 100 2.5V 10 1 0.1 1 100 10 1k US% 60 50 OS% 40 30 20 10 0 10k 20 0 40 FREQUENCY (Hz) 60 80 100 120 CLOAD (pF) Figure 26. Figure 27. THD+N vs. Frequency THD+N vs. Frequency 0.04 0.05 RL = 600: RL = 600: 0.04 THD+N (%) THD+N (%) 0.03 + 0.03 V = 2.5V 0.02 AV = +10 VO = 1 VPP 0.02 VO = 4 VPP 0.01 0.01 0 10 + V = 5V AV = +10 RL = 100 k: 100 1k RL = 100 k: 10k 100k 0 10 100 1k 10k 100k FREQUENCY (Hz) FREQUENCY (Hz) Figure 28. Figure 29. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 9 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. THD+N vs. Peak-to-Peak Output Voltage (VOUT) THD+N vs. Peak-to-Peak Output Voltage (VOUT) -40 -40 -50 THD+N (dB) THD+N (dB) -50 -60 RL = 600: -70 V+ = 2.5V f = 1 kHz AV = +10 -80 0.01 1 -70 -80 V+ = 5V f = 1 kHz AV = +10 -90 0.01 RL = 100 k: 0.1 RL = 600: -60 10 OUTPUT AMPLITUDE (VPP) RL = 100 k: 0.1 Figure 30. 10 Figure 31. Open Loop Gain and Phase 100 1 OUTPUT AMPLITUDE (VPP) Closed Loop Output Impedance vs. Frequency 1000 100 80 60 60 GAIN 40 40 20 20 + V = 5V CL = 20 pF RL = 2 k:, 10 k: 0 -20 100k OUTPUT IMPEDANCE (:) 80 PHASE () GAIN (dB) PHASE 100 10 1 0 1M 10M -20 100M 0.1 10k 100k FREQUENCY (Hz) 10M 100M Figure 33. Small Signal Transient Response, AV = +10 Large Signal Transient Response, AV = +10 10 mV/DIV 200 mV/DIV Figure 32. VIN = 100 mVPP VIN = 2 mVPP f = 1 MHz, AV = +10 f = 1 MHz, AV = +10 V = 5V, CL = 10 pF V = 2.5V, CL = 10 pF + + 10 1M FREQUENCY (Hz) 100 ns/DIV 100 ns/DIV Figure 34. Figure 35. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Typical Performance Characteristics (continued) Unless otherwise specified, TA = 25C, V- = 0, V+ = Supply Voltage = 5V, VCM = V+/2. Large Signal Transient Response, AV = +10 10 mV/DIV 200 mV/DIV Small Signal Transient Response, AV = +10 VIN = 100 mVPP VIN = 2 mVPP f = 1 MHz, AV = +10 f = 1 MHz, AV = +10 V = 2.5V, CL = 10 pF V = 5V, CL = 10 pF + + 100 ns/DIV 100 ns/DIV Figure 36. Figure 37. PSRR vs. Frequency CMRR vs. Frequency 120 100 -PSRR, 5V 90 + V = 2.5V +PSRR, 5V 80 +PSRR, 2.5V CMRR (dB) PSRR (dB) 80 100 -PSRR, 2.5V 70 60 + V = 5V 60 40 50 20 40 100 1k 10k 100k 0 10 1M 100 FREQUENCY (Hz) 1k 10k 100k 1M 10M FREQUENCY (Hz) Figure 38. Figure 39. Input Common Mode Capacitance vs. VCM 25 + V = 5V CCM (pF) 20 15 10 5 0 0 1 2 3 4 VCM (V) Figure 40. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 11 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com APPLICATION INFORMATION ADVANTAGES OF THE LMV793/LMV794 Wide Bandwidth at Low Supply Current The LMV793/LMV794 are high performance op amps that provide a GBW of 88 MHz with a gain of 10 while drawing a low supply current of 1.15 mA. This makes them ideal for providing wideband amplification in data acquisition applications. With the proper external compensation the LMV793/LMV794 can be operated at gains of 1 and still maintain much faster slew rates than comparable unity gain stable amplifiers. The increase in bandwidth and slew rate is obtained without any additional power consumption over the LMV796. Low Input Referred Noise and Low Input Bias Current The LMV793/LMV794 have a very low input referred voltage noise density (5.8 nV/Hz at 1 kHz). A CMOS input stage ensures a small input bias current (100 fA) and low input referred current noise (0.01 pA/Hz). This is very helpful in maintaining signal integrity, and makes the LMV793/LMV794 ideal for audio and sensor based applications. Low Supply Voltage The LMV793 and LMV794 have performance specified at 2.5V and 5V supply. These parts are specified to be operational at all supply voltages between 2.0V and 5.5V, for ambient temperatures ranging from -40C to 125C, thus utilizing the entire battery lifetime. The LMV793/LMV794 are also specified to be operational at 1.8V supply voltage, for temperatures between 0C and 125C optimizing their usage in low-voltage applications. RRO and Ground Sensing Rail-to-rail output swing provides the maximum possible dynamic range. This is particularly important when operating at low supply voltages. An innovative positive feedback scheme is used to boost the current drive capability of the output stage. This allows the LMV793/LMV794 to source more than 40 mA of current at 1.8V supply. This also limits the performance of these parts as comparators, and hence the usage of the LMV793 and the LMV794 in an open-loop configuration is not recommended. The input common-mode range includes the negative supply rail which allows direct sensing at ground in single supply operation. Small Size The small footprint of the LMV793 and the LMV794 package saves space on printed circuit boards, and enables the design of smaller electronic products, such as cellular phones, pagers, or other portable systems. Long traces between the signal source and the op amp make the signal path more susceptible to noise pick up. The physically smaller LMV793/LMV794 packages, allow the op amp to be placed closer to the signal source, thus reducing noise pickup and maintaining signal integrity. USING THE DECOMPENSATED LMV793 Advantages of Decompensated Op Amps A unity gain stable op amp, which is fully compensated, is designed to operate with good stability down to gains of 1. The large amount of compensation does provide an op amp that is relatively easy to use; however, a decompensated op amp is designed to maximize the bandwidth and slew rate without any additional power consumption. This can be very advantageous. The LMV793/LMV794 require a gain of 10 to be stable. However, with an external compensation network (a simple RC network) these parts can be stable with gains of 1 and still maintain the higher slew rate. Looking at the Bode plots for the LMV793 and its closest equivalent unity gain stable op amp, the LMV796, one can clearly see the increased bandwidth of the LMV793. Both plots are taken with a parallel combination of 20 pF and 10 k for the output load. 12 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 100 100 PHASE 80 80 GAIN (dB) GAIN 40 40 20 20 0 0 -20 1k 10k 100k PHASE () 60 60 1M 10M -20 100M FREQUENCY (Hz) 100 100 80 80 60 60 40 40 20 20 0 0 -20 1k 10k 100k 1M 10M PHASE () GAIN (dB) Figure 41. LMV793 AVOL vs. Frequency -20 100M FREQUENCY (Hz) Figure 42. LMV796 AVOL vs. Frequency Figure 41 shows the much larger 88 MHz bandwidth of the LMV793 as compared to the 17 MHz bandwidth of the LMV796 shown in Figure 42. The decompensated LMV793 has five times the bandwidth of the LMV796. What is a Decompensated Op Amp? The differences between the unity gain stable op amp and the decompensated op amp are shown in Figure 43. This Bode plot assumes an ideal two pole system. The dominant pole of the decompensated op amp is at a higher frequency, f1, as compared to the unity-gain stable op amp which is at fd as shown in Figure 43. This is done in order to increase the speed capability of the op amp while maintaining the same power dissipation of the unity gain stable op amp. The LMV793/LMV794 have a dominant pole at 1.6 kHz. The unity gain stable LMV796/LMV797 have their dominant pole at 300 Hz. DECOMPENSATED OP AMP AOL UNITY-GAIN STABLE OP AMP Gmin fGBWP fd f1 fu f2 fu' Figure 43. Open Loop Gain for Unity-Gain Stable Op Amp and Decompensated Op Amp Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 13 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com Having a higher frequency for the dominate pole will result in: 1. The DC open-loop gain (AVOL) extending to a higher frequency. 2. A wider closed loop bandwidth. 3. Better slew rate due to reduced compensation capacitance within the op amp. The second open loop pole (f2) for the LMV793/LMV794 occurs at 45 MHz. The unity gain (fu') occurs after the second pole at 51 MHz. An ideal two pole system would give a phase margin of 45 at the location of the second pole. The LMV793/LMV794 have parasitic poles close to the second pole, giving a phase margin closer to 0. Therefore it is necessary to operate the LMV793/LMV794 at a closed loop gain of 10 or higher, or to add external compensation in order to assure stability. For the LMV796, the gain bandwidth product occurs at 17 MHz. The curve is constant from fd to fu which occurs before the second pole. For the LMV793/LMV794, the GBW = 88 MHz and is constant between f1 and f2. The second pole at f2 occurs before AVOL = 1. Therefore fu' occurs at 51 MHz, well before the GBW frequency of 88 MHz. For decompensated op amps the unity gain frequency and the GBW are no longer equal. Gmin is the minimum gain for stability and for the LMV793/LMV794 this is a gain of 18 to 20 dB. Input Lead-Lag Compensation The recommended technique which allows the user to compensate the LMV793/LMV794 for stable operation at any gain is lead-lag compensation. The compensation components added to the circuit allow the user to shape the feedback function to make sure there is sufficient phase margin when the loop gain is as low as 0 dB and still maintain the advantages over the unity gain op amp. Figure 44 shows the lead-lag configuration. Only RC and C are added for the necessary compensation. RF RIN RC LMV793 C Figure 44. LMV793 with Lead-Lag Compensation for Inverting Configuration To cover how to calculate the compensation network values it is necessary to introduce the term called the feedback factor or F. The feedback factor F is the feedback voltage VA-VB across the op amp input terminals relative to the op amp output voltage VOUT. F= VA - V B VOUT (1) From feedback theory the classic form of the feedback equation for op amps is: VOUT A = 1 + AF VIN (2) A is the open loop gain of the amplifier and AF is the loop gain. Both are highly important in analyzing op amps. Normally AF >>1 and so the above equation reduces to: VOUT 1 = F VIN (3) Deriving the equations for the lead-lag compensation is beyond the scope of this datasheet. The derivation is based on the feedback equations that have just been covered. The inverse of feedback factor for the circuit in Figure 44 is: 14 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com (c) (c) F = 1 + RF 1 + s(Rc + RIN || RF) C 1 + sRcC RIN (c) (c) 1 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 (4) where 1/F's pole is located at 1 fp = 2SRcC (5) 1/F's zero is located at 1 fz = 2S Rc + RIN || RF)C (6) 1 F =1+ f=0 RF RIN (7) The circuit gain for Figure 44 at low frequencies is -RF/RIN, but F, the feedback factor is not equal to the circuit gain. The feedback factor is derived from feedback theory and is the same for both inverting and non-inverting configurations. Yes, the feedback factor at low frequencies is equal to the gain for the non-inverting configuration. f=f (c) (c) F = 1 + RIN || RF RF 1 + RC RIN (c) (c) 1 (8) From this formula, we can see that * 1/F's zero is located at a lower frequency compared with 1/F's pole. * 1/F's value at low frequency is 1 + RF/RIN. * This method creates one additional pole and one additional zero. * This pole-zero pair will serve two purposes: - To raise the 1/F value at higher frequencies prior to its intercept with A, the open loop gain curve, in order to meet the Gmin = 10 requirement. For the LMV793/LMV794 some overcompensation will be necessary for good stability. - To achieve the previous purpose above with no additional loop phase delay. Please note the constraint 1/F Gmin needs to be satisfied only in the vicinity where the open loop gain A and 1/F intersect; 1/F can be shaped elsewhere as needed. The 1/F pole must occur before the intersection with the open loop gain A. In order to have adequate phase margin, it is desirable to follow these two rules: 1. 1/F and the open loop gain A should intersect at the frequency where there is a minimum of 45 of phase margin. When over-compensation is required the intersection point of A and 1/F is set at a frequency where the phase margin is above 45, therefore increasing the stability of the circuit. 2. 1/F's pole should be set at least one decade below the intersection with the open loop gain A in order to take advantage of the full 90 of phase lead brought by 1/F's pole which is F's zero. This ensures that the effect of the zero is fully neutralized when the 1/F and A plots intersect each other. Calculating Lead-Lag Compensation for LMV793/LMV794 Figure 45 is the same plot as Figure 41, but the AVOL and phase curves have been redrawn as smooth lines to more readily show the concepts covered, and to clearly show the key parameters used in the calculations for lead-lag compensation. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 15 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com 100 PHASE ADDITIONAL COMPENSATION GAIN (dB) and PHASE (C) 80 AVOL 60 45 PHASE MARGIN 40 20 1 with ADDITIONAL F COMPENSATION 1 F 0 -20 1k 2nd POLE f2 10k 100k 1M GBP 10M 100M FREQUENCY (Hz) Figure 45. LMV793/LMV794 Simplified Bode Plot To obtain stable operation with gains under 10 V/V the open loop gain margin must be reduced at high frequencies to where there is a 45 phase margin when the gain margin of the circuit with the external compensation is 0 dB. The pole and zero in F, the feedback factor, control the gain margin at the higher frequencies. The distance between F and AVOL is the gain margin; therefore, the unity gain point (0 dB) is where F crosses the AVOL curve. For the example being used RIN = RF for a gain of -1. Therefore F = 6 dB at low frequencies. At the higher frequencies the minimum value for F is 18 dB for 45 phase margin. From Equation 8 we have the following relationship: (c) + (c) 1 (c) RIN || RF RF = 18 dB = 7.9 1+ RC RIN (c) (9) Now set RF = RIN = R. With these values and solving for RC we have RC = R/5.9. Note that the value of C does not affect the ratio between the resistors. Once the value of the resistors are set, then the position of the pole in F must be set. A 2 k resistor is used for RF and RIN in this design. Therefore the value for RC is set at 330, the closest standard value for 2 k/5.9. Rewriting Equation 5 to solve for the minimum capacitor value gives the following equation: C = 1/(2fpRC) (10) The feedback factor curve, F, intersects the AVOL curve at about 12 MHz. Therefore the pole of F should not be any larger than 1.2 MHz. Using this value and Rc = 330 the minimum value for C is 390 pF. Figure 46 shows that there is too much overshoot, but the part is stable. Increasing C to 2.2 nF did not improve the ringing, as shown in Figure 47. Figure 46. First Try at Compensation, Gain = -1 16 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Figure 47. C Increased to 2.2 nF, Gain = -1 Some over-compensation appears to be needed for the desired overshoot characteristics. Instead of intersecting the AVOL curve at 18 dB, 2 dB of over-compensation will be used, and the AVOL curve will be intersected at 20 dB. Using Equation 8 for 20 dB, or 10 V/V, the closest standard value of RC is 240. The following two waveforms show the new resistor value with C = 390 pF and 2.2 nF. Figure 49 shows the final compensation and a very good response for the 1 MHz square wave. Figure 48. RC = 240 and C = 390 pF, Gain = -1 Figure 49. RC = 240 and C = 2.2 nF, Gain = -1 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 17 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com To summarize, the following steps were taken to compensate the LMV793 for a gain of -1: 1. Values for Rc and C were calculated from the Bode plot to give an expected phase margin of 45. The values were based on RIN = RF = 2 k. These calculations gave Rc = 330 and C = 390 pF. 2. To reduce the ringing C was increased to 2.2 nF which moved the pole of F, the feedback factor, farther away from the AVOL curve. 3. There was still too much ringing so 2 dB of over-compensation was added to F. This was done by decreasing RC to 240. The LMV796 is the fully compensated part which is comparable to the LMV793. Using the LMV796 in the same setup, but removing the compensation network, provide the response shown in Figure 50 . Figure 50. LMV796 Response For large signal response the rise and fall times are dominated by the slew rate of the op amps. Even though both parts are quite similar the LMV793 will give rise and fall times about 2.5 times faster than the LMV796. This is possible because the LMV793 is a decompensated op amp and even though it is being used at a gain of -1, the speed is preserved by using a good technique for external compensation. Non-Inverting Compensation For the non-inverting amp the same theory applies for establishing the needed compensation. When setting the inverting configuration for a gain of -1, F has a value of 2. For the non-inverting configuration both F and the actual gain are the same, making the non-inverting configuration more difficult to compensate. Using the same circuit as shown in Figure 44, but setting up the circuit for non-inverting operation (gain of +2) results in similar performance as the inverting configuration with the inputs set to half the amplitude to compensate for the additional gain. Figure 51 below shows the results. Figure 51. RC = 240 and C = 2.2 nF, Gain = +2 18 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Figure 52. LMV796 Response Gain = +2 The response shown in Figure 51 is close to the response shown in Figure 49. The part is actually slightly faster in the non-inverting configuration. Decreasing the value of RC to around 200 can decrease the negative overshoot but will have slightly longer rise and fall times. The other option is to add a small resistor in series with the input signal. Figure 52 shows the performance of the LMV796 with no compensation. Again the decompensated parts are almost 2.5 times faster than the fully compensated op amp. The most difficult op amp configuration to stabilize is the gain of +1. With proper compensation the LMV793/LMV794 can be used in this configuration and still maintain higher speeds than the fully compensated parts. Figure 53 shows the gain = 1, or the buffer configuration, for these parts. RF RC RP LMV793 C + Figure 53. LMV793 with Lead-Lag Compensation for Non-Inverting Configuration Figure 53 is the result of using Equation 8 and additional experimentation in the lab. RP is not part of Equation 8, but it is necessary to introduce another pole at the input stage for good performance at gain = +1. Equation 8 is shown below with RIN = . + (c) 1 (c) RF = 18 dB = 7.9 Rc (11) Using 2 k for RF and solving for RC gives RC = 2000/6.9 = 290. The closest standard value for RC is 300. After some fine tuning in the lab RC = 330 and RP = 1.5 k were choosen as the optimum values. RP together with the input capacitance at the non-inverting pin inserts another pole into the compensation for the LMV793/LMV794. Adding this pole and slightly reducing the compensation for 1/F (using a slightly higher resistor value for RC) gives the optimum response for a gain of +1. Figure 54 is the response of the circuit shown in Figure 53. Figure 55 shows the response of the LMV796 in the buffer configuration with no compensation and RP = RF = 0. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 19 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com Figure 54. RC = 330 and C = 10 nF, Gain = +1 Figure 55. LMV796 Response Gain = +1 With no increase in power consumption the decompensated op amp offers faster speed over the compensated equivalent part. These examples used RF = 2 k. This value is high enough to be easily driven by the LMV793/LMV794, yet small enough to minimize the effects from the parasitic capacitance of both the PCB and the op amp. Note: When using the LMV793/LMV794, proper high frequency PCB layout must be followed. The GBW of these parts is 88 MHz, making the PCB layout significantly more critical than when using the compensated counterparts which have a GBW of 17 MHz. TRANSIMPEDANCE AMPLIFIER An excellent application for either the LMV793 or the LMV794 is as a transimpedance amplifier. With a GBW product of 88 MHz these parts are ideal for high speed data transmission by light. The circuit shown on the front page of the datasheet is the circuit used to test the LMV793/LMV794 as transimpedance amplifiers. The only change is that VB is tied to the VCC of the part, thus the direction of the diode is reversed from the circuit shown on the front page. Very high speed components were used in testing to check the limits of the LMV793/LMV794 in a transimpedance configuration. The photo diode part number is PIN-HR040 from OSI Optoelectronics. The diode capacitance for this part is only about 7 pF for the 2.5V bias used (VCC to virtual ground). The rise time for this diode is 1 nsec. A laser diode was used for the light source. Laser diodes have on and off times under 5 nsec. The speed of the selected optical components allowed an accurate evaluation of the LMV793 as a transimpedance amplifier. TIs Evaluation Board for decompensated op amps, PN 551013271-001 A, was used and only minor modifications were necessary and no traces had to be cut. 20 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 CF 2.5V 2.5V RF DPHOTO CD LMV793 CCM VOUT + -2.5V Figure 56. Transimpedance Amplifier Figure 56 is the complete schematic for a transimpedance amplifier. Only the supply bypass capacitors are not shown. CD represents the photo diode capacitance which is given on its datasheet. CCM is the input common mode capacitance of the op amp and, for the LMV793 it is shown in the last drawing of the Typical Performance Characteristics section of this datasheet. In Figure 56 the inverting input pin of the LMV793 is kept at virtual ground. Even though the diode is connected to the 2.5V line, a power supply line is AC ground, thus CD is connected to ground. Figure 57 shows the schematic needed to derive F, the feedback factor, for a transimpedance amplifier. In this figure CD + CCM = CIN. Therefore it is critical that the designer knows the diode capacitance and the op amp input capacitance. The photo diode is close to an ideal current source once its capacitance is included in the model. What kind of circuit is this? Without CF there is only an input capacitor and a feedback resistor. This circuit is a differentiator! Remember, differentiator circuits are inherently unstable and must be compensated. In this case CF compensates the circuit. CF RF VA IDIODE CIN LMV793 VOUT + Figure 57. Transimpedance Feedback Model Using feedback theory, F = VA/VOUT, this becomes a voltage divider giving the following equation: F= 1 + sCFRF 1 + sRF (CF + CIN) (12) The noise gain is 1/F. Because this is a differentiator circuit, a zero must be inserted. The location of the zero is given by: 1 z = 1 + sRF (CF + CIN) (13) CF has been added for stability. The addition of this part adds a pole to the circuit. The pole is located at: p = 1 1 + sCFRF (14) To attain maximum bandwidth and still have good stability the pole is to be located on the open loop gain curve which is A. If additional compensation is required one can always increase the value of CF, but this will also reduce the bandwidth of the circuit. Therefore A = 1/F, or AF = 1. For A the equation is: Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 21 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 A= ZGBW Z = www.ti.com GBW (15) The expression fGBW is the gain bandwidth product of the part. For a unity gain stable part this is the frequency where A = 1. For the LMV793 fGBW = 88 MHz. Multiplying A and F results in the following equation: GBW x 1 + sCFRF 1 + sRF (CF + CIN) CFRF (c) CFRF 1+ x 1+ (c) = 2 =1 RF (CF + CIN) CFRF (c) P GBW (c) AF = 2 (16) For the above equation s = j. To find the actual amplitude of the equation the square root of the square of the real and imaginary parts are calculated. At the intersection of F and A, we have: Z= 1 CFRF (17) After a bit of algebraic manipulation the above equation reduces to: (c) 1+ C + C IN F C F (c) 2 2 = 8S2 GBW RF CF 2 2 (18) In the above equation the only unknown is CF. In trying to solve this equation the fourth power of CF must be dealt with. An excel spread sheet with this equation can be used and all the known values entered. Then through iteration, the value of CF when both sides are equal will be found. That is the correct value for CF, and of course the closest standard value is used for CF. Before moving the lab, the transfer function of the transimpedance amplifier must be found and the units must be in Ohms. -RF x IDIODE VOUT = 1 + sCFRF (19) The LMV793 was evaluated for RF = 10 k and 100 k, representing a somewhat lower gain configuration and with the 100 k feedback resistor a fairly high gain configuration. The RF = 10 k is covered first. Looking at the Figure 39 chart for CCM for the operating point selected CCM = 15 pF. Note that for split supplies VCM = 2.5V, CIN = 22 pF and fGBW = 88 MHz. Solving for CF the calculated value is 1.75 pF, so 1.8 pF is selected for use. Checking the frequency of the pole finds that it is at 8.8 MHz, which is right at the minimum gain recommended for this part. Some over compensation was necessary for stability and the final selected value for CF is 2.7 pF. This moves the pole to 5.9 MHz. Figure 58 and Figure 59 show the rise and fall times obtained in the lab with a 1V output swing. The laser diode was difficult to drive due to thermal effects making the starting and ending point of the pulse quite different, therefore the two separate scope pictures. Figure 58. Fall Time 22 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 LMV793, LMV794 www.ti.com SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 Figure 59. Rise Time In Figure 58 the ringing and the hump during the on time is from the laser. The higher drive levels for the laser gave ringing in the light source as well as light changing from the thermal characteristics. The hump is due to the thermal characteristics. Solving for CF using a 100 k feedback resistor, the calculated value is 0.54 pF. One of the problems with more gain is the very small value for CF. A 0.5 pF capacitor was used, its measured value being 0.64 pF. For the 0.64 pF location the pole is at 2.5 MHz. Figure 60 shows the response for a 1V output. Figure 60. High Gain Response A transimpedance amplifier is an excellent application for the LMV793. Even with the high gain using a 100 k feedback resistor, the bandwidth is still well over 1 MHz. Other than a little over compensation for the 10 k feedback resistor configuration using the LMV793 was quite easy. Of course a very good board layout was also used for this test. For information on photo diodes please contact OSI Optoelectronics, (310) 978-0516. For further information on transimpedance amplifiers please contact your Texas Instruments representative. Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 23 LMV793, LMV794 SNOSAX6D - MARCH 2007 - REVISED MARCH 2013 www.ti.com REVISION HISTORY Changes from Revision C (March 2013) to Revision D * 24 Page Changed layout of National Data Sheet to TI format .......................................................................................................... 23 Submit Documentation Feedback Copyright (c) 2007-2013, Texas Instruments Incorporated Product Folder Links: LMV793 LMV794 PACKAGE OPTION ADDENDUM www.ti.com 12-Oct-2017 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (C) Device Marking (4/5) LMV793MA/NOPB ACTIVE SOIC D 8 95 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LMV7 93MA LMV793MAX/NOPB ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LMV7 93MA LMV793MF/NOPB ACTIVE SOT-23 DBV 5 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AS4A LMV793MFX/NOPB ACTIVE SOT-23 DBV 5 3000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AS4A LMV794MA/NOPB ACTIVE SOIC D 8 95 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LMV7 94MA LMV794MAX/NOPB ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LMV7 94MA LMV794MM/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AN4A LMV794MMX/NOPB ACTIVE VSSOP DGK 8 3500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 AN4A (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. 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Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 20-Dec-2016 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LMV793MAX/NOPB SOIC D 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1 LMV793MF/NOPB SOT-23 DBV 5 1000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3 LMV793MFX/NOPB SOT-23 DBV 5 3000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3 LMV794MAX/NOPB SOIC D 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1 LMV794MM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LMV794MMX/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 20-Dec-2016 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LMV793MAX/NOPB SOIC D 8 2500 367.0 367.0 35.0 LMV793MF/NOPB SOT-23 DBV 5 1000 210.0 185.0 35.0 LMV793MFX/NOPB SOT-23 DBV 5 3000 210.0 185.0 35.0 LMV794MAX/NOPB SOIC D 8 2500 367.0 367.0 35.0 LMV794MM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0 LMV794MMX/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0 Pack Materials-Page 2 PACKAGE OUTLINE DBV0005A SOT-23 - 1.45 mm max height SCALE 4.000 SMALL OUTLINE TRANSISTOR C 3.0 2.6 1.75 1.45 PIN 1 INDEX AREA 1 0.1 C B A 5 2X 0.95 1.9 1.45 MAX 3.05 2.75 1.9 2 4 0.5 5X 0.3 0.2 3 (1.1) C A B 0.15 TYP 0.00 0.25 GAGE PLANE 8 TYP 0 0.22 TYP 0.08 0.6 TYP 0.3 SEATING PLANE 4214839/C 04/2017 NOTES: 1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing per ASME Y14.5M. 2. This drawing is subject to change without notice. 3. Refernce JEDEC MO-178. www.ti.com EXAMPLE BOARD LAYOUT DBV0005A SOT-23 - 1.45 mm max height SMALL OUTLINE TRANSISTOR PKG 5X (1.1) 1 5 5X (0.6) SYMM (1.9) 2 2X (0.95) 3 4 (R0.05) TYP (2.6) LAND PATTERN EXAMPLE EXPOSED METAL SHOWN SCALE:15X SOLDER MASK OPENING METAL SOLDER MASK OPENING METAL UNDER SOLDER MASK EXPOSED METAL EXPOSED METAL 0.07 MIN ARROUND 0.07 MAX ARROUND NON SOLDER MASK DEFINED (PREFERRED) SOLDER MASK DEFINED SOLDER MASK DETAILS 4214839/C 04/2017 NOTES: (continued) 4. Publication IPC-7351 may have alternate designs. 5. Solder mask tolerances between and around signal pads can vary based on board fabrication site. www.ti.com EXAMPLE STENCIL DESIGN DBV0005A SOT-23 - 1.45 mm max height SMALL OUTLINE TRANSISTOR PKG 5X (1.1) 1 5 5X (0.6) SYMM (1.9) 2 2X(0.95) 4 3 (R0.05) TYP (2.6) SOLDER PASTE EXAMPLE BASED ON 0.125 mm THICK STENCIL SCALE:15X 4214839/C 04/2017 NOTES: (continued) 6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate design recommendations. 7. Board assembly site may have different recommendations for stencil design. www.ti.com PACKAGE OUTLINE DBV0005A SOT-23 - 1.45 mm max height SCALE 4.000 SMALL OUTLINE TRANSISTOR C 3.0 2.6 1.75 1.45 PIN 1 INDEX AREA 1 0.1 C B A 5 2X 0.95 1.9 1.45 MAX 3.05 2.75 1.9 2 4 0.5 5X 0.3 0.2 3 (1.1) C A B 0.15 TYP 0.00 0.25 GAGE PLANE 8 TYP 0 0.22 TYP 0.08 0.6 TYP 0.3 SEATING PLANE 4214839/C 04/2017 NOTES: 1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing per ASME Y14.5M. 2. This drawing is subject to change without notice. 3. Refernce JEDEC MO-178. www.ti.com EXAMPLE BOARD LAYOUT DBV0005A SOT-23 - 1.45 mm max height SMALL OUTLINE TRANSISTOR PKG 5X (1.1) 1 5 5X (0.6) SYMM (1.9) 2 2X (0.95) 3 4 (R0.05) TYP (2.6) LAND PATTERN EXAMPLE EXPOSED METAL SHOWN SCALE:15X SOLDER MASK OPENING METAL SOLDER MASK OPENING METAL UNDER SOLDER MASK EXPOSED METAL EXPOSED METAL 0.07 MIN ARROUND 0.07 MAX ARROUND NON SOLDER MASK DEFINED (PREFERRED) SOLDER MASK DEFINED SOLDER MASK DETAILS 4214839/C 04/2017 NOTES: (continued) 4. Publication IPC-7351 may have alternate designs. 5. Solder mask tolerances between and around signal pads can vary based on board fabrication site. www.ti.com EXAMPLE STENCIL DESIGN DBV0005A SOT-23 - 1.45 mm max height SMALL OUTLINE TRANSISTOR PKG 5X (1.1) 1 5 5X (0.6) SYMM (1.9) 2 2X(0.95) 4 3 (R0.05) TYP (2.6) SOLDER PASTE EXAMPLE BASED ON 0.125 mm THICK STENCIL SCALE:15X 4214839/C 04/2017 NOTES: (continued) 6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate design recommendations. 7. Board assembly site may have different recommendations for stencil design. www.ti.com IMPORTANT NOTICE Texas Instruments Incorporated (TI) reserves the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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