MIC2208
3mm x 3mm 1MHz 3A PWM Buck
Regulator
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
September 2007 M9999-090707-C
General Description
The Micrel MIC2208 is a high efficiency PWM buck (step-
down) regulator that provides up to 3A of output current.
The MIC2208 operates at 1MHz and has external
compensation that allows a closed loop bandwidth of over
100kHz.
The low on-resistance internal p-channel MOSFET of the
MIC2208 allows efficiencies over 94% reduces external
component count and eliminates the need for an
expensive current sense resistor.
The MIC2208 operates from 2.7V to 5.5V input and the
output can be adjusted down to 1V. The devices can
operate with a maximum duty cycle of 100% for use in low-
dropout conditions.
The MIC2208 is available in the exposed pad 12-pin
3mm x 3mm MLF® package with a junction operating
range from –40°C to +125°C.
Datasheets and support documentation can be found on
Micrel’s web site at: www.micrel.com.
Features
• 2.7 to 5.5V supply voltage
• 1MHz PWM mode
• Output current to 3A
• >90% efficiency
• Adjustable output voltage option down to 1V
• Ultra-fast transient response
• External Compensation
• Stable with a wide range of output capacitance
• Fully integrated 5A MOSFET switch
• Micropower shutdown
• Thermal shutdown and current limit protection
• Pb-free 12-pin 3mm x 3mm MLF® package
• –40°C to +125°C junction temperature range
Applications
• 5V or 3.3V Point of Load Conversion
• Telecom/Networking Equipment
• Set Top Boxes
• Storage Equipment
• Video Cards
___________________________________________________________________________________________________________
Typical Application
MIC2208
3A 1MHz Buck Regulator
Micrel, Inc. MIC2208
September 2007 2 M9999-090707-C
Ordering Information
Part Number Voltage Temperature Range Package Lead Finish
MIC2208YML Adj. –40° to +125°C 12-Pin 3x3 MLF® Pb-Free
Note:
MLF® is GREEN RoHS compliant package. Lead finish is NiPdAu. Mold compound is Halogen Free.
Pin Configur ation
BIAS EN
SW
VIN
PGND
SGND
SW
VIN
PGND
PGOOD
5
1
2
3
4
8
FB COMP
67
12
11
10
9
EP
12-Pin 3mm x 3mm MLF® (ML)
Pin Description
Pin Number Pin Name Pin Function
1, 12 SW Switch (Output): Internal power P-Channel MOSFET output switch.
2, 11 VIN Supply Voltage (Input): Supply voltage for the source of the internal P-channel
MOSFET and driver. Requires bypass capacitor to GND
3, 10 PGND Power Ground. Provides the ground return path for the high-side drive current.
4 SGND Signal Ground. Provides return path for control circuitry and internal reference.
5 BIAS
Internal circuit bias supply. Must be bypassed with a 0.1µF ceramic capacitor to
SGND.
6 FB
Feedback. Input to the error amplifier, connect to the external resistor divider
network to set the output voltage.
7 COMP
Compensation. This is the internal error amplifier output. Connect external
compensation components for type II or type III compensation.
8 EN
Enable (Input). Logic level low will shutdown the device, reducing the current
draw to less than 5µA.
9 PGOOD
Power Good. Open drain output that is pulled to ground when the output voltage
is within ±7.5% of the set regulation voltage
EP GND Connect to ground.
Micrel, Inc. MIC2208
September 2007 3 M9999-090707-C
Absolute Maximum Ratings(1)
Supply Voltage (VIN)........................................ –0.3V to +6V
Output Switch Voltage (VSW) .............................. –1V to +6V
Output Switch Current (ISW)............................................10A
Logic Input Voltage (VEN) .................................. –0.3V to VIN
Storage Temperature (Ts) ...........................–60°C to 150°C
ESD Rating(3).......................................................2kV (HBM)
Operating Ratings(2)
Supply Voltage (VIN)..................................... +2.7V to +5.5V
Logic Input Voltage (VEN, VLOWQ)............................ 0V to VIN
Junction Temperature (TJ) ........................ –40°C to +125°C
Junction Thermal Resistance
3x3 MLF-12 (θJA) ...............................................60°C/W
Electrical Characteristics(4)
VIN = VEN = 3.6V; L = 1µH; COUT = 4.7µF; TA = 25°C, unless noted. Bold values indicate –40°C< TJ < +125°C.
Parameter Condition Min Typ Max Units
Supply Voltage Range 2.7 5.5 V
Under-Voltage Lockout
Threshold
(turn-on) 2.45 2.55 2.65 V
UVLO Hysteresis 100 mV
Quiescent Current VFB = 0.9 * VNOM (not switching) 720 950 µA
Shutdown Current VEN = 0V 0.1 5 µA
[Adjustable] Feedback
Voltage
± 1%
± 2% (over temperature)
0.99
0.98
1 1.01
1.02
V
V
FB pin input current 1 100 nA
Current Limit in PWM Mode VFB = 0.9 * VNOM 8 10 A
Output Voltage Line
Regulation
VOUT > 2.2V; VIN = VOUT + 500mV to 5.5V; ILOAD = 20mA
VOUT < 2.2V; VIN = 2.7V to 5.5V; ILOAD = 20mA
0.13 %
Output Voltage Load
Regulation
20mA < ILOAD < 3A 0.2 1 %
PWM Switch ON-
Resistance
ISW = 50mA VFB = 0.7VFB_NOM (High Side Switch) 95 20
300
m
m
Oscillator Frequency 0.9 1 1.1 MHz
Enable Threshold 0.5 0.85 1.3 V
Enable Input Current 0.1 2 µA
Soft Start Time VOUT =10% to VOUT = 90% 450 µs
Over-Temperature
Shutdown
160 °C
over-Temperature
Hysteresis
20 °C
Power Good Range ±7 ±10 %
Power Good Resistance IPGOOD 145
200
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
Micrel, Inc. MIC2208
September 2007 4 M9999-090707-C
Typical Characteristics
80
82
84
86
88
90
92
94
96
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
3.3VOUT Efficiency
4.5VIN
5VIN 5.5VIN
75
77
79
81
83
85
87
89
91
93
95
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.8VOUT Efficiency
3.6VIN
3VIN
3.3VIN
70
72
74
76
78
80
82
84
86
88
90
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EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.8VOUT Efficiency
4.5VIN
5VIN
70
75
80
85
90
95
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.5VOUT Efficiency
3.3VIN
3VIN
3.6VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.5VOUT Efficiency
5.5VIN
5VIN
4.5VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.2VOUT Efficiency
3.6VIN
3VIN
3.3VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2208
1.2VOUT Efficiency
5.5VIN
5VIN
4.5VIN
0.990
0.992
0.994
0.996
0.998
1.000
1.002
1.004
1.006
1.008
1.010
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OUTPUT VOLTAGE (V)
OUTPUT CURRENT (A)
Load Regul ati on
3.3VIN
0.9900
0.9920
0.9940
0.9960
0.9980
1.0000
1.0020
1.0040
1.0060
1.0080
1.0100
-40
-20
0
20
40
60
80
100
120
FEEDBACK VOLTAGE (V)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
3.3VIN
0.80
0.85
0.90
0.95
1.00
1.05
1.10
1.15
1.20
-40
-20
0
20
40
60
80
100
120
FEEDBACK VOLTAGE (V)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
3.3VIN
Micrel, Inc. MIC2208
September 2007 5 M9999-090707-C
Typical Characteristics (c ontinued)
0
0.2
0.4
0.6
0.8
1.0
1.2
012345
FEEDBACK VOLTAGE (V)
SUPPLY VOLTAGE (V)
Feedback Voltage
vs. Supply Voltage
0
100
200
300
400
500
600
700
012345
QUIESCENT CURRENT (µA)
SUPPLY VOLTAGE (V)
Quiescent Current
vs. Supply Voltage
70
75
80
85
90
95
100
105
110
115
120
2.7 3.2 3.7 4.2 4.7 5.2
P-CHANNEL RDSON (mOhms)
SUPPLY VOLTAGE (V)
RDSON
vs. Supply Voltage
0
20
40
60
80
100
120
140
160
-40
-20
0
20
40
60
80
100
120
P-CHANNEL RDSON (mOhms)
TEMPERATURE (°C)
RDSON
vs. Temperature
3.3VIN
0
0.2
0.4
0.6
0.8
1
1.2
2.7 3.2 3.7 4.2 4.7
ENABLE THRESHOLD (V)
SUPPLY VOLTAGE (V)
Enable Threshold
vs. Supply Voltage
0
0.2
0.4
0.6
0.8
1
1.2
-40
-20
0
20
40
60
80
100
120
ENABLE THRESHOLD (V)
TEMPERATURE (°C)
Enable Threshold
vs. Temperature
0
0.5
1
1.5
2
2.5
3
3.5
60 70 80 90 100 110 120
MAX. OUTPUT CURRENT (A)
AMBIENT TEMPERATURE (°C)
Max. Continuous Current
vs. Ambient Temp 3.3VOUT*
*Using recommended
layout (1oz copper)
and B.O.M.
5VIN
0
0.5
1
1.5
2
2.5
3
3.5
60 70 80 90 100 110 120
MAX. OUTPUT CURRENT (A)
AMBIENT TEMPERATURE (°C)
Max. Continuous Current
vs. Ambient Temp 2.5VOUT*
*Using
recommended
layout (1oz copper)
and B.O.M.
3.3VIN 5VIN
0
0.5
1
1.5
2
2.5
3
3.5
60 70 80 90 100 110 120
MAX. OUTPUT CURRENT (A)
AMBIENT TEMPERATURE (°C)
Max. Continuous Current
vs. Ambient Temp 1.8VOUT*
*Using
recommended
layout (1oz copper)
and B.O.M.
3.3VIN 5VIN
0
0.5
1
1.5
2
2.5
3
3.5
60 70 80 90 100 110 120
MAX. OUTPUT CURRENT (A)
AMBIENT TEMPERATURE (°C)
Max. Continuous Current
vs. Ambient Temp 1.0VOUT*
*Using
recommended
layout (1oz copper)
and B.O.M.
3.3VIN 5VIN
Micrel, Inc. MIC2208
September 2007 6 M9999-090707-C
Functional Characteristics
Micrel, Inc. MIC2208
September 2007 7 M9999-090707-C
Functional Diagram
VIN
VIN
BIAS
EN
SW
SW
FB
PGOOD
PGND
Enable and
Control Logic
PWM
Control
P-Channel
Current Limit
SGND
1.0V
1.0V
Soft
Start
Bias,
UVLO,
Thermal
Shutdown
HSD
EA
COMP
MIC2208 Block Diagram
Micrel, Inc. MIC2208
September 2007 8 M9999-090707-C
Pin Description
VIN
Two pins for VIN provide power to the source of the
internal P-channel MOSFET along with the current
limiting sensing. The VIN operating voltage range is from
2.7V to 5.5V. Due to the high switching speeds, a 10µF
capacitor is recommended close to VIN and the power
ground (PGND) for each pin for bypassing. Please refer
to layout recommendations.
BIAS
The bias (BIAS) provides power to the internal reference
and control sections of the MIC2208. A 10 resistor
from VIN to BIAS and a 0.1µF from BIAS to SGND is
required for clean operation.
EN
The enable pin provides a logic level control of the
output. In the off state, supply current of the device is
greatly reduced (typically <1µA). Do not drive the enable
pin above the supply voltage.
FB
The feedback pin (FB) provides the control path to
control the output. For adjustable versions, a resistor
divider connecting the feedback to the output is used to
adjust the desired output voltage. The output voltage is
calculated as follows:
⎟
⎠
⎞
⎜
⎝
⎛+×= 1
R2
R1
VV REFOUT
where VREF is equal to 1.0V.
COMP
The COMP pin is the output of the internal error
amplifier. This pin is used to compensate the MIC2208
for stability over a varying range of external components.
Refer to the compensation section of the datasheet for
determining necessary component values.
SW
The switch (SW) pin connects directly to the inductor
and provides the switching current necessary to operate
in PWM mode. Due to the high speed switching on this
pin, the switch node should be routed away from
sensitive nodes. This pin also connects to the cathode of
the free-wheeling diode.
PGOOD
Power good is an open drain pull down that indicates
when the output voltage has reached regulation. For a
power good low, the output voltage is within ±10% of the
set regulation voltage. For output voltages greater or
less than 10%, the PGOOD pin is high. This should be
connected to the input supply through a pull up resistor.
A delay can be added by placing a capacitor from
PGOOD to ground.
PGND
Power ground (PGND) is the ground path for the
MOSFET drive current. The current loop for the power
ground should be as small as possible and separate
from the Analog ground (AGND) loop. Refer to the layout
considerations fro more details.
SGND
Signal ground (SGND) is the ground path for the biasing
and control circuitry. The current loop for the signal
ground should be separate from the power ground
(PGND) loop. Refer to the layout considerations for more
details.
Micrel, Inc. MIC2208
September 2007 9 M9999-090707-C
Application Information
The MIC2208 is a 3A PWM non-synchronous buck
regulator. By switching an input voltage supply, and
filtering the switched voltage through an Inductor and
capacitor, a regulated DC voltage is obtained. Figure 1
shows a simplified example of a non-synchronous buck
converter.
Figure 1.
For a non-synchronous buck converter, there are two
modes of operation; continuous and discontinuous.
Continuous or discontinuous refer to the inductor
current. If current is continuously flowing through the
inductor throughout the switching cycle, it is in
continuous operation. If the inductor current drops to
zero during the off time, it is in discontinuous operation.
Critically continuous is the point where any decrease in
output current will cause it to enter discontinuous
operation. The critically continuous load current can be
calculated as follows:
L21MHz
V
V
V
IIN
2
OUT
OUT
OUT ××
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡−
=
Continuous or discontinuous operation determines how
we calculate peak inductor current.
Continuous Operation
Figure 2 illustrates the switch voltage and inductor
current during continuous operation.
Figure 2. Continuous Operation
The output voltage is regulated by pulse width
modulating (PWM) the switch voltage to the average
required output voltage. The switching can be broken up
into two cycles; On and Off.
During the on-time, the high side switch is turned on,
current flows from the input supply through the inductor
and to the output. The inductor current is
Figure 3. On-Time
charged at the rate:
(
)
L
VV OUTIN
−
To determine the total on-time, or time at which the
inductor charges, the duty cycle needs to be calculated.
The duty cycle can be calculated as:
IN
OUT
V
V
D=
and the On time is:
1MHz
D
TON =
Therefore, peak to peak ripple current is:
()
L1MHz
V
V
VV
IIN
OUT
OUTIN
PKPK ×
×−
=
−
Figure 4 demonstrates the off-time. During the off-
time, the high-side internal P-channel MOSFET turns off.
Since the current in the inductor has to discharge, the
current flows through the free-wheeling Schottky diode
to the output. In this case, the inductor discharge rate is
Micrel, Inc. MIC2208
September 2007 10 M9999-090707-C
(where VD is the diode forward voltage):
1MHz
D1
TOFF
−
=
Figure 4. Off-Time
Discontinuous Operation
Discontinuous operation is when the inductor current
discharges to zero during the off cycle. Figure 5
demonstrates the switch voltage and inductor currents
during discontinuous operation.
Figure 5. Discontinuous Operation
When the inductor current (IL) has completely
discharged, the voltage on the switch node rings at the
frequency determined by the parasitic capacitance and
the inductor value. In Figure 5, it is drawn as a DC
voltage, but to see actual operation (with ringing and all)
refer to the functional characteristics.
Discontinuous mode of operation has the advantage
over full PWM in that at light loads, the MIC2208 will skip
pulses as necessary, reducing gate drive losses,
drastically improving light load efficiency.
Efficiency Considerations
Calculating the efficiency is as simple as measuring
power out and dividing it by the power in:
100
P
P
Efficiency
IN
OUT ×=
Where input power (PIN) is:
P
IN = VIN × IIN
and output power (POUT) is calculated as:
P
OUT = VOUT × IOUT
The Efficiency of the MIC2208 is determined by several
factors.
• RDSON (Internal P-channel Resistance)
• Diode conduction losses
• Inductor Conduction losses
• Switching losses
RDSON losses are caused by the current flowing through
the high side P-channel MOSFET. The amount of power
loss can be approximated by:
P
SW = RDSON × IOUT
2 × D
where D is the duty cycle.
Since the MIC2208 uses an internal P-channel
MOSFET, RDSON losses are inversely proportional to
supply voltage. Higher supply voltage yields a higher
gate to source voltage, reducing the RDSON, reducing the
MOSFET conduction losses. A graph showing typical
RDSON vs. input supply voltage can be found in the typical
characteristics section of this datasheet.
Diode conduction losses occur due to the forward
voltage drop (VF) and the output current. Diode power
losses can be approximated as follows:
P
D = VF × IOUT × (1-D)
For this reason, the Schottky diode is the rectifier of
choice. Using the lowest forward voltage drop will help
reduce diode conduction losses, and improve efficiency.
Duty cycle, or the ratio of output voltage-to-input voltage,
determines whether the dominant factor in conduction
losses will be the internal MOSFET or the Schottky
diode. Higher duty cycles place the power losses on the
high side switch, and lower duty cycles place the power
losses on the schottky diode. Inductor conduction losses
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September 2007 11 M9999-090707-C
(PL) can be calculated by multiplying the DC resistance
(DCR) times the square of the output current:
P
L = DCR × IOUT
2
Also, be aware that there are additional core losses
associated with switching current in an inductor. Since
most inductor manufacturers do not give data on the
type of material used, approximating core losses
becomes very difficult, so verify inductor temperature
rise.
Switching losses occur twice each cycle, when the
switch turns on and when the switch turns off. This is
caused by a non-ideal world where switching transitions
are not instantaneous, and neither are currents. Figure 6
demonstrates (Or exaggerates.) how switching losses
due to the transitions dissipate power in the switch.
Figure 6. Switch ing Transition Losses
Normally, when the switch is on, the voltage across the
switch is low (virtually zero) and the current through the
switch is high. This equates to low power dissipation.
When the switch is off, voltage across the switch is high
and the current is zero, again with power dissipation
being low. During the transitions, the voltage across the
switch (VS-D) and the current through the switch (IS-D) are
at middle, causing the transition to be the highest
instantaneous power point. During continuous mode,
these losses are the highest. Also, with higher load
currents, these losses are higher. For discontinuous
operation, the transition losses only occur during the “off”
transition since the “on” transitions there is no current
flow through the inductor.
Micrel, Inc. MIC2208
September 2007 12 M9999-090707-C
Component Selection
Input Capacitor
A 10µF ceramic is recommended on each VIN pin for
bypassing. X5R or X7R dielectrics are recommended for
the input capacitor. Y5V dielectrics lose most of their
capacitance over temperature and are therefore, not
recommended. Also, tantalum and electrolytic capacitors
alone are not recommended due their reduced RMS
current handling, reliability, and ESR increases.
An additional 0.1µF is recommended close to the VIN
and PGND pins for high frequency filtering. Smaller case
size capacitors are recommended due to their lower
ESR and ESL. Please refer to layout recommendations
for proper layout of the input capacitor.
Inductor Selection
The MIC2208 is designed for use with a 1µH inductor.
Proper selection should ensure the inductor can handle
the maximum average and peak currents required by the
load. Maximum current ratings of the inductor are
generally given in two methods; permissible DC current
and saturation current. Permissible DC current can be
rated either for a 40°C temperature rise or a 10% to 20%
loss in inductance. Ensure the inductor selected can
handle the maximum operating current. When saturation
current is specified, make sure that there is enough
margin that the peak current will not saturate the
inductor.
Diode Selection
Since the MIC2208 is non-synchronous, a free-wheeling
diode is required for proper operation. A schottky diode
is recommended due to the low forward voltage drop
and their fast reverse recovery time. The diode should
be rated to be able to handle the average output current.
Also, the reverse voltage rating of the diode should
exceed the maximum input voltage. The lower the
forward voltage drop of the diode the better the
efficiency. Please refer to the layout recommendations to
minimize switching noise.
Feedback Resistors
The feedback resistor set the output voltage by dividing
down the output and sending it to the feedback pin. The
feedback voltage is 1.0V. Calculating the set output
voltage is as follows:
⎟
⎠
⎞
⎜
⎝
⎛+= 1
R2
R1
VV FBOUT
Where R1 is the resistor from VOUT to FB and R2 is the
resistor from FB to GND. The recommended feedback
resistor values for common output voltages is available
in the bill of materials on page x. Although the range of
resistance for the FB resistors is very wide, R1 is
recommended to be 10K. This minimizes the effect the
parasitic capacitance of the FB node.
Bias filter
A small 10 resistor is recommended from the input
supply to the bias pin along with a small 0.1µF ceramic
capacitor from bias-to-ground. This will bypass the high
frequency noise generated by the violent switching of
high currents from reaching the internal reference and
control circuitry. Tantalum and electrolytic capacitors are
not recommended for the bias, these types of capacitors
lose their ability to filter at high frequencies.
20dB/Decade
Dominant
Pole
Zero
LC
Frequency
Gain (dB)
Compensation
The MIC2208 utilizes voltage mode compensation and
has the error amplifier pin (COMP) pinned out to allow it
to be compensated using external components. This
allows the MIC2208 to be stable with a wide range of
inductor and capacitor values.
TYPE II compensation
Type II compensation can be expressed as pole-zero-
pole. In our case, a dominant pole (R1 and C3) followed
by a zero (C3 and R4), allowing the final pole to be
provided by the output inductor and output capacitor (L
and COUT). This mode of compensation works well
when using higher ESR output capacitors, such as
tantalum and electrolytic dielectrics. The ESR of the
capacitor, along with the output capacitance provides a
zero (COUT and ESR) that negates one of the two poles
created by the inductor-output capacitor filter. This
allows the gain to cross the 0dB point with a -1 slope
(-20dB/decade).
Micrel, Inc. MIC2208
September 2007 13 M9999-090707-C
-20
-10
0
10
20
30
40
50
60
70
80
100 1k 10k 100k 1M
Frequency (KHz)
Type II Compensation
Gain (dB)
-72
-36
0
36
72
108
144
180
216
252
288
VIN=5VIN
VOUT=1.0V
IOUT=3A
MIC2208
Bill of Materials
Item Part Number Manufacturer Description Qty.
C2012JB0J106K TDK(1)
GRM219R60J106KE19 Murata(2)
C1a, C1b
08056D106MAT AVX(3)
10µF Ceramic Capacitor X5R 0805 6.3V 2
C2 0402ZD104MAT AVX(3) 0.1µF Ceramic Capacitor X5R 0402 10V 1
C3 0402ZD100MAT AVX(3) 100pF Ceramic Capacitor X5R 0402 10V 1
C4 TPME477M010R0030 AVX(3) 470µF Tantalum Capacitor 10V 1
D1 SSA33L Vishay Semi(4) 3A Schottky 30V SMA 1
RLF7030-1R0N6R4 TDK(1) 1µH Inductor 8.8m 7.1mm(L) x 6.8mm (W)x 3.2mm(H)
744 778 9001 Wurth Elektronik(5) 1µH Inductor 12m 7.3mm(L)x7.3mm(W)x3.2mm(H)
L1
IHLP2525AH-01 1 Vishay Dale(4) 1µH Inductor 17.5m (L)6.47mmx(W)6.86mmx(H) 1.8mm
1
R1 CRCW04023012F Vishay Dale(4) 30.1K 1% 0402 Resistor 1
CRCW04022002F Vishay Dale(4) 20 k 1% 0402 For 2.5VOUT
CRCW04023742F Vishay Dale(4) 37.4 k 1% 0402 For 1.8 VOUT
CRCW04026042F Vishay Dale(4) 60.4 k 1% 0402 For 1.5 VOUT
CRCW04021503F Vishay Dale(4) 150 k 1% 0402 For 1.2 VOUT
R2
Vishay Dale(4) Open For 1.0 VOUT
1
R4 CRCW04024993F Vishay Dale(4) 499K 1% 0402 Resistor 1
R5 CRCW040210R0F Vishay Dale(4) 10 1% 0402 Resistor 1
R6 CRCW04021002F Vishay Dale(4) 10K 1% 0402 Resistor 1
U1 MIC2208BML Micrel, Inc.(6) 1MHz 3A Buck Regulator 1
Notes:
1. TDK: www.tdk.com
2. Murata: www.murata.com
3. AVX: www.avx.com
4. Vishay: www.vishay.com
5. Wurth Elektronik Midcom, Inc.: www.midcom-inc..com
6. Micrel, Inc.: www.micrel.com
Micrel, Inc. MIC2208
September 2007 14 M9999-090707-C
TYPE III compensation
Type III in our case, is a dominant pole (C3 and R1)
followed by a zero (C3 and R4) and an additional zero
(C5 and R4), allowing the final pole to be provided by the
output inductor and output capacitor. This mode of
compensation is required when using low ESR output
capacitors, such as ceramic capacitors. The additional
zero offsets the double pole created by the
inductor/output capacitor filter. 20dB/Decade
Dominant
Pole
Zero
LC
Frequency
Frequency (Hz)
Gain(dB)
Zero
Type III Open Loop Gain Response
-20
-10
10
20
30
40
50
60
70
80
100 1k 10k 100k 1M
Frequency (KHz)
Type III Compensation
Gain (dB)
-72
-36
0
36
72
108
144
180
216
252
288
Gain
Phase
VIN=5VIN
VOUT=1.0V
IOUT=3A
COUT=47µF
MIC2208
Bill of Materials
Item Part Number Manufacturer Description Qty.
C2012JB0J106K TDK(1)
GRM219R60J106KE19 Murata(2)
C1a, C1b
08056D106MAT AVX(3)
10µF Ceramic Capacitor X5R 0805 6.3V 2
C2 0402ZD104MAT AVX(3) 0.1µF Ceramic Capacitor X5R 0402 10V 1
C3 0402ZD103MAT AVX(3) 1nF Ceramic Capacitor X5R 0402 10V 1
C3216X5R0J476K TDK(1)
GRM32ER60J476ME20 Murata(2) 47µF Ceramic Capacitor X5R 1206 6.3V
C4
12106D476MAT2A AVX(3) 47µF Ceramic Capacitor X5R 1210 6.3V
1
C5 VJ0402A330KXAA Vishay VT(4) 33pF Ceramic Capacitor 0402 1
D1 SSA33L Vishay Semi(4) 3A Schottky 30V SMA 1
RLF7030-1R0N6R4 TDK(1) 1µH Inductor 8.8m 7.1mm(L) x 6.8mm (W)x 3.2mm(H)
744 778 9001 Wurth Elektronik(5) 1µH Inductor 12m 7.3mm(L)x7.3mm(W)x3.2mm(H)
L1
IHLP2525AH-01 1 Vishay Dale(4) 1µH Inductor 17.5m (L)6.47mmx(W)6.86mmx(H) 1.8mm
1
Micrel, Inc. MIC2208
September 2007 15 M9999-090707-C
R1 CRCW04024992F Vishay Dale(4) 49.9K 1% 0402 Resistor 1
CRCW04023322F Vishay Dale(4) 33.3 k 1% 0402 For 2.5VOUT
CRCW04026192F Vishay Dale(4) 61.9 k 1% 0402 For 1.8 VOUT
CRCW04021003F Vishay Dale(4) 100 k 1% 0402 For 1.5 VOUT
CRCW04022493F Vishay Dale(4) 249 k 1% 0402 For 1.2 VOUT
R2
Vishay Dale(4) Open For 1.0 VOUT
1
R3 CRCW04024991F Vishay Dale(4) 499K 1% 0402 Resistor
R4 CRCW04024991F Vishay Dale(4) 90.9K 1% 0402 Resistor 1
R5 CRCW040210R0F Vishay Dale(4) 10 1% 0402 Resistor 1
R6 CRCW04021002F Vishay Dale(4) 10K 1% 0402 Resistor 1
U1 MIC2208BML Micrel, Inc.(6) 1MHz 3A Buck Regulator 1
Notes:
1. TDK: www.tdk.com
2. Murata: www.murata.com
3. AVX: www.avx.com
4. Vishay: www.vishay.com
5. Wurth Elektronik Midcom, Inc.: www.midcom-inc..com
6. Micrel, Inc.: www.micrel.com
Loop Stability and Bode Analysis
Bode analysis is an excellent way to measure small
signal stability and loop response in power supply
designs. Bode analysis monitors gain and phase of a
control loop. This is done by breaking the feedback loop
and injecting a signal into the feedback node and
comparing the injected signal to the output signal of the
control loop. This will require a network analyzer to
sweep the frequency and compare the injected signal to
the output signal. The most common method of injection
is the use of transformer. Figure 7 demonstrates how a
transformer is used to inject a signal into the feedback
network.
Figure 7. Transformer Injection
A 50 resistor allows impedance matching from the
network analyzer source. This method allows the DC
loop to maintain regulation and allow the network
analyzer to insert an AC signal on top of the DC voltage.
The network analyzer will then sweep the source while
monitoring A and R for an A/R measurement. While this
is the most common method for measuring the gain and
phase of a power supply, it does have significant
limitations. First, to measure low frequency gain and
phase, the transformer needs to be high in inductance.
This makes frequencies <100Hz require an extremely
large and expensive transformer. Conversely, it must be
able to inject high frequencies. Transformers with these
wide frequency ranges generally need to be custom
made and are extremely expensive (usually in the tune
of several hundred dollars!). By using an op-amp, cost
and frequency limitations used by an injection
transformer are completely eliminated. Figure 8
demonstrates using an op-amp in a summing amplifier
configuration for signal injection.
Network Analyzer
Source
+8V R1
1k
R3
1k R4
1k
50
Feedback Output
Netwo
r
k
Analyzer
“A” Input
Network
Analyzer
“R” Input MIC922BC5
Figure 8. Op Amp Injection
Micrel, Inc. MIC2208
September 2007 16 M9999-090707-C
R1 and R2 reduce the DC voltage from the output to the
non-inverting input by half. The network analyzer is
generally a 50 source. R1 and R2 also divide the AC
signal sourced by the network analyzer by half. These
two signals are “summed” together at half of their
original input. The output is then gained up by 2 by R3
and R4 (the 50 is to balance the network analyzer’s
source impedance) and sent to the feedback signal. This
essentially breaks the loop and injects the AC signal on
top of the DC output voltage and sends it to the
feedback. By monitoring the feedback “R” and output
“A”, gain and phase are measured. This method has no
minimum frequency. Ensure that the bandwidth of the
op-amp being used is much greater than the expected
bandwidth of the power supplies control loop. An op-amp
with >100MHz bandwidth is more than sufficient for most
power supplies (which includes both linear and
switching) and are more common and significantly
cheaper than the injection transformers previously
mentioned. The one disadvantage to using the op-amp
injection method, that the supply voltages need to below
the maximum operating voltage of the op-amp. Also, the
maximum output voltage for driving 50 inputs using the
MIC922 is 3V. For measuring higher output voltages, a
1M input impedance is required for the A and R
channels. Remember to always measure the output
voltage with an oscilloscope to ensure the measurement
is working properly. You should see a single sweeping
sinusoidal waveform without distortion on the output. If
there is distortion of the sinusoid, reduce the amplitude
of the source signal. You could be overdriving the
feedback causing a large signal response.
Output Impeda nc e and Transient
response
Output impedance, simply stated, is the amount of
output voltage deviation vs. the load current deviation.
The lower the output impedance, the better.
OUT
OUT
OUT I
V
Z=
Output impedance for a buck regulator is the parallel
impedance of the output capacitor and the MOSFET and
inductor divided by the gain:
COUT
LDSON
TOTAL X||
GAIN
XDCRR
Z++
=
To measure output impedance vs. frequency, the load
current must be load current must be swept across the
frequencies measured, while the output voltage is
monitored. Figure 9 shows a test set-up to measure
output impedance from 10Hz to 1MHz using the
MIC5190 high speed controller.
Figure 9. Output Impedan ce Measur ement
By setting up a network analyzer to sweep the feedback
current, while monitoring the output of the voltage
regulator and the voltage across the load resistance,
output impedance is easily obtainable. To keep the
current from being too high, a DC offset needs to be
applied to the network analyzer’s source signal. This can
be done with an external supply and 50 resistor. Make
sure that the currents are verified with an oscilloscope
first, to ensure the integrity of the signal measurement. It
is always a good idea to monitor the A and R
measurements with a scope while you are sweeping it.
To convert the network analyzer data from dBm to
something more useful (such as peak-to-peak voltage
and current in our case):
0.707
2501mW
10
dBm
10
V
×××
=
and peak to peak current:
LOAD
R0.707
2501mW
10
dBm
10
I×
×××
=
The following graph shows output impedance vs.
frequency at 2A load current sweeping the AC current
from 10Hz to 10MHz, at 1A peak to peak amplitude.
Output Impedance vs Frequency
0.001
0.01
0.1
1
10 10 0 1k 10 k 10 0 k 1M 10 M
Frequency (Hz)
Output
I
mpedance (Ohms)
3.3VIN
5V IN
VOUT = 1.8V
L =1µH
COUT = 4.7µF+0.1µF
Micrel, Inc. MIC2208
September 2007 17 M9999-090707-C
From this graph, you can see the effects of bandwidth
and output capacitance. For frequencies <100KHz, the
output impedance is dominated by the gain and
inductance. For frequencies >100KHz, the output
impedance is dominated by the capacitance. A good
approximation for transient response can be calculated
from determining the frequency of the load step in amps
per second:
2
A/sec
f=
Then, determine the output impedance by looking at the
output impedance vs. frequency graph. Then calculating
the voltage deviation times the load step:
VOUT = IOUT × ZOUT
The output impedance graph shows the relationship
between supply voltage and output impedance. This is
caused by the lower RDSON of the high side MOSFET
and the increase in gain with increased supply voltages.
This explains why higher supply voltages have better
transient response.
COUT
LDSON
TOTAL X||
GAIN
XDCRR
Z↑
++↓
=↓
Ripple measurements
To properly measure ripple on either input or output of a
switching regulator, a proper ring in tip measurement is
required. Standard oscilloscope probes come with a
grounding clip, or a long wire with an alligator clip.
Unfortunately, for high frequency measurements, this
ground clip can pick-up high frequency noise and
erroneously inject it into the measured output ripple.
The standard evaluation board accommodates a home
made version by providing probe points for both the
input and output supplies and their respective grounds.
This requires the removing of the oscilloscope probe
sheath and ground clip from a standard oscilloscope
probe and wrapping a non-shielded bus wire around the
oscilloscope probe. If there does not happen to be any
non shielded bus wire immediately available, the leads
from axial resistors will work. By maintaining the shortest
possible ground lengths on the oscilloscope probe, true
ripple measurements can be obtained
Micrel, Inc. MIC2208
September 2007 18 M9999-090707-C
Package Information
12-Pin MLF® (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
