LM1876
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LM1876 Overture™ Audio Power Amplifier Series
Dual 20W Audio Power Amplifier with Mute and Standby Modes
Check for Samples: LM1876
1FEATURES DESCRIPTION
The LM1876 is a stereo audio amplifier capable of
23• SPiKe Protection delivering typically 20W per channel of continuous
• Minimal Amount of External Components average output power into a 4Ωor 8Ωload with less
Necessary than 0.1% THD+N.
• Quiet Fade-In/Out Mute Mode Each amplifier has an independent smooth transition
• Standby-Mode fade-in/out mute and a power conserving standby
mode which can be controlled by external logic.
• Isolated 15-Lead TO-220 Package (PFM)
• Non-Isolated 15-lead TO-220 Package The performance of the LM1876, utilizing its Self
Peak Instantaneous Temperature (°Ke) ( SPiKe™)
• Wide Supply Range 20V - 64V protection circuitry, places it in a class above discrete
and hybrid amplifiers by providing an inherently,
APPLICATIONS dynamically protected Safe Operating Area (SOA).
• High-End Stereo TVs SPiKe protection means that these parts are
safeguarded at the output against overvoltage,
• Component Stereo undervoltage, overloads, including thermal runaway
• Compact Stereo and instantaneous temperature peaks.
KEY SPECIFICATIONS
• THD+N at 1kHz at 2 x 15W continuous
averageoutput power into 4Ωor 8Ω: 0.1%
(max)
• THD+N at 1kHz at continuous average output
power of 2 x 20W into 8Ω: 0.009% (typ)
• Standby current: 4.2mA (typ)
Connection Diagram
Figure 1. Plastic Package- Top View
Isolated Package (PFM)
See Package Number NDB0015B
Non-Isolated Package
See Package Number NDL0015A
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2Overture, SPiKe are trademarks of dcl_owner.
3All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 1999–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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Typical Application
Numbers in parentheses represent pinout for amplifier B.
*Optional component dependent upon specific design requirements.
Figure 2. Typical Audio Amplifier Application Circuit
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1)(2)(3)
Supply Voltage |VCC| + |VEE| (No Input) 64V
Supply Voltage |VCC| + |VEE| (with Input) 64V
Common Mode Input Voltage (VCC or VEE) and
|VCC| + |VEE|≤54V
Differential Input Voltage 54V
Output Current Internally Limited
Power Dissipation (4) 62.5W
ESD Susceptability (5) 2000V
Junction Temperature (6) 150°C
Thermal Resistance Isolated NDB-Package θJC 2°C/W
Non-Isolated NDL-Package θJC 1°C/W
Soldering Information NDB Package (10 sec.) 260°C
Storage Temperature −40°C to +150°C
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which specify performance limits. This assumes that the device is within the Operating
Ratings. Specifications are not ensure for parameters where no limit is given, however, the typical value is a good indication of device
performance.
(2) All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(4) For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a
thermal resistance of θJC = 2°C/W (junction to case) for the NDB package and θJC = 1°C/W for the NDL package. Refer to
DETERMINING THE CORRECT HEAT SINK in Application Information.
(5) Human body model, 100 pF discharged through a 1.5 kΩresistor.
(6) The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.
Operating Ratings (1)(2)
Temperature Range TMIN ≤TA≤TMAX −20°C ≤TA≤+85°C
Supply Voltage |VCC| + |VEE|(3) 20V to 64V
(1) All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which specify performance limits. This assumes that the device is within the Operating
Ratings. Specifications are not ensure for parameters where no limit is given, however, the typical value is a good indication of device
performance.
(3) Operation is specified up to 64V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations
are not taken into account. Refer to Application Information for a complete explanation.
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Electrical Characteristics (1)(2)
The following specifications apply for VCC = +22V, VEE =−22V with RL= 8Ωunless otherwise specified. Limits apply for TA=
25°C.
Symbol Parameter Conditions LM1876 Units
(Limits)
Typical(3 Limit(4)
)
|VCC| + Power Supply Voltage (5) GND −VEE ≥9V 20 V (min)
|VEE| 64 V (max)
PO(6) Output Power THD + N = 0.1% (max),
(Continuous Average) f = 1 kHz
|VCC| = |VEE| = 22V, RL= 8Ω20 15 W/ch (min)
|VCC| = |VEE| = 20V, RL= 4Ω(7) 22 15 W/ch (min)
THD + N Total Harmonic Distortion 15 W/ch, RL= 8Ω0.08 %
Plus Noise 15 W/ch, RL= 4Ω, |VCC| = |VEE| = 20V 0.1 %
20 Hz ≤f≤20 kHz, AV= 26 dB
Xtalk Channel Separation f = 1 kHz, VO= 10.9 Vrms 80 dB
SR (6) Slew Rate VIN = 1.414 Vrms, trise = 2 ns 18 12 V/μs (min)
Itotal (8) Total Quiescent Power Both Amplifiers VCM = 0V,
Supply Current VO= 0V, IO= 0 mA
Standby: Off 50 80 mA (max)
Standby: On 4.2 6 mA (max)
VOS (8) Input Offset Voltage VCM = 0V, IO= 0 mA 2.0 15 mV (max)
IBInput Bias Current VCM = 0V, IO= 0 mA 0.2 0.5 μA (max)
IOS Input Offset Current VCM = 0V, IO= 0 mA 0.002 0.2 μA (max)
IOOutput Current Limit |VCC| = |VEE| = 10V, tON = 10 ms, 3.5 2.9 Apk (min)
VO= 0V
VOD (8) Output Dropout Voltage (9) |VCC–VO|, VCC = 20V, IO= +100 mA 1.8 2.3 V (max)
|VO–VEE|, VEE =−20V, IO=−100 mA 2.5 3.2 V (max)
PSRR (8) Power Supply Rejection Ratio VCC = 25V to 10V, VEE =−25V, 115 85 dB (min)
VCM = 0V, IO= 0 mA
VCC = 25V, VEE =−25V to −10V 110 85 dB (min)
VCM = 0V, IO= 0 mA
CMRR (8) Common Mode Rejection Ratio VCC = 35V to 10V, VEE =−10V to −35V, 110 80 dB (min)
VCM = 10V to −10V, IO= 0 mA
AVOL (8) Open Loop Voltage Gain RL= 2 kΩ,ΔVO= 20 V 110 90 dB (min)
GBWP Gain Bandwidth Product fO= 100 kHz, VIN = 50 mVrms 7.5 5 MHz (min)
(1) All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which specify performance limits. This assumes that the device is within the Operating
Ratings. Specifications are not ensure for parameters where no limit is given, however, the typical value is a good indication of device
performance.
(3) Typicals are measured at 25°C and represent the parametric norm.
(4) Limits are ensure that all parts are tested in production to meet the stated values.
(5) VEE must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In
addition, the voltage differential between VCC and VEE must be greater than 14V.
(6) AC Electrical Test; Refer to Test Circuit #2 (AC Electrical Test Circuit).
(7) For a 4Ωload, and with ±20V supplies, the LM1876 can deliver typically 22W of continuous average output power with less than 0.1%
(THD + N). With supplies above ±20V, the LM1876 cannot deliver more than 22W into a 4Ωdue to current limiting of the output
transistors. Thus, increasing the power supply above ±20V will only increase the internal power dissipation, not the possible output
power. Increased power dissipation will require a larger heat sink as explained in Application Information.
(8) DC Electrical Test; Refer to Test Circuit #1 (DC Electrical Test Circuit).
(9) The output dropout voltage, VOD, is the supply voltage minus the clipping voltage. Refer to Figure 16,Figure 17, and Figure 18 in
Typical Performance Characteristics.
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Electrical Characteristics (1)(2) (continued)
The following specifications apply for VCC = +22V, VEE =−22V with RL= 8Ωunless otherwise specified. Limits apply for TA=
25°C.
Symbol Parameter Conditions LM1876 Units
(Limits)
Typical(3 Limit(4)
)
eIN (6) Input Noise IHF—A Weighting Filter 2.0 8 μV (max)
RIN = 600Ω(Input Referred)
SNR Signal-to-Noise Ratio PO= 1W, A—Weighted, 98 dB
Measured at 1 kHz, RS= 25Ω
PO= 15W, A—Weighted 108 dB
Measured at 1 kHz, RS= 25Ω
AMMute Attenuation Pin 6,11 at 2.5V 115 80 dB (min)
Standby Pin
VIL Standby Low Input Voltage Not in Standby Mode 0.8 V (max)
VIH Standby High Input Voltage In Standby Mode 2.0 2.5 V (min)
Mute pin
VIL Mute Low Input Voltage Outputs Not Muted 0.8 V (max)
VIH Mute High Input Voltage Outputs Muted 2.0 2.5 V (min)
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Bridged Amplifier Application Circuit
Figure 3. Bridged Amplifier Application Circuit
Single Supply Application Circuit
*Optional components dependent upon specific design requirements.
Figure 4. Single Supply Amplifier Application Circuit
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Auxiliary Amplifier Application Circuit
Figure 5. Special Audio Amplifier Application Circuit
Equivalent Schematic
(excluding active protection circuitry)
Figure 6. LM1876 (per Amp)
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(excluding active protection circuitry)
External Components Description
Components Functional Description
1 RBPrevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon
power down of the system due to the low input impedance of the circuitry when the undervoltage circuitry is off.
This phenomenon occurs when the supply voltages are below 1.5V.
2 RiInverting input resistance to provide AC gain in conjunction with Rf.
3 RfFeedback resistance to provide AC gain in conjunction with Ri.
4 Ci(1) Feedback capacitor which ensures unity gain at DC. Also creates a highpass filter with Riat fC= 1/(2πRiCi).
5 CSProvides power supply filtering and bypassing. Refer to SUPPLY BYPASSING for proper placement and selection
of bypass capacitors.
6 RV(1) Acts as a volume control by setting the input voltage level.
7 RIN (1) Sets the amplifier's input terminals DC bias point when CIN is present in the circuit. Also works with CIN to create a
highpass filter at fC= 1/(2πRINCIN). Refer to Figure 5.
8 CIN (1) Input capacitor which blocks the input signal's DC offsets from being passed onto the amplifier's inputs.
9 RSN (1) Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
10 CSN (1) Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The pole is
set at fC= 1/(2πRSNCSN). Refer to Figure 5.
11 L (1) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of
the series resonant circuit. Also provides a low impedance at low frequencies to short out R and pass audio signals
12 R (1) to the load. Refer to Figure 5.
13 RAProvides DC voltage biasing for the transistor Q1 in single supply operation.
14 CAProvides bias filtering for single supply operation.
15 RINP (1) Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to CLICKS AND
POPS for a more detailed explanation of the function of RINP.
16 RBI Provides input bias current for single supply operation. Refer to CLICKS AND POPS for a more detailed
explanation of the function of RBI.
17 REEstablishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the half-
supply point along with CA.
(1) Optional components dependent upon specific design requirements.
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Typical Performance Characteristics
THD + N THD + N
vs vs
Frequency Frequency
Figure 7. Figure 8.
THD + N
vs THD + N vs
Frequency Output Power
Figure 9. Figure 10.
THD + N vs THD + N vs
Output Power Output Power
Figure 11. Figure 12.
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Typical Performance Characteristics (continued)
THD + N vs THD + N vs
Output Power Output Power
Figure 13. Figure 14.
THD + N vs Clipping Voltage vs
Output Power Supply Voltage
Figure 15. Figure 16.
Clipping Voltage vs Clipping Voltage vs
Supply Voltage Supply Voltage
Figure 17. Figure 18.
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Typical Performance Characteristics (continued)
Output Power vs Power Dissipation vs
Load Resistance Output Power
Figure 19. Figure 20.
Power Dissipation vs Output Power vs
Output Power Supply Voltage
Figure 21. Figure .
Output Mute vs Output Mute vs
Mute Pin Voltage Mute Pin Voltage
Figure 22. Figure 23.
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Typical Performance Characteristics (continued)
Channel Separation vs
Frequency Pulse Response
Figure 24. Figure 25.
Power Supply
Large Signal Response Rejection Ratio
Figure 26. Figure 27.
Common-Mode Open Loop
Rejection Ratio Frequency Response
Figure 28. Figure 29.
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Typical Performance Characteristics (continued)
SPiKe Protection
Safe Area Response
Figure 30. Figure 31.
Supply Current vs Pulse Thermal
Supply Voltage Resistance
Figure 32. Figure 33.
Pulse Thermal Supply Current vs
Resistance Output Voltage
Figure 34. Figure 35.
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Typical Performance Characteristics (continued)
Pulse Power Limit Pulse Power Limit
Figure 36. Figure 37.
Supply Current vs Supply Current (ICC) vs
Case Temperature Standby Pin Voltage
Figure 38. Figure 39.
Supply Current (IEE) vs Input Bias Current vs
Standby Pin Voltage Case Temperature
Figure 40. Figure 41.
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Typical Performance Characteristics (continued)
Output Power/Channel vs Output Power/Channel vs
Supply Voltage Supply Voltage
f = 1kHz, RL= 4Ω, 80kHz BW f = 1kHz, RL= 6Ω, 80kHz BW
Figure 42. Figure 43.
Output Power/Channel vs
Supply Voltage
f = 1kHz, RL= 8Ω, 80kHz BW
Figure 44.
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APPLICATION INFORMATION
MUTE MODE
By placing a logic-high voltage on the mute pins, the signal going into the amplifiers will be muted. If the mute
pins are left floating or connected to a logic-low voltage, the amplifiers will be in a non-muted state. There are
two mute pins, one for each amplifier, so that one channel can be muted without muting the other if the
application requires such a configuration. Refer to Typical Performance Characteristics for Figure 22 and
Figure 23.
STANDBY MODE
The standby mode of the LM1876 allows the user to drastically reduce power consumption when the amplifiers
are idle. By placing a logic-high voltage on the standby pins, the amplifiers will go into Standby Mode. In this
mode, the current drawn from the VCC supply is typically less than 10 μA total for both amplifiers. The current
drawn from the VEE supply is typically 4.2 mA. Clearly, there is a significant reduction in idle power consumption
when using the standby mode. There are two Standby pins, so that one channel can be put in standby mode
without putting the other amplifier in standby if the application requires such flexibility. Refer to Typical
Performance Characteristics for Figure 39 and Figure 40.
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry allows the power supplies and their corresponding
capacitors to come up close to their full values before turning on the LM1876 such that no DC output spikes
occur. Upon turn-off, the output of the LM1876 is brought to ground before the power supplies such that no
transients occur at power-down.
OVER-VOLTAGE PROTECTION
The LM1876 contains over-voltage protection circuitry that limits the output current to approximately 3.5 Apk
while also providing voltage clamping, though not through internal clamping diodes. The clamping effect is quite
the same, however, the output transistors are designed to work alternately by sinking large current spikes.
SPiKe PROTECTION
The LM1876 is protected from instantaneous peak-temperature stressing of the power transistor array. The
Figure 30 in Typical Performance Characteristics shows the area of device operation where SPiKe Protection
Circuitry is not enabled. The waveform to the right of the SOA graph exemplifies how the dynamic protection will
cause waveform distortion when enabled.
THERMAL PROTECTION
The LM1876 has a sophisticated thermal protection scheme to prevent long-term thermal stress of the device.
When the temperature on the die reaches 165°C, the LM1876 shuts down. It starts operating again when the die
temperature drops to about 155°C, but if the temperature again begins to rise, shutdown will occur again at
165°C. Therefore, the device is allowed to heat up to a relatively high temperature if the fault condition is
temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion between the thermal
shutdown temperature limits of 165°C and 155°C. This greatly reduces the stress imposed on the IC by thermal
cycling, which in turn improves its reliability under sustained fault conditions.
Since the die temperature is directly dependent upon the heat sink used, the heat sink should be chosen such
that thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the
cost and space constraints of the system will improve the long-term reliability of any power semiconductor
device, as discussed in DETERMINING THE CORRECT HEAT SINK.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a very important parameter requiring a thorough
understanding if optimum power output is to be obtained. An incorrect maximum power dissipation calculation
may result in inadequate heat sinking causing thermal shutdown and thus limiting the output power.
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Equation 1 exemplifies the theoretical maximum power dissipation point of each amplifier where VCC is the total
supply voltage.
PDMAX = VCC2/2π2RL(1)
Thus by knowing the total supply voltage and rated output load, the maximum power dissipation point can be
calculated. The package dissipation is twice the number which results from Equation 1 since there are two
amplifiers in each LM1876. Refer to Figure 21 and Figure 20 in Typical Performance Characteristics which show
the actual full range of power dissipation not just the maximum theoretical point that results from Equation 1.
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level
such that the thermal protection circuitry does not operate under normal circumstances.
The thermal resistance from the die (junction) to the outside air (ambient) is a combination of three thermal
resistances, θJC,θCS, and θSA. In addition, the thermal resistance, θJC (junction to case), of the LM1876TF is
2°C/W and the LM1876T is 1°C/W. Using Thermalloy Thermacote thermal compound, the thermal resistance,
θCS (case to sink), is about 0.2°C/W. Since convection heat flow (power dissipation) is analogous to current flow,
thermal resistance is analogous to electrical resistance, and temperature drops are analogous to voltage drops,
the power dissipation out of the LM1876 is equal to the following:
PDMAX = (TJMAX−TAMB)/θJA
where
• TJMAX = 150°C
• TAMB is the system ambient temperature
•θJA =θJC +θCS +θSA (2)
Once the maximum package power dissipation has been calculated using Equation 1, the maximum thermal
resistance, θSA, (heat sink to ambient) in °C/W for a heat sink can be calculated. This calculation is made using
Equation 3 which is derived by solving for θSA in Equation 2.
θSA = [(TJMAX−TAMB)−PDMAX(θJC +θCS)]/PDMAX (3)
Again it must be noted that the value of θSA is dependent upon the system designer's amplifier requirements. If
the ambient temperature that the audio amplifier is to be working under is higher than 25°C, then the thermal
resistance for the heat sink, given all other things are equal, will need to be smaller.
SUPPLY BYPASSING
The LM1876 has excellent power supply rejection and does not require a regulated supply. However, to improve
system performance as well as eliminate possible oscillations, the LM1876 should have its supply leads
bypassed with low-inductance capacitors having short leads that are located close to the package terminals.
Inadequate power supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or
by high frequency instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large
tantalum or electrolytic capacitor (10 μF or larger) which is used to absorb low frequency variations and a small
ceramic capacitor (0.1 μF) to prevent any high frequency feedback through the power supply lines.
If adequate bypassing is not provided, the current in the supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal causes distortion at high frequencies requiring that the
supplies be bypassed at the package terminals with an electrolytic capacitor of 470 μF or more.
BRIDGED AMPLIFIER APPLICATION
The LM1876 has two operational amplifiers internally, allowing for a few different amplifier configurations. One of
these configurations is referred to as “bridged mode” and involves driving the load differentially through the
LM1876's outputs. This configuration is shown in Figure 3. Bridged mode operation is different from the classical
single-ended amplifier configuration where one side of its load is connected to ground.
A bridge amplifier design has a distinct advantage over the single-ended configuration, as it provides differential
drive to the load, thus doubling output swing for a specified supply voltage. Consequently, theoretically four times
the output power is possible as compared to a single-ended amplifier under the same conditions. This increase in
attainable output power assumes that the amplifier is not current limited or clipped.
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A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal
power dissipation. For each operational amplifier in a bridge configuration, the internal power dissipation will
increase by a factor of two over the single ended dissipation. Thus, for an audio power amplifier such as the
LM1876, which has two operational amplifiers in one package, the package dissipation will increase by a factor
of four. To calculate the LM1876's maximum power dissipation point for a bridged load, multiply Equation 1 by a
factor of four.
This value of PDMAX can be used to calculate the correct size heat sink for a bridged amplifier application. Since
the internal dissipation for a given power supply and load is increased by using bridged-mode, the heatsink's θSA
will have to decrease accordingly as shown by Equation 3. Refer to DETERMINING THE CORRECT HEAT SINK
for a more detailed discussion of proper heat sinking for a given application.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM1876 is a split supply amplifier. But as shown in Figure 4, the LM1876 can also
be used in a single power supply configuration. This involves using some external components to create a half-
supply bias which is used as the reference for the inputs and outputs. Thus, the signal will swing around half-
supply much like it swings around ground in a split-supply application. Along with proper circuit biasing, a few
other considerations must be accounted for to take advantage of all of the LM1876 functions.
The LM1876 possesses a mute and standby function with internal logic gates that are half-supply referenced.
Thus, to enable either the Mute or Standby function, the voltage at these pins must be a minimum of 2.5V above
half-supply. In single-supply systems, devices such as microprocessors and simple logic circuits used to control
the mute and standby functions, are usually referenced to ground, not half-supply. Thus, to use these devices to
control the logic circuitry of the LM1876, a “level shifter,” like the one shown in Figure 45, must be employed. A
level shifter is not needed in a split-supply configuration since ground is also half-supply.
Figure 45. Level Shift Circuit
When the voltage at the Logic Input node is 0V, the 2N3904 is “off” and thus resistor Rcpulls up mute or standby
input to the supply. This enables the mute or standby function. When the Logic Input is 5V, the 2N3904 is “on”
and consequently, the voltage at the collector is essentially 0V. This will disable the mute or standby function,
and thus the amplifier will be in its normal mode of operation. Rshift, along with Cshift, creates an RC time constant
that reduces transients when the mute or standby functions are enabled or disabled. Additionally, Rshift limits the
current supplied by the internal logic gates of the LM1876 which insures device reliability. Refer to MUTE MODE
and STANDBY MODE in Application Information for a more detailed description of these functions.
CLICKS AND POPS
In the typical application of the LM1876 as a split-supply audio power amplifier, the IC exhibits excellent “click”
and “pop” performance when utilizing the mute and standby modes. In addition, the device employs Under-
Voltage Protection, which eliminates unwanted power-up and power-down transients. The basis for these
functions are a stable and constant half-supply potential. In a split-supply application, ground is the stable half-
supply potential. But in a single-supply application, the half-supply needs to charge up just like the supply rail,
VCC. This makes the task of attaining a clickless and popless turn-on more challenging. Any uneven charging of
the amplifier inputs will result in output clicks and pops due to the differential input topology of the LM1876.
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To achieve a transient free power-up and power-down, the voltage seen at the input terminals should be ideally
the same. Such a signal will be common-mode in nature, and will be rejected by the LM1876. In Figure 4, the
resistor RINP serves to keep the inputs at the same potential by limiting the voltage difference possible between
the two nodes. This should significantly reduce any type of turn-on pop, due to an uneven charging of the
amplifier inputs. This charging is based on a specific application loading and thus, the system designer may need
to adjust these values for optimal performance.
As shown in Figure 4, the resistors labeled RBI help bias up the LM1876 off the half-supply node at the emitter of
the 2N3904. But due to the input and output coupling capacitors in the circuit, along with the negative feedback,
there are two different values of RBI, namely 10 kΩand 200 kΩ. These resistors bring up the inputs at the same
rate resulting in a popless turn-on. Adjusting these resistors values slightly may reduce pops resulting from
power supplies that ramp extremely quick or exhibit overshoot during system turn-on.
AUDIO POWER AMPLlFIER DESIGN
Design a 15W/8ΩAudio Amplifier
Given:
Power Output 15 Wrms
Load Impedance 8Ω
Input Level 1 Vrms(max)
Input Impedance 47 kΩ
Bandwidth 20 Hz−20 kHz ±0.25 dB
A designer must first determine the power supply requirements in terms of both voltage and current needed to
obtain the specified output power. VOPEAK can be determined from Equation 4 and IOPEAK from Equation 5.
(4)
(5)
To determine the maximum supply voltage the following conditions must be considered. Add the dropout voltage
to the peak output swing VOPEAK, to get the supply rail at a current of IOPEAK. The regulation of the supply
determines the unloaded voltage which is usually about 15% higher. The supply voltage will also rise 10% during
high line conditions. Therefore the maximum supply voltage is obtained from the following equation.
Max supplies ≈± (VOPEAK + VOD) (1 + regulation) (1.1) (6)
For 15W of output power into an 8Ωload, the required VOPEAK is 15.49V. A minimum supply rail of 20.5V results
from adding VOPEAK and VOD. With regulation, the maximum supplies are ±26V and the required IOPEAK is 1.94A
from Equation 5. It should be noted that for a dual 15W amplifier into an 8Ωload the IOPEAK drawn from the
supplies is twice 1.94 Apk or 3.88 Apk. At this point it is a good idea to check the Power Output vs Supply
Voltage to ensure that the required output power is obtainable from the device while maintaining low THD+N. In
addition, the designer should verify that with the required power supply voltage and load impedance, that the
required heatsink value θSA is feasible given system cost and size constraints. Once the heatsink issues have
been addressed, the required gain can be determined from Equation 7.
(7)
From Equation 7, the minimum AVis: AV≥11.
By selecting a gain of 21, and with a feedback resistor, Rf= 20 kΩ, the value of Rifollows from Equation 8.
Ri= Rf(AV−1) (8)
Thus with Ri= 1 kΩa non-inverting gain of 21 will result. Since the desired input impedance was 47 kΩ, a value
of 47 kΩwas selected for RIN. The final design step is to address the bandwidth requirements which must be
stated as a pair of −3 dB frequency points. Five times away from a −3 dB point is 0.17 dB down from passband
response which is better than the required ±0.25 dB specified. This fact results in a low and high frequency pole
of 4 Hz and 100 kHz respectively. As stated in External Components Description, Riin conjunction with Cicreate
a high-pass filter.
Ci≥1/(2π* 1 kΩ* 4 Hz) = 39.8 μF; use 39 μF. (9)
20 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated
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LM1876
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SNAS097C –MAY 1999–REVISED APRIL 2013
The high frequency pole is determined by the product of the desired high frequency pole, fH, and the gain, AV.
With a AV= 21 and fH= 100 kHz, the resulting GBWP is 2.1 MHz, which is less than the minimum GBWP of the
LM1876 of 5 MHz. This will ensure that the high frequency response of the amplifier will be no worse than 0.17
dB down at 20 kHz which is well within the bandwidth requirements of the design.
Copyright © 1999–2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LM1876
LM1876
SNAS097C –MAY 1999–REVISED APRIL 2013
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REVISION HISTORY
Changes from Revision B (April 2013) to Revision C Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 21
22 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated
Product Folder Links: LM1876
PACKAGE OPTION ADDENDUM
www.ti.com 27-Oct-2016
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM1876TF/NOPB ACTIVE TO-220 NDB 15 20 Pb-Free (RoHS
Exempt) CU SN Level-1-NA-UNLIM -20 to 85 LM1876TF
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
PACKAGE OPTION ADDENDUM
www.ti.com 27-Oct-2016
Addendum-Page 2
MECHANICAL DATA
NDB0015B
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TF15B (REV B)
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