MP1583
3A, 23V, 385KHz
Step-Down Converter
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The Future of Analog IC Technology
DESCRIPTION
The MP1583 is a step-down regulator with a
built-in internal Power MOSFET. It achieves 3A
of continuous output current over a wide input
supply range with excellent load and line
regulation.
Current mode operation provides fast transient
response and eases loop stabilization.
Fault condition protection includes
cycle-by-cycle current limiting and thermal
shutdown. An adjustable soft-start reduces the
stress on the input source at startup. In
shutdown mode the regulator draws 20A of
supply current.
The MP1583 requires a minimum number of
external components, providing a compact
solution.
FEATURES
• 3A Output Current
• Programmable Soft-Start
• 100m Internal Power MOSFET Switch
• Stable with Low ESR Output Ceramic Capacitors
• Up to 95% Efficiency
• 20A Shutdown Mode
• Fixed 385KHz Frequency
• Thermal Shutdown
• Cycle-by-Cycle Over Current Protection
• Wide 4.75V to 23V Operating Input Range
• Output Adjustable from 1.22V to 21V
• Under-Voltage Lockout
APPLICATIONS
• Distributed Power Systems
• Battery Chargers
• Pre-Regulator for Linear Regulators
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
INPUT
4.75V to 23V
OUTPUT
2.5V
3A
MP1583
BSIN
FB
SW
SS
GND COMP
EN
1
3
5
64
8
7
2
OPEN =
AUTOMATIC
STARTUP
10nF 5.6nF
10nF
10μF
CERAMIC
22μF
CERAMIC
3.9kΩ
10kΩ
10.5kΩ
15μH
B330A
MP1583_EC01
Efficiency Curve
V
IN
= 10V
100
90
80
70
60
50
EFFICIENCY (%)
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5
LOAD CURRENT (A)
V
OUT
=5.0V
V
OUT
=2.5V
V
OUT
=3.3V
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
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ORDERING INFORMATION
Part Number* Package Top Marking Free Air Temperature (TA)
MP1583DN SOIC8E MP1583DN
–40°C to +85°C
MP1583DP PDIP8 MP1583DP
–40°C to +85°C
* For Tape & Reel, add suffix –Z (e.g. MP8736DL–Z)
For RoHS compliant packaging, add suffix –LF (e.g. MP8736DL–LF–Z)
PACKAGE REFERENCE
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage VIN .......................–0.3V to +28V
Switch Voltage VSW ................. –1V to VIN + 0.3V
Bootstrap Voltage VBS ....VSW – 0.3V to VSW + 6V
FB, COMP and SS Pins.................–0.3V to +6V
Continuous Power Dissipation (TA = +25°C)(2)
SOIC8E...................................................... 2.5W
PDIP8 ........................................................ 1.2W
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ............. –65°C to +150°C
Recommended Operating Conditions (3)
Input Voltage VIN ............................4.75V to 23V
Operating Junct. Temp (TJ)...... -40°C to +125°C
Thermal Resistance (4) θJA θJC
SOIC8E .................................. 50 ...... 10... °C/W
PDIP8 .................................... 104 ..... 45... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-to-
ambient thermal resistance JA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)-
TA)/JA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB
BS
IN
SW
GND
SS
EN
COMP
FB
1
2
3
4
8
7
6
5
TOP VIEW
SOIC8N/PDIP8
MP1583_PD01
EXPOSED PAD
(SOIC8N ONLY)
CONNECT TO PIN 4
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
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ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameters Symbol Condition Min Typ Max Units
Shutdown Supply Current VEN = 0V 20 30 µA
Supply Current VEN = 2.8V, VFB =1.4V 1.0 1.2 mA
Feedback Voltage VFB 4.75V ≤ VIN ≤ 23V 1.194 1.222 1.250 V
Error Amplifier Voltage Gain AVEA 400 V/V
Error Amplifier Transconductance GEA ΔICOMP = ±10A 500 800 1120 µA/V
High-Side Switch On-Resistance RDS(ON)1 0.1
Low-Side Switch On-Resistance RDS(ON)2 10
High-Side Switch Leakage Current VEN = 0V, VSW = 0V 0 10 µA
Current Limit 4.0 4.9 6.0 A
Current Sense to COMP Transconductance GCS 3.8 A/V
Oscillation Frequency fS 335 385 435 KHz
Short Circuit Oscillation Frequency VFB = 0V 25 40 55 KHz
Maximum Duty Cycle DMAX V
FB = 1.0V 90 %
Minimum Duty Cycle VFB = 1.5V 0 %
EN Shutdown Threshold Voltage 0.9 1.2 1.5 V
Enable Pull Up Current VEN = 0V 1.1 1.8 2.5 µA
EN UVLO Threshold VEN Rising 2.37 2.54 2.71 V
EN UVLO Threshold Hysteresis 210 mV
Soft-Start Period CSS = 0.1µF 10 ms
Thermal Shutdown 160 °C
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
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TYPICAL PERFORMANCE CHARACTERISTICS
V
EN
5V/div.
V
OUT
2V/div.
I
L
1A/div.
MP1583-TPC02
Soft-Start
CSS Open, VIN = 10V, VOUT = 3.3V,
1.5A Resistive Load
V
EN
5V/div.
V
OUT
2V/div.
I
L
1A/div.
MP1583-TPC03
V
EN
5V/div.
V
OUT
2V/div.
I
L
1A/div.
1ms/div.
MP1583-TPC04
MP1583-TPC01
100
90
80
70
60
50
EFFICIENCY (%)
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5
LOAD CURRENT (A)
Efficiency Curve
VIN = 7V
VOUT=5.0V
VOUT=2.5V
VOUT=3.3V
PIN FUNCTIONS
Pin # Name Description
1 BS
High-Side Gate Drive Bootstrap Input. BS supplies the drive for the high-side N-Channel MOSFET
switch. Connect a 4.7nF or greater capacitor from SW to BS to power the high-side switch.
2 IN
Power Input. IN supplies the power to the IC. Drive IN with a 4.75V to 23V power source. Bypass
IN to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input
Capacitor
3 SW
Power Switching Output. SW is the switching node that supplies power to the output. Connect the
output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to
power the high-side switch.
4 GND Ground. (Note: For the SOIC8E package, connect the exposed pad on backside to Pin 4).
5 FB
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage
divider from the output voltage. The feedback threshold is 1.222V. See Setting the Output Voltage
6 COMP
Compensation Node. COMP is used to compensate the regulation control loop. Connect a series
RC network from COMP to GND to compensate the regulation control loop. See Compensation
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PIN FUNCTIONS (continued)
Pin # Name Description
7 EN
Enable/UVLO. A voltage greater than 2.71V enables operation. For complete low
current shutdown the EN pin voltage needs to be at less than 900mV. When the
voltage on EN exceeds 1.2V, the internal regulator will be enabled and the soft-start
capacitor will begin to charge. The MP1583 will start switching after the EN pin
voltage reaches 2.71V. There is 7V zener connected between EN and GND. If EN is
driven by external signal, the voltage should never exceed 7V.
8 SS
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to
set the soft-start period. To disable the soft-start feature, leave SS unconnected.
OPERATION
The MP1583 is a current-mode step-down
regulator. It regulates input voltages from 4.75V to
23V down to an output voltage as low as 1.222V,
and is able to supply up to 3A of load current.
The MP1583 uses current-mode control to
regulate the output voltage. The output voltage is
measured at FB through a resistive voltage
divider and amplified through the internal error
amplifier. The output current of the
transconductance error amplifier is presented at
COMP where a RC network compensates the
regulation control system.
The voltage at COMP is compared to the
internally measured switch current to control the
output voltage.
The converter uses an internal N-Channel
MOSFET switch to step-down the input voltage to
the regulated output voltage. Since the MOSFET
requires a gate voltage greater than the input
voltage, a boost capacitor connected between
SW and BS drives the gate. The capacitor is
internally charged when SW is low.
An internal 10 switch from SW to GND is used
to insure that SW is pulled to GND when SW is
low in order to fully charge the BS capacitor.
LOCKOUT
COMPARATOR
ERROR
AMPLIFIER
FREQUENCY
FOLDBACK
COMPARATOR GM = 800μA/V
INTERNAL
REGULATORS
1μA
7V
1.8V
SLOPE
COMP
CLK
CURRENT
COMPARATOR
CURRENT
SENSE
AMPLIFIER
SHUTDOWN
COMPARATOR
SS
8
COMP
6
IN 2
EN 7
GND
4
OSCILLATOR
40/385KHz
S
R
Q
SW
3
BS
1
5V
+
Σ
Q
1.2V
++
2.54V +
1.222V0.7V +
+
FB
5
--
--
--
--
--
--
Figure 1—Functional Block Diagram
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
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APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to the
FB pin. The voltage divider divides the output
voltage down to the feedback voltage by the
ratio:
21 2
RR
R
VV OUTFB +
=
Where VFB is the feedback voltage and VOUT is
the output voltage.
Thus the output voltage is:
221
22.1 R
RR
VVOUT
+
×=
A typical value for R2 can be as high as 100k,
but a typical value is 10k. Using that value, R1
is determined by:
))(22.1(18.81 Ω−×= kVVR OUT
For example, for a 3.3V output voltage, R2 is
10k, and R1 is 17k.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current and
lower output ripple voltage. However, larger
value inductors have a larger physical size,
higher series resistance, and/or lower
saturation current. A good rule for determining
the inductance to use is to allow the inductor
peak-to-peak ripple current to be approximately
30% of the maximum switch current limit. Also,
make sure that the peak inductor current is
below the maximum switch current limit. The
inductance value can be calculated by:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−×
×
=
IN
OUT
LS
OUT
V
V
1
ΔIf
V
L
Where VIN is the input voltage, fS is the 385KHz
switching frequency and IL is the peak-to-peak
inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current.
The peak inductor current can be calculated by:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−×
××
+=
IN
OUT
S
OUT
LOADLP V
V
1
Lf2
V
II
Where ILOAD is the load current.
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which inductor to use mainly depends on the
price vs. size requirements and any EMI
requirements.
Table 1—Inductor Selection Guide
Package
Dimensions
(mm)
Vendor/
Model
Core
Type
Core
Material WL H
Sumida
CR75 Open Ferrite 7.0 7.8 5.5
CDH74 Open Ferrite 7.3 8.0 5.2
CDRH5D28 Shielded Ferrite 5.5 5.7 5.5
CDRH5D28 Shielded Ferrite 5.5 5.7 5.5
CDRH6D28 Shielded Ferrite 6.7 6.7 3.0
CDRH104R Shielded Ferrite 10.1 10.0 3.0
Toko
D53LC
Type A Shielded Ferrite 5.0 5.0 3.0
D75C Shielded Ferrite 7.6 7.6 5.1
D104C Shielded Ferrite 10.0 10.0 4.3
D10FL Open Ferrite 9.7 1.5 4.0
Coilcraft
DO3308 Open Ferrite 9.4 13.0 3.0
DO3316 Open Ferrite 9.4 13.0 5.1
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
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Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off.
Use a Schottky diode to reduce losses due to
the diode forward voltage and recovery times.
Choose a diode whose maximum reverse voltage
rating is greater than the maximum input voltage,
and whose current rating is greater than the
maximum load current. Table 2 lists example
Schottky diodes and manufacturers.
Table 2—Diode Selection Guide
Diode
V
oltage/Current
Rating
Manufacture
SK33 30V, 3A Diodes Inc.
SK34 40V, 3A Diodes Inc.
B330 30V, 3A Diodes Inc.
B340 40V, 3A Diodes Inc.
MBRS330 30V, 3A On Semiconductor
MBRS340 40V, 3A On Semiconductor
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
will also suffice.
Since the input capacitor absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
⎟
⎟
⎟
⎠
⎞
⎜
⎜
⎜
⎝
⎛
×−×=
IN
OUT
IN
OUT
LOADC V
V
1
V
V
II 1
The worst-case condition occurs at VIN = 2VOUT,
where:
2
1LOAD
C
I
I=
For simplification, choose an input capacitor
whose RMS current rating is greater than half of
the maximum load current.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor (i.e. 0.1F) should be placed as close
to the IC as possible.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive voltage
ripple at the input. The input voltage ripple
caused by capacitance can be estimated by:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−××
×
=Δ
IN
OUT
IN
OUT
S
LOAD
IN V
V
1
V
V
Cf
I
V1
Where C1 is the input capacitance value.
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred so as to
keep the output voltage ripple low. The output
voltage ripple can be estimated by:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
××
+×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−×
×
=Δ 28 1
1Cf
R
V
V
Lf
V
V
S
ESR
IN
OUT
S
OUT
OUT
Where L is the inductor value, C2 is the output
capacitance value and RESR is the equivalent
series resistance (ESR) value of the output
capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance, which is the
main cause for the output voltage ripple. For
simplification, the output voltage ripple can be
estimated by:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−×
×××
=
IN
OUT
S
OUT
OUT V
V
1
CLf
V
ΔV28 2
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
ESR
IN
OUT
S
OUT
OUT R
V
V
1
Lf
V
ΔV×
⎟
⎠
⎞
⎜
⎝
⎛−×
×
=
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The MP1583 can be optimized for a wide range
of capacitance and ESR values.
Compensation Components
The MP1583 employs current mode control for
easy compensation and fast transient response.
The system stability and transient response are
controlled through the COMP pin. COMP is the
output of the internal transconductance error
amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The DC gain of the voltage feedback loop is:
OUT
FB
VEACSLOADVDC V
V
AGRA ×××=
Where AVEA is the error amplifier voltage gain,
GCS is the current sense transconductance and
RLOAD is the load resistor value.
The system has two poles of importance. One
is due to the compensation capacitor (C3) and
the output resistor of error amplifier while the
other is due to the output capacitor and the load
resistor. These poles are located at:
VEA
EA
PAC
G
f××
=32
1
π
LOAD
PRC
f××
=22 1
2
π
Where GEA is the error amplifier
transconductance.
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
332 1
1RC
fZ××
=
π
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero
due to the ESR and capacitance of the output
capacitor is located at:
ESR
ESR RC
f××
=22 1
π
In this case, a third pole set by the
compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
362 1
3RC
fP××
=
π
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
(where the feedback loop has unity gain) is
important.
Lower crossover frequencies result in slower
line and load transient responses, while higher
crossover frequencies could cause system
instability. A good standard is to set the
crossover frequency to approximately one-tenth
of the switching frequency. The switching
frequency for the MP1583 is 385KHz, so the
desired crossover frequency is around 38KHz.
Table 3 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
(Please reference Fig. 3 and Fig. 4)
VOUT C2 R3 C3 C6
2.5V 22F
Ceramic 3.9k 5.6nF None
3.3V 22F
Ceramic 4.7k 4.7nF None
5V 22F
Ceramic 7.5k 4.7nF None
12V 22F
Ceramic 16.9k 1.5nF None
2.5V 560F Al.
30m ESR 91k 1nF 150pF
3.3V 560F Al
30m ESR 120k 1nF 120pF
5V 470F Al.
30m ESR 100k 1nF 120pF
12V 220F Al.
30m ESR 169k 1nF 39pF
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To optimize the compensation components for
conditions not listed in Table 2, the following
procedure can be used.
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine R3
by the following equation:
FB
OUT
CSEA
C
V
V
GG
f2C2
3R ×
×
××π
=
Where fC is the desired crossover frequency
(which typically has a value no higher than
38KHz).
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of
the crossover frequency provides sufficient
phase margin. Determine C3 by the following
equation:
C
f3R2
4
3C ××π
>
Where R3 is the compensation resistor value.
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the 385KHz switching
frequency, or if the following relationship is valid:
2
f
R2C2
1S
ESR
<
××π
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine C6
by the equation:
3R
R2C
6C ESR
×
=
PCB Layout Guide
PCB layout is very important to achieve stable
operation. Please follow these guidelines and
take Figure2 and 3 for references.
1) Keep the path of switching current short
and minimize the loop area formed by Input
cap, high-side and low-side MOSFETs.
2) Keep the connection of low-side MOSFET
between SW pin and input power ground
as short and wide as possible.
3) Ensure all feedback connections are short
and direct. Place the feedback resistors
and compensation components as close to
the chip as possible.
4) Route SW away from sensitive analog
areas such as FB.
5) Connect IN, SW, and especially GND
respectively to a large copper area to cool
the chip to improve thermal performance
and long-term reliability. For single layer,
do not solder exposed pad of the IC
C1 D1
R1
C2
R4
SGND
PGND
5
1
2
3
4
8
7
6
FB
COMP
EN
SS/REFBS
IN
SW
GND
L1
C3
C6
R2
R3
SGND
C4 C5
Figure 2―PCB Layout (Single Layer)
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D1
C4 R3
R2
C5
R4
L1
SGND
PGND
1
2
3
4
8
7
6
5FB
COMP
EN
SS/REFBS
IN
SW
GND
C6
C2
C3
R1
C1
Top Layer
Bottom Layer
Figure 3―PCB Layout (Double Layer)
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator, the applicable
conditions of external BST diode are:
z VOUT=5V or 3.3V; and
z Duty cycle is high: D=
IN
OUT
V
V>65%
In these cases, an external BST diode is
recommended from the output of the voltage
regulator to BST pin, as shown in Fig.4
MP1583
SW
BST
C
L
BST
C
5V or 3.3V
OUT
External BST Diode
IN4148
+
Figure 4—Add Optional External Bootstrap
Diode to Enhance Efficiency
The recommended external BST diode is
IN4148, and the BST cap is 0.1~1µF.
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TYPICAL APPLICATION CIRCUITS
INPUT
4.75V to 23V
OUTPUT
3.3V
3A
C1
10μF/25V
CERAMIC
C2
22μF/10V
Murata
C3
4.7nF
C4
10nF
C6
NS
D1
C5
10nF
L1
15μH
R3
4.7kΩ
R2
10kΩ
R1
16.9kΩ
MP1583
BSIN
FB
SW
SS
GND COMP
EN
1
3
5
64
8
7
2
OPEN = AUTOMATIC
STARTUP
Figure 5—3.3V output 3A solution with Murata 22µF, 10V Ceramic Output Capacitor
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
MP1583 Rev. 3.1 www.MonolithicPower.com 12
6/20/2011 MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
PACKAGE INFORMATION
SOIC8E (EXPOSED PAD)
SEE DETAIL "A"
0.0075(0.19)
0.0098(0.25)
0.050(1.27)
BSC
0.013(0.33)
0.020(0.51)
SEATING PLANE
0.000(0.00)
0.006(0.15)
0.051(1.30)
0.067(1.70)
TOP VIEW
FRONT VIEW
SIDE VIEW
BOTTOM VIEW
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
0.089(2.26)
0.101(2.56)
0.124(3.15)
0.136(3.45)
RECOMMENDED LAND PATTERN
0.213(5.40)
0.063(1.60)
0.050(1.27)
0.024(0.61)
0.103(2.62)
0.138(3.51)
0.150(3.80)
0.157(4.00)
PIN 1 ID
0.189(4.80)
0.197(5.00)
0.228(5.80)
0.244(6.20)
14
85
0.016(0.41)
0.050(1.27)
0o-8o
DETAIL "A"
0.010(0.25)
0.020(0.50) x 45o
0.010(0.25) BSC
GAUGE PLANE
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP1583 Rev. 3.1 www.MonolithicPower.com 13
6/20/2011 MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
PDIP8
NOTE:
1) Control dimension is in inches. Dimension in bracket is millimeters.
0.021(0.533)
0.015(0.381)
0.100 BSC(2.540)
0.145(3.683)
0.134(3.404)
0.140(3.556)
0.120(3.048)
0.035 (0.889)
0.015 (0.381)
0.260 (6.604)
0.240 (6.096)
0.387 (9.830)
0.367 (9.322)
PIN 1 IDENT.
0.040 (1.016)
0.020 (0.508)
0.065 (1.650)
0.050 (1.270)
0.014 (0.356)
0.008 (0.200)
0.325(8.255)
0.300(7.620)
0.392(9.957)
0.332(8.433)
3°~11°
Le ad Be n d
