LT8616
1
8616f
For more information www.linear.com/LT8616
Typical applicaTion
FeaTures DescripTion
Dual 42V Synchronous
Monolithic Step-Down Regulator
with 6.5µA Quiescent Current
The LT
®
8616 is a high efficiency, high speed, dual synchro-
nous monolithic step-down switching regulator that con-
sumes only 6.5µA of quiescent current with both channels
enabled. Both channels contain all switches and necessary
circuitry to minimize the need for external components.
Low ripple Burst Mode operation enables high efficiency
down to very low output currents while minimizing output
ripple. A SYNC pin allows synchronization to an external
clock. Internal compensation with peak current mode
topology allows the use of small inductors and results in
fast transient response and good loop stability. The en-
able pins have accurate 1V thresholds and can be used to
program undervoltage lockout. Capacitors on the TR/SS
pins programs the output voltage ramp rate during start-
up while the PG pins signal when each output is within
10% of the programmed output voltage. The LT8616 is
available in a TSSOP package for high reliability.
applicaTions
n Wide Input Voltage Range: 3.4V to 42V
n 2.5A and 1.5A Buck Regulators with Separate Inputs
n Fast Minimum Switch On-Time: 35ns
n Ultralow Quiescent Current Burst Mode
®
Operation:
n 6.5µA IQ Regulating 12VIN to 5VOUT and 3.3VOUT
n Output Ripple < 15mV
n 180° Out of Phase Switching
n Adjustable and Synchronizable: 200kHz to 3MHz
n Accurate 1V Enable Pin Thresholds
n Internal Compensation
n Output Soft-Start and Tracking
n TSSOP Package: Output Stays at or Below Regulation
Voltage During Adjacent Pin Short or When a Pin Is
Left Floating
n Thermally Enhanced 28-Lead TSSOP Package
n Automotive and Industrial Supplies
n General Purpose Step-Down
L, LT , LT C , LT M , Burst Mode, Linear Technology and the Linear logo are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
BOOST1VIN1
EN/UV1
INTVCC
RT
PG1
PG2
SYNC/MODE
TR/SS1
TR/SS2
SW1
LT8616
GND
FB1
BIAS
BOOST2
SW2
8616 TA01a
FB2
0.1µF
0.1µF
VOUT1
5V, 1.5A
10pF
2 x 47µF
4.7µF
VIN
12V
VOUT2
3.3V, 2.5A
10nF
10µH
4.7µH
1M
316k
1M
1M
187k
56.2k
VIN2
EN/UV2
4.7µF 5.6pF 47µF
1µF
5V, 3.3V, 700kHz Step-Down Converter Efficiency
LOAD (mA)
0.01
30
EFFICIENCY (%)
90
80
70
60
50
40
100
1 10 100 10000.1
8616 TA01b
CH2, 3.3VOUT
CH1, 5VOUT
VIN1 = VIN2 = 12V
fSW = 700kHz
LT8616
2
8616f
For more information www.linear.com/LT8616
pin conFiguraTionabsoluTe MaxiMuM raTings
VIN1, VIN2, EN/UV1, EN/UV2, PG1, PG2 .....................42V
BIAS .......................................................................... 30V
BST1 Above SW1, BST2 Above SW2, FB1, FB2,
TR/SS1, TR/SS2 .....................................................4V
SYNC/MODE ...............................................................6V
Operating Junction Temperature Range (Note 2)
LT8616E ............................................. –40°C to 125°C
LT8616I .............................................. –40°C to 125°C
LT8616H ............................................ –40°C to 150°C
Storage Temperature Range .................. –60°C to 150°C
1
2
3
4
5
6
7
8
9
10
11
12
13
14
TOP VIEW
FE PACKAGE
28-LEAD PLASTIC TSSOP
28
27
26
25
24
23
22
21
20
19
18
17
16
15
EN/UV2
PG2
SW2
SW2
SW2
BOOST2
NC
BOOST1
SW1
SW1
PG1
TR/SS1
FB1
FB1
TR/SS2
FB2
FB2
NC
VIN2
NC
BIAS
INTVCC
NC
VIN1
NC
SYNC/MODE
EN/UV1
RT
29
GND
TJMAX = 150°C, θJA = 30°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8616EFE#PBF LT8616EFE#TRPBF LT8616FE 28-Lead Plastic TSSOP –40 to 125°C
LT8616IFE#PBF LT8616IFE#TRPBF LT8616FE 28-Lead Plastic TSSOP –40 to 125°C
LT8616HFE#PBF LT8616HFE#TRPBF LT8616FE 28-Lead Plastic TSSOP –40 to 150°C
Consult LT C Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LT C Marketing for information on nonstandard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
(Note 1)
LT8616
3
8616f
For more information www.linear.com/LT8616
elecTrical characTerisTics
PARAMETER CONDITIONS MIN TYP MAX UNITS
Common
Quiescent Current EN/UV1 = EN/UV2 = 0V, Current from VIN1
l
1.7
1.7
4.0
8.0
µA
µA
EN/UV1 = EN/UV2 = 2V, SYNC = 0V (Burst Mode), Not Switching,
Current from VIN1
l
3.0
3.0
5.0
12.0
µA
µA
EN/UV1 = EN/UV2 = 2V, SYNC = 3V (Pulse-Skipping Mode), Not
Switching, Current from BIAS or VIN1
l0.5 1.0 mA
FB Voltage VIN = 6V, Load = 0.5A
l
782
778
790
790
798
802
mV
mV
FB Voltage Line Regulation VIN = 4V to 25V, Load = 0.5A 0.005 %/V
FB Pin Input Current FB = 0.79V –20 20 nA
EN/UV Pin Threshold Rising l0.97 1.03 1.09 V
EN/UV Pin Hysteresis 50 mV
EN/UV Pin Current EN/UV = 2V –20 20 nA
PG Upper Threshold from VFB FB Rising l6 10 13 %
PG Lower Threshold from VFB FB Falling l–6 –10 –13 %
PG Hysteresis 1 %
PG Leakage PG = 3.3V –100 100 nA
PG Pull-Down Resistance PG = 0.1V 350 Ω
TR/SS Source Current 1 2 3 µA
TR/SS Pull-Down Resistance TR/SS = 0.1V 250 Ω
BIAS Pin Current Consumption VOUT1 = 3.3V, Load1 = 0.5A, VOUT2 = 3.3V, Load2 = 0.5A, fSW = 1MHz 7 mA
Oscillator Frequency RT = 14.7kΩ
RT = 37.4kΩ
RT = 221kΩ
l
l
l
1.85
900
160
2.05
1000
200
2.25
1100
240
MHz
kHz
kHz
SYNC Threshold SYNC Falling
SYNC Rising
0.4
2.4
V
V
SYNC Pin Current SYNC = 3V –100 100 nA
Channel 1
Minimum VIN1 Voltage l3.0 3.4 V
Supply Current in Regulation VIN = 6V, VOUT1 = 3.3V, Load = 100µA
VIN = 6V, VOUT1 = 3.3V Load = 1mA
80
620
110
910
µA
µA
SW1 Minimum On-Time Load = 0.25A, Pulse-Skipping Mode l20 35 55 ns
SW1 Top NMOS On-Resistance 310 mΩ
SW1 Peak Current Limit (Note 3) l3.2 4.2 5.2 A
SW1 Bottom NMOS On-Resistance 190 mΩ
SW1 Valley Current Limit l1.5 2.0 3.0 A
SW1 Leakage Current VIN1 = 42V, VSW1 = 0V, 42V –2 2 µA
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C.
LT8616
4
8616f
For more information www.linear.com/LT8616
PARAMETER CONDITIONS MIN TYP MAX UNITS
Channel 2
Minimum VIN1 Voltage to Use
Channel2
l3.0 3.4 V
Supply Current in Regulation VIN = 6V, VOUT2 = 3.3V, Load2 = 100µA
VIN = 6V, VOUT2 = 3.3V Load2 = 1mA
80
620
110
910
µA
µA
SW2 Minimum On-Time Load = 0.25A, Pulse-Skipping Mode l20 35 55 ns
SW2 Top NMOS On-Resistance 145 mΩ
SW2 Peak Current Limit (Note 3) l4.5 5.5 6.5 A
SW2 Bottom NMOS On-Resistance 120 mΩ
SW2 Valley Current Limit l2.5 3.5 4.5 A
SW2 Leakage Current VIN2 = 42V, VSW2 = 0V, 42V –2 2 µA
elecTrical characTerisTics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT8616E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT8616I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8616H is guaranteed over the full –40°C to
150°C operating junction temperature range.
Note 3: Current limit guaranteed by design and/or correlation to static test.
Slope compensation reduces current limit at higher duty cycle.
Note 4: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified maximum operating junction temperature may impair device
reliability. See High Temperature Considerations section.
LT8616
5
8616f
For more information www.linear.com/LT8616
Typical perForMance characTerisTics
Efficiency at 3.3VOUT Efficiency at 3.3VOUT No Load Supply Current
No Load Supply Current
BIAS Current
vs Switching Frequency EN/UV Threshold
Efficiency at 5VOUT Efficiency at 5VOUT Efficiency vs Frequency
LOAD (A)
0
80
EFFICIENCY (%)
98
96
94
92
90
88
86
84
82
100
0.25 1.5 1.75 2.0 2.25 2.50.5 0.75 1.0
8616 G01
1.25
CH2, 24VIN
CH2, 12VIN
CH1, 24VIN
CH1, 12VIN
fSW = 700kHz
LOAD (mA)
0.01
30
EFFICIENCY (%)
95
90
85
80
75
70
65
60
55
50
45
40
35
100
0.1 100 10001
8616 G02
10
fSW = 700kHz
CH2, 24VIN
CH2, 12VIN
CH1, 24VIN
CH1, 12VIN
FREQUENCY (MHz)
0.2
85
EFFICIENCY (%)
95
94
93
92
91
90
89
88
87
86
1.4 2.6 3.01.8
8616 G03
2.21.00.6
VIN1 = VIN2 = 12V
CH2, 3.3V, 1.5A
CH1, 5V, 1A
LOAD (A)
0
80
EFFICIENCY (%)
100
98
96
94
92
90
88
86
84
82
0.75 1.0 1.25 2.0 2.25 2.51.5
8616 G04
1.750.50.25
CH2, 24VIN
CH2, 12VIN
CH1, 24VIN
CH1, 12VIN
fSW = 700kHz
LOAD (mA)
0.01
30
EFFICIENCY (%)
100
90
95
85
80
75
70
65
60
55
50
45
40
35
0.1 1 10 1000
8616 G05
100
CH2, 24VIN
CH2, 12VIN
CH1, 24VIN
CH1, 12VIN
fSW = 700kHz
VIN (V)
0
0
INPUT CURRENT (µA)
10
9
8
7
6
5
4
3
2
1
5 10 15 20 25 30 40
8616 G06
35
VOUT1 = 3.3V
VOUT2 = 5V
TEMPERATURE (°C)
–50
0
SUPPLY CURRENT (µA)
30
25
20
15
10
5
–25 0 25 50 75 100 150
8616 G07
125
VIN1 = VIN2 = 12V
VOUT1 = 5V
VOUT2 = 3.3V
IN REGULATION
FREQUENCY (MHz)
0
0
BIAS CURRENT (mA)
18
16
14
12
10
8
6
4
2
0.5 1.0 1.5 2.0 3.0
8616 G08
2.5
VIN1 = VIN2 = 12V
VOUT1 = 5V
LOAD1 = 1A
VOUT2 = 3.3V
LOAD2 = 1A
TEMPERATURE (°C)
–50
0.92
EN THRESHOLD (V)
1.1
1.08
1.06
1.04
0.98
0.96
0.94
1.02
1.0
–25 50250 75 100 150
8616 G09
125
EN FALLING
EN RISING
LT8616
6
8616f
For more information www.linear.com/LT8616
Typical perForMance characTerisTics
Power-Good Thresholds Soft-Start Tracking Switching Frequency
Switching Period vs RTMinimum On-Time Minimum Off-Time
FB Voltage vs Temperature Line Regulation vs VIN1 Load Regulation
TEMPERATURE (°C)
–50
774
VOLTAGE (mV)
802
800
798
796
794
792
790
788
786
784
782
780
778
776
806
804
75 100 125 150–25 0 25
8616 G10
50
VIN1 (V)
0
–0.1
VOLTAGE (%)
0.06
0.04
0.02
0
–0.02
–0.04
–0.06
–0.08
0.1
0.08
25 30 35 405 10
8616 G11
15 20
VIN1 = VIN2
LOAD (A)
0
–0.2
CHANGE IN VOUT (%)
0.1
0.05
0
–0.05
–0.1
–0.15
0.2
0.15
1.75 2.0 2.25 2.50.25 1.00.750.5
8616 G12
1.25 1.5
FB2
FB1
TEMPERATURE (°C)
–50
–15
PG THRESHOLD RELATIVE TO FB (%)
5
0
–5
–10
15
10
75 100 125 150–25 0
8616 G13
25 50
PG HIGH FALLING
PG LOW RISING
PG HIGH RISING
PG LOW FALLING
SS/TR VOLTAGE (mV)
0
0
FB VOLTAGE (mV)
700
300
400
500
600
200
100
900
800
700 800 900 1000100 200 300
8616 G14
400 500 600
TEMPERATURE (°C)
–50
1.85
FREQUENCY (MHz)
2.15
2.1
2.05
2
1.95
1.9
2.25
2.2
75 100 125 150–25 0
8616 G15
25 50
RT = 14.7kΩ
SWITCHING PERIOD (µs)
0
0
RT RESISTOR (kΩ)
200
180
160
140
120
100
80
60
40
20
240
220
2345
8616 G16
1
TEMPERATURE (°C)
–50
20
TIME (ns)
45
40
35
30
25
50
75 100 125 150
8616 G17
–25 0 25 50
TEMPERATURE (°C)
–50
50
TIME (ns)
85
80
75
70
65
60
55
90
75 100 125 150
8616 G18
–25 0 25 50
LT8616
7
8616f
For more information www.linear.com/LT8616
Typical perForMance characTerisTics
Current Limit vs Temperature
Switch Resistance
vs Temperature Dropout Voltage vs Load
Start-Up Dropout (CH1, 5V) Start-Up Dropout (CH2, 3.3V)
Burst Frequency vs Load Minimum Load for Full Frequency
Top FET Current Limit
vs Duty Cycle
LOAD CURRENT (mA)
0
0
SWITCHING FREQUENCY (kHz)
700
600
500
400
300
200
100
800
50 200 250 300 350 400100
8616 G19
150
VIN1 = VIN2 =12V
fSW = 700kHz
SYNC = 0V
CH2, 3.3V
CH1, 5V
CH2, 3.3V
CH1, 5V
VIN (V)
0
0
LOAD CURRENT (mA)
70
60
50
40
30
20
10
80
5 20 25 30 35 4010
8616 G20
15
fSW = 700kHz
SYNC = 3V
CH2
CH1
DUTY CYCLE (%)
0
2.0
CURRENT LIMIT (A)
5.0
4.5
4.0
3.5
3.0
2.5
5.5
20 40 60 80 100
8616 G21
TEMPERATURE (°C)
–50
0
CURRENT LIMIT (A)
5
4
3
2
1
6
75 100 125 150–25 0 25
8616 G22
50
CH2, VALLEY
CH2, 0% DC PEAK
CH1, VALLEY
CH1, 0% DC PEAK
TEMPERATURE (°C)
–50
0
RESISTANCE (mΩ)
500
400
300
200
100
600
75 100 125 150–25 0 25
8616 G23
50
CH2, BOTTOM
CH2, TOP
CH1, BOTTOM
CH1, TOP
LOAD (A)
0
0
DROPOUT VOLTAGE (mV)
900
800
700
600
500
400
300
200
100
1000
1 1.5 2 2.50.5
8616 G24
CH2, 5V
CH1, 5V
fSW = 2MHz
100ms/DIV
VIN1
1V/DIV
VOUT1
1V/DIV
8616 G26
RLOAD1 = 3.33Ω (1.5A)
100ms/DIV
VIN1 AND VIN2
1V/DIV
VOUT2
1V/DIV
8616 G27
VIN1 = VIN2
RLOAD2 = 1.32Ω (2.5A)
Start-Up Dropout (CH2, 3.3V)
100ms/DIV
VIN2
1V/DIV
VOUT2
1V/DIV
8616 G28
VIN1 = 6V
RLOAD2 = 1.32Ω (2.5A)
LT8616
8
8616f
For more information www.linear.com/LT8616
Typical perForMance characTerisTics
CCM Burst Mode DCM
CH1 CCM, CH2 CCM CH1 CCM, CH2 Burst Mode CH1 Shorted, CH2 CCM
Transient CH1, 5V Transient CH2, 3.3V
20µs/DIV
VOUT1 (AC)
200mV/DIV
IL1
500mA/DIV
8616 G29
VIN1 = 12V
VOUT1 = 5V
L1 = 10µH
COUT1 = 47µF
CFF = 5.6pF
20µs/DIV 8616 G30
VOUT2 (AC)
50mV/DIV
IL2
1A/DIV
VIN2 = 12V
VOUT2 = 3.3V
L2 = 4.7µH
COUT2 = 2 x 47µF
CFF2 = 10pF
1µs/DIV
VSW
5V/DIV
IL
500mA/DIV
8616 G31
12VIN TO 5VOUT AT 500mA
1µs/DIV
VSW
5V/DIV
IL
500mA/DIV
8616 G32
12VIN TO 5VOUT AT 50mA
SYNC = 0V
1µs/DIV
VSW
5V/DIV
IL
500mA/DIV
8616 G33
12VIN TO 5VOUT AT 50mA
SYNC = 3V
500ns/DIV
SW1
5V/DIV
SW2
5V/DIV
8616 G34
VIN = 12V
CH1 = 5V, 1A
CH2 = 3.3V, 1A
SYNC = 0V
1µs/DIV
SW1
5V/DIV
SW2
5V/DIV
8616 G35
VIN = 12V
CH1 = 5V, 1A
CH2 = 3.3V, 0.1A
SYNC = 0V
5µs/DIV
SW1
5V/DIV
SW2
5V/DIV
8616 G36
VIN = 12V
CH1 = 0V SHORT
CH2 = 3.3V, 1A
SYNC = 0V
VIN1 UVLO
TEMPERATURE (°C)
–50
2.0
VIN1 (V)
3.4
3.2
3.0
2.8
2.6
2.4
2.2
3.6
75 100 125 150–25 0 25 50
8616 G25
LT8616
9
8616f
For more information www.linear.com/LT8616
pin FuncTions
BIAS: The BIAS pin supplies the internal regulator when tied
to a voltage higher than 3.1V. For output voltages of 3.3V
and above this pin should be tied to the appropriate VOUT.
Connect a 1µF bypass capacitor to this pin if it is connected
to a supply other than VOUT1 or VOUT2. Ground if unused.
BOOST1, BOOST2: The BOOST pins are used to provide
drive voltages, higher than the input voltage, to the internal
topside power switches. Place 0.1µF capacitors between
BOOST and its corresponding SW pin as close as possible
to the IC. BOOST nodes should be kept small on the PCB
for good performance.
EN/UV1, EN/UV2: The EN/UV pins are used to indepen-
dently disable each channel when pulled low and enable
when pulled high. The hysteretic threshold voltage is 1.03V
going up and 0.98V going down. Tie to VIN supply if the
shutdown feature is not used. External resistor dividers
from VIN can be used to program thresholds below which
each channel is disabled. Don’t float these pins.
FB1, FB2: The FB pins are regulated to 0.790V. Connect
the feedback resistor divider taps to the FB pins. Also
connect phase lead capacitors between FB pins and VOUT
nodes. Typical phase lead capacitors are 1.5pF to 10pF.
GND: The GND pins and exposed pad must be con-
nected to the negative terminal of the input capacitors
and soldered to the PCB in order to lower the thermal
resistance.
INTVCC: The INTVCC pin provides power to internal gate
drivers and control circuits. INTVCC current will be sup-
plied from BIAS if VBIAS > 3.1V, otherwise current will be
drawn from VIN1. Decouple this pin to ground with at least
a 1μF low ESR ceramic capacitor. Do not load the INTVCC
pin with external circuitry.
NC: The NC pins have no internal connection. Float NC
pins to increase fault tolerance or connect to ground to
facilitate PCB layout.
PG1, PG2: The PG pins are the open-drain outputs of the
internal power good comparators. Each channel's PG pin
remains low until the respective FB pin is within ±10% of
the final regulation voltage and there are no fault conditions.
RT: A resistor is tied between RT and ground to set the
switching frequency.
SW1, SW2: The SW pins are the outputs of each chan-
nel's internal power switches. Connect these pins to the
inductors and boost capacitors. SW nodes should be kept
small on the PCB for good performance.
SYNC/MODE: Ground the SYNC/MODE pin for low ripple
Burst Mode operation at low output loads. Tie to a clock
source for synchronization to an external frequency. Apply
a DC voltage of 2.4V or higher or tie to INTVCC for pulse-
skipping mode. When in pulse-skipping mode, the IQ will
increase to several hundred μA. Channel 1 will align its
positive switching edge to the positive edge of the external
clock and channel 2 will align its positive switching edge
to the negative external clock edge. Do not float this pin.
TR/SS1, TR/SS2: The TR/SS pins are used to soft-start
the two channels, to allow one channel to track the other
output, or to allow both channels to track another output.
For tracking, tie a resistor divider to the TR/SS pin from
the tracked output. For soft-start, tie a capacitor to TR/
SS. Internal 2μA pull-up currents from INTVCC charge
soft-start capacitors to create voltage ramps. A TR/SS
voltage below 0.79V forces the LT8616 to regulate the
corresponding FB pins to equal the TR/SS pin voltage.
When TR/SS voltages are above 0.79V, the tracking func-
tion is disabled and the internal reference resumes control
of the error amplifiers. TR/SS pins are individually pulled
to ground with internal 250Ω MOSFETs during shutdown
and fault conditions; use series resistors if driving from
a low impedance output.
VIN1: VIN1 supplies current to the LT8616's internal circuitry
and to channel 1's topside power switch. This pin must
be locally bypassed. Be sure to place the positive terminal
of the input capacitor as close as possible to the pin, and
the negative capacitor terminal as close as possible to the
GND pins. VIN1 must be connected to 3.4V or above even
if only channel 2 is in use.
VIN2: VIN2 supplies current to internal channel 2's topside
power switch. This pin must be locally bypassed. Be sure
to place the positive terminal of the input capacitor as close
as possible to the pin, and the negative capacitor terminal
as close as possible to the GND pins. Please note VIN1
must be 3.4V or above to operate channel 2.
LT8616
10
8616f
For more information www.linear.com/LT8616
block DiagraM
+
–
+
–
+
+
–
INTERNAL
REFERENCE AND
3.3V REGULATOR
DRIVER
SWITCH
LOGIC
AND ANTI-
SHOOT
THROUGH
BURST
LOGIC
+
+
–
OSCILLATOR
200kHz TO 3MHz
BURST
LOGIC
VIN1
VIN2
R5
R6
R7
R8
EN/UV2
EN/UV1
SHDN1
SHDN2
1.03V
1.03V
INTVCC
VC1
INTVCC
CVCC
CFF1
CFF2
BIAS
BOOST1
SW1
GND
VIN2
BOOST2
CBST2
L2
SW2
GND
8616 BD
GND
CBST1
VIN1 VIN1
VOUT1
VIN2
VOUT2
COUT2
CIN2
COUT1
CIN1
L1
INTVCC
INTVCC
VC2
SHDN2
TR/SS2
FB2
R3
R4
R1
R2
PG2
SYNC/MODE
RT
TR/SS1
FB1
2µA
2µA
PG1
CSS1
RT
TSD
VIN1 UVLO
SHDN2
VIN1 UVLO
INTVCC UVLO
TSD
SHDN1
VIN1 UVLO
INTVCC UVLO
TSD
SHDN1
±10%
0.79V
VIN1 UVLO
INTVCC
TSD
CSS2
VOUT2
VOUT1
ERROR
AMP
ERROR
AMP
SLOPE COMP
SLOPE COMP
DRIVER
SWITCH
LOGIC
AND ANTI-
SHOOT
THROUGH
±10%
0.79V
LT8616
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Foreword
The LT8616 is a dual monolithic step down regulator. The
two channels differ in maximum current and input range.
The following sections describe the operation of channel
1 and common circuits. They will highlight channel 2 dif-
ferences and interactions only when relevant. To simplify
the application, both VIN1 and VIN2 are assumed to be con-
nected to the same input supply. However, note that VIN1
must be greater than 3.4V for either channel to operate.
Operation
The LT8616 is a dual monolithic, constant frequency, peak
current mode step-down DC/DC converter.
An oscillator, with frequency set using a resistor on the RT
pin, turns on the internal top power switch at the beginning
of each clock cycle. Current in the inductor then increases
until the top switch current comparator trips and turns off
the top power switch. The peak inductor current at which
the top switch turns off is controlled by the voltage on the
internal VC node. The error amplifier servos the VC node
by comparing the voltage on the FB pin with an internal
0.790V reference. When the load current increases it causes
a reduction in the feedback voltage relative to
the reference,
causing the error amplifier to raise the VC voltage until the
average inductor current matches the new load current.
When the top power switch turns off, the synchronous
power switch turns on until the next clock cycle begins or
inductor current falls to zero. If overload conditions result
in more than the valley current limit flowing through the
bottom switch, the next clock cycle will be delayed until
current returns to a safe level.
If either EN/UV pin is low, the corresponding channel is
shut down. If both EN/UV pins are low, the LT8616 is
fully shut down and draws 1.7µA from the input supply.
When the EN/UV pins are above 1.03V, the corresponding
switching regulators will become active. 1.3μA is supplied
by VIN1 to common bias circuits for both channels.
Each channel can independently enter Burst Mode opera-
tion to optimize efficiency at light load. Between bursts,
all circuitry associated with controlling the output switch
is shut down, reducing the channel's contribution to in-
put supply current. In a typical application, 6.5μA will be
consumed from the input supply when regulating both
channels with no load. Ground the SYNC/MODE pin for
Burst Mode operation or apply a DC voltage above 2.4V
to use pulse-skipping mode. If a clock is applied to the
SYNC/MODE pin, both channels will synchronize to the
external clock frequency and operate in pulse-skipping
mode. While in pulse-skipping mode the oscillator operates
continuously and SW transitions are aligned to the clock.
During light loads, switch pulses are skipped to regulate
the output and the quiescent current per channel will be
several hundred µA.
To improve efficiency across all loads, supply current to
internal circuitry can be sourced from the BIAS pin when
biased at 3.1V or above. Otherwise, the internal circuitry
will draw current exclusively from VIN1. The BIAS pin
should be connected to the lowest VOUT programmed at
3.3V or above.
Comparators monitoring the FB pin voltage will pull the
corresponding PG pin low if the output voltage varies
more than ±10% (typical) from the regulation voltage or
if a fault condition is present.
Tracking soft-start is implemented by providing constant
current via the TR/SS pin to an external soft-start capaci-
tor to generate a voltage ramp. FB voltage is regulated to
the voltage at the TR/SS pin until it exceeds 0.790V; FB
is then regulated to the 0.790V reference. Soft-start also
reduces the valley current limit to avoid inrush current
during start-up. The SS capacitor is reset during shutdown,
VIN1 undervoltage, or thermal shutdown.
Channel 1 is designed for 1.5A load, whereas channel 2
is designed for 2.5A load. Channel 1 has a minimum VIN1
requirement of 3.4V, but channel 2 can operate with no
minimum VIN2 provided that the minimum VIN1 has been
satisfied.
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Achieving Ultralow Quiescent Current
To enhance efficiency at light loads, the LT8616 operates
in low ripple Burst Mode operation, which keeps the out-
put capacitor charged to the desired output voltage while
minimizing the input quiescent current and output voltage
ripple. 1.7μA is supplied by VIN1 to common bias circuits.
In Burst Mode operation the LT8616 delivers single small
pulses of current to the output capacitor followed by sleep
periods where the output power is supplied by the output
capacitor. While in sleep mode the LT8616 consumes 3μA.
As the output load decreases, the frequency of single cur-
rent pulses decreases (see Figure 1a) and the percentage
of time that the LT8616 is in sleep mode increases, result-
ing in much higher light load efficiency than for typical
converters. By maximizing the time between pulses, the
converter quiescent current approaches 6.5µA for a typi-
cal application when there is no output load. Therefore,
to optimize the quiescent current performance at light
loads, the current in the feedback resistor divider must
be minimized as it appears to the output as load current.
While in Burst Mode operation the current limit of the top
switch is approximately 400mA for channel 1 (600mA
for channel 2) resulting in output voltage ripple shown in
Figure 2. Increasing the output capacitance will decrease
the output ripple. As load increases from zero the switch-
ing frequency will increase but only up to the switching
frequency programmed by the resistor at the RT pin as
shown in Figure 1a. The output load at which the LT8616
reaches the programmed frequency varies based on input
voltage, output voltage, and inductor value.
For some applications it is desirable to select pulse-skipping
mode to maintain full switching frequency at lower output
load (see Figure 1b). See Pulse-Skipping Mode section.
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin (R1 to R2 for channel 1,
R3 to R4 for channel 2). Choose the resistor values ac-
cording to:
R1=R2
V
OUT1
0.790V –1
⎛
⎝
⎜⎞
⎠
⎟
(1)
applicaTions inForMaTion
Figure 1a. Frequency vs Load in Burst Mode Operation
Figure 1b. Minimum Load for Full
Frequency in Pulse-Skipping Mode
Figure 2. Burst Mode Operation
LOAD CURRENT (mA)
0
0
SWITCHING FREQUENCY (kHz)
800
700
600
500
400
300
200
100
200 350 400250
8616 F01a
30015010050
VIN1 = VIN2 = 12V
fSW = 700kHz
SYNC = 0V
CH2, 3.3V
CH1, 5V
VIN (V)
0
0
LOAD CURRENT (mA)
80
70
60
50
40
30
20
10
20 35 4025
8616 F01b
3015105
fSW = 700kHz
SYNC = 3V
CH2, 3.3V
CH1, 5V
5µs/DIV
VOUT (AC)
5mV/DIV
IL
200mA/DIV
8616 F02
CH1 = 5V, 25mA
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Figure 3. Switching Frequency vs RT
Reference designators refer to the Block Diagram. 1% resis-
tors are recommended to maintain output voltage accuracy.
If low input quiescent current and good light-load efficiency
are desired, use large resistor values for the FB resistor
divider. The current flowing in the divider acts as a load
current and will increase the no-load input current to the
converter, which is approximately:
IQ=3µA +VOUT1
R1+R2
⎛
⎝
⎜⎞
⎠
⎟VOUT1
V
IN1
⎛
⎝
⎜⎞
⎠
⎟1
η
⎛
⎝
⎜⎞
⎠
⎟
(2)
where 3µA is the quiescent current, the second term is
the current in the feedback divider reflected to the input
of channel 1 operating at its light load efficiency η. For a
3.3V application with R1 = 1M and R2 = 316k, the feedback
divider draws 2.5µA. With VIN = 12V and η = 70%, this
adds 1µA to the 3µA quiescent current resulting in 4µA
no-load current from the 12V supply.
Substitute R1 and R2 with R3 and R4 in the above equa-
tion if VIN1 and VIN2 are connected to the same voltage.
Assuming channel 2 feedback divider contributes 2.5µA
to the quiescent current, then the total quiescent current
is 6.5µA.
For a typical FB resistor of 1MΩ, a 1.5pF to 10pF phase-
lead capacitor should be connected from VOUT to FB.
Setting the Switching Frequency
The LT8616 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 3MHz
by using a resistor tied from the RT pin to ground. Table 1
and Figure 3 show the necessary RT value for a desired
switching frequency.
The RT resistor required for a desired switching frequency
can be calculated using:
RT=
0.6
fSW2+
42.6
fSW
– 6.
1
(3)
where RT is in kΩ and fSW is the desired switching fre-
quency in MHz.
Table 1. SW Frequency vs RT Value
fSW (MHz) RT kΩ) fSW (MHz) RT kΩ)
0.2 221 1.6 20.5
0.3 143 1.8 17.8
0.4 105 2.0 15.4
0.5 80.6 2.05 14.7
0.6 66.5 2.2 13.3
0.7 56.2 2.4 11.8
0.8 47.5 2.6 10.3
1.0 37.4 2.8 9.31
1.2 29.4 3.0 8.25
1.4 24.3
The two channels of the LT8616 operate 180° out of
phase to avoid aligned switching edge noise and input
current ripple.
Operating Frequency Selection and Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, and input voltage range. The
advantage of high frequency operation is that smaller induc-
tor and capacitor values may be used. The disadvantages
are lower efficiency and a smaller input voltage range for
full frequency operation.
SWITCHING PERIOD (µs)
0
0
RT RESISTOR (kΩ)
120
140
160
180
200
220
100
80
60
40
20
240
12345
8616 F03
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The highest switching frequency (fSW(MAX)) for a given
application can be calculated as follows:
fSW(MAX) =
V
OUT
+V
SW(BOT)
tON(MIN) VIN – VSW(TOP) +VSW(BOT)
( )
(4)
where VIN is the typical input voltage, VOUT is the output
voltage, VSW(TOP) and VSW(BOT) are the internal switch
drops (~0.53V, ~0.38V, respectively at maximum load
for channel 1 and ~0.78V, ~0.48V for channel 2) and
tON(MIN) is the minimum top switch on-time of 55ns (see
the Electrical Characteristics). This equation shows that a
lower switching frequency is necessary to accommodate a
high VIN/VOUT ratio. Choose the lower frequency between
channel 1 and 2.
For transient operation, VIN may go as high as the absolute
maximum rating of 42V regardless of the RT value, how-
ever the LT8616 will reduce switching frequency on each
channel independently as necessary to maintain control
of inductor current to assure safe operation.
The LT8616 is capable of a maximum duty cycle of greater
than 99%, and the VIN to VOUT dropout is limited by the
RDS(ON) of the top switch. In this mode the channel that
enters dropout skips switch cycles, resulting in a lower
than programmed switching frequency.
For applications that cannot allow deviation from the pro-
grammed switching frequency at low VIN/VOUT ratios, use
the following formula to set switching frequency:
VIN(MIN) =
V
OUT
+V
SW(BOT)
1– fSW •tOFF(MIN)
– VSW(BOT) +VSW(TOP)
(5)
where VIN(MIN) is the minimum input voltage without
skipped cycles, VOUT is the output voltage, VSW(TOP) and
VSW(BOT) are the internal switch drops (~0.53V, ~0.38V,
respectively at maximum load for channel 1 and ~0.78V,
~0.48V for channel 2), fSW is the switching frequency (set
by RT), and tOFF(MIN) is the minimum switch off-time. Note
that higher switching frequency will increase the minimum
input voltage below which cycles will be dropped to achieve
higher duty cycle.
Note there is no minimum VIN2 voltage requirement as it
does not supply the internal common bias circuits, mak-
ing channel 2 uniquely capable of operating from very
low input voltages.
Inductor Selection and Maximum Output Current
The LT8616 is designed to minimize solution size by
allowing the inductor to be chosen based on the output
load requirements of the application. During overload or
short-circuit conditions the LT8616 safely tolerates opera-
tion with a saturated inductor through the use of a high
speed peak-current mode architecture.
A good first choice for the inductor value is:
L1=
V
OUT1
+V
SW1(BOT)
fSW
•1.6
L2 =VOUT2 +VSW2(BOT)
fSW
(6a)
(6b)
where fSW is the switching frequency in MHz, VOUT is
the output voltage, VSW(BOT) is the bottom switch drop
(~0.38V, ~0.48V) and L is the inductor value in μH. To
avoid overheating and poor efficiency, an inductor must
be chosen with an RMS current rating that is greater than
the maximum expected output load of the application. In
addition, the saturation current (typically labeled ISAT) rat-
ing of the inductor must be higher than the load current
plus 1/2 of in inductor ripple current:
IL(PEAK) =ILOAD(MAX) +
1
2
ΔIL
(7)
where ∆IL is the inductor ripple current as calculated in
equation 9 and ILOAD(MAX) is the maximum output load
for a given application.
As a quick example, an application requiring 1A output
should use an inductor with an RMS rating of greater than
1A and an ISAT of greater than 1.3A. During long duration
overload or short-circuit conditions, the inductor RMS
rating requirement is greater to avoid overheating of the
inductor. To keep the efficiency high, the series resistance
(DCR) should be less than 0.04Ω, and the core material
should be intended for high frequency applications.
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The LT8616 limits the peak switch current in order to protect
the switches and the system from overload faults. The top
switch current limit (ILIM) is 4.2A at 0% duty cycle and
decreases linearly to 2.9A at DC = 80% (channel 2 current
limit are 5.5A at 0% duty cycle and 3.7A at DC = 80%).
The inductor value must then be sufficient to supply the
desired maximum output current (IOUT(MAX)), which is a
function of the switch current limit (ILIM) and the ripple
current.
IOUT(MAX) =ILIM –
ΔI
L
2
(8)
The peak-to-peak ripple current in the inductor can be
calculated as follows:
ΔIL=VOUT
L•fSW
•1– VOUT
VIN(MAX)
⎛
⎝
⎜⎞
⎠
⎟
(9)
where fSW is the switching frequency of the LT8616, and
L is the value of the inductor. Therefore, the maximum
output current that the LT8616 will deliver depends on
the switch current limit, the inductor value, and the input
and output voltages.
Each channel has a secondary valley current limit. After
the top switch has turned off, the bottom switch carries
the inductor current. If for any reason the inductor current
is too high, the bottom switch will remain on, delaying the
top switch turning on until the inductor current returns
to a safe level. This level is specified as the valley Current
Limit, and is independent of duty cycle. Maximum output
current in the application circuit is limited to this valley
current plus one half of the inductor ripple current.
In most cases current limit is enforced by the top switch.
The bottom switch limit controls the inductor current when
the minimum on-time condition is violated (high input
voltage, high frequency or saturated inductor).
The bottom switch current limit is designed to avoid any
contribution to the maximum rated current of the LT8616.
The optimum inductor for a given application may differ
from the one indicated by this design guide. A larger value
inductor provides a higher maximum load current and
reduces the output voltage ripple. For applications requir-
ing smaller load currents, the value of the inductor may
be lower and the LT8616 may operate with higher ripple
current. This allows use of a physically smaller inductor,
or one with a lower DCR resulting in higher efficiency. Be
aware that low inductance may result in discontinuous
mode operation, which further reduces maximum load
current.
For more information about maximum output current
and discontinuous operation, see Linear Technology’s
Application Note 44.
Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5),
a minimum inductance is required to avoid sub-harmonic
oscillation. See Application Note 19.
Table 2. Inductor Manufacturers
VENDOR URL
Coilcraft www.coilcraft.com
Sumida www.sumida.com
Toko www.toko.com
Würth Elektronik www.we-online.com
Vishay www.vishay.com
Input Capacitor
Bypass the input of the LT8616 circuit with a ceramic ca-
pacitor of X7R or X5R type placed as close as possible to
the VIN and GND pins. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 2.2μF to 10μF ceramic capacitor is adequate to
bypass the LT8616 and will easily handle the ripple current.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input sup-
ply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT8616 and to force this very high frequency
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switching current into a tight local loop, minimizing EMI.
A 2.2μF capacitor is capable of this task, but only if it is
placed close to the LT8616 (see the PCB Layout section).
A second precaution regarding the ceramic input capacitor
concerns the maximum input voltage rating of the LT8616.
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank cir-
cuit. If the LT8616 circuit is plugged into a live supply, the
input voltage can ring to twice its nominal value, possibly
exceeding the LT8616’s voltage rating. This situation is
easily avoided (see Linear Technology Application Note 88).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by
the LT8616 to produce the DC output. In this role it deter-
mines the output voltage ripple, thus, low impedance at
the switching frequency is important. The second function
is to store energy in order to satisfy transient loads and
stabilize the LT8616’s control loop. Ceramic capacitors
have very low equivalent series resistance (ESR) and
provide the best ripple performance. For good starting
values, see the Typical Applications section.
Use X5R or X7R types. This choice will provide low output
ripple and good transient response. Transient performance
can be improved with a higher value output capacitor and
the addition of a feed forward capacitor placed between
VOUT and FB. Increasing the output capacitance will also
decrease the output voltage ripple. A lower value of output
capacitor can be used to save space and cost but transient
performance will suffer and may cause loop instability. See
the Typical Applications in this data sheet for suggested
capacitor values.
When choosing a capacitor, special attention should be
given to the data sheet to calculate the effective capacitance
under the relevant operating conditions of voltage bias and
temperature. A physically larger capacitor or one with a
higher voltage rating may be required.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT8616 due to their piezoelectric nature.
When in Burst Mode operation, the LT8616’s switching
frequency depends on the load current, and at very light
loads the LT8616 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT8616
operates at a lower current limit during Burst Mode op-
eration, the noise is typically very quiet to a casual ear.
If this is unacceptable, use a high performance tantalum
or electrolytic capacitor at the output. Low noise ceramic
capacitors are also available.
Enable Pin
The LT8616 is in shutdown when both EN/UV pins are low
and active when either pin is high. The rising threshold of
the EN/UV comparator is 1.03V, with 50mV of hysteresis.
The EN/UV pins can be tied to VIN if the shutdown feature
is not used, or tied to a logic level if shutdown control is
required.
Adding a resistor divider from VIN to EN/UV programs
the LT8616 to operate only when VIN is above a desired
voltage (see the Block Diagram). Typically, this threshold,
VIN(EN), is used in situations where the input supply is cur-
rent limited, or has a relatively high source resistance. A
switching regulator draws constant power from the source,
so source current increases as source voltage drops. This
looks like a negative resistance load to the source and can
cause the source to current limit or latch low under low
Table 3. Ceramic Capacitor Manufacturers
MANUFACTURER WEB
Taiyo Yuden www.t-yuden.com
AVX www.avxcorp.com
Murata www.murata.com
TDK www.tdk.com
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source voltage conditions. The VIN(EN) threshold prevents
the regulator from operating at source voltages where the
problems might occur. This threshold can be adjusted by
setting the values R5 and R6 (R7, R8 for channel 2) such
that they satisfy the following equation:
R5 =R6
IN1(EN)
1.03V – 1
⎛
⎝
⎜⎞
⎠
⎟
(10)
where the corresponding channel will remain off until VIN is
above VIN(EN). Due to the comparator’s hysteresis, switch-
ing will not stop until the input falls slightly below VIN(EN).
When operating in Burst Mode operation for light load
currents, the current through the VIN(EN) resistor network
can easily be greater than the supply current consumed
by the LT8616. Therefore, the VIN(EN) resistors should be
large to minimize their effect on efficiency at low loads.
INTVCC Regulator
An internal low dropout (LDO) regulator produces the 3.4V
supply from VIN1 that powers the drivers and the internal
bias circuitry. For this reason, VIN1 must be present and
valid to use either channel. The INTVCC pin supplies cur-
rent for the LT8616’s circuitry and must be bypassed to
ground with a 1μF ceramic capacitor. Good bypassing is
necessary to supply the high transient currents required
by the power MOSFET gate drivers. To improve efficiency,
the internal LDO will draw current from the BIAS pin when
the BIAS pin is at 3.1V or higher. Typically, the BIAS pin
is tied to the lowest output or external supply above 3.1V.
If BIAS is connected to a supply other than VOUT, bypass
it with a local ceramic capacitor. If the BIAS pin is below
3.0V, the internal LDO will consume current from VIN1.
Applications with high input voltage and high switching
frequency where the internal LDO pulls current from VIN1
will increase die temperature because of the higher power
dissipation across the LDO. Do not connect an external
load to the INTVCC pin.
Output Voltage Tracking and Soft-Start
The LT8616 allows the user to program its output voltage
ramp rate with the TR/SS pin. An internal 2μA current pulls
up the TR/SS pin to INTVCC. Putting an external capacitor
on TR/SS enables soft starting the output to prevent cur-
rent surge on the input supply. During the soft-start ramp
the output voltage will proportionally track the TR/SS pin
voltage. For output tracking applications, TR/SS can be
externally driven by another voltage source. From 0V to
0.790V, the TR/SS voltage will override the internal 0.790V
reference input to the error amplifier, thus regulating the
FB pin voltage to that of TR/SS pin (figure 4). When TR/SS
is above 0.790V, tracking is disabled and the feedback
voltage will regulate to the internal reference voltage. The
TR/SS pin may be left floating if the function is not needed.
Note the LT8616 will not discharge the output to regulate
to a lower TR/SS voltage (figure 5).
An active pull-down circuit is connected to the TR/SS pin
which will discharge the external soft-start capacitor in
the case of fault conditions and restart the ramp when the
faults are cleared. Fault conditions that clear the soft-start
capacitor are the corresponding EN/UV pin below 0.92V,
VIN1 voltage falling too low, or thermal shutdown.
Figure 4. FB Tracking TR/SS Voltage Until 0.790V
TR/SS VOLTAGE (mV)
0
0
FB VOLTAGE (mV)
600
700
800
500
400
300
200
100
900
600500400300200100 700 800 900 1000
8616 F04
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Output Power Good
When the LT8616’s output voltage is within the ±10%
window of the regulation point, which is a FB voltage in
the range of 0.72V to 0.88V (typical), the output voltage
is considered good and the open-drain PG pin goes high
impedance and is typically pulled high with an external
resistor. Otherwise, the internal pull-down device will pull
the PG pin low. To prevent glitching, both the upper and
lower thresholds include 1% of hysteresis. See figure 6.
The PG pin is also actively pulled low during several fault
conditions: corresponding EN/UV pin below 0.92V, INTVCC
voltage falling too low, VIN1 UVLO, or thermal shutdown.
applicaTions inForMaTion
Figure 5. TR/SS Does Not Discharge VOUT
Figure 6. Power-Good Thresholds
Synchronization
To select low ripple Burst Mode operation, tie the SYNC/
MODE pin below 0.4V (this can be ground or a logic low
output). To select pulse skip mode, tie the SYNC/MODE
pin above 2.4V (SYNC/MODE can be tied to INTVCC). To
synchronize the LT8616 oscillator to an external frequency
connect a square wave (with 20% to 80% duty cycle) to
the SYNC/MODE pin. The square wave amplitude should
have valleys that are below 0.4V and peaks above 2.4V
(up to 6V).
Channel 1 will synchronize its positive switch edge transi-
tions to the positive edge of the SYNC signal, and channel
2 will synchronize to the negative edge of the SYNC signal.
The LT8616 will not enter Burst Mode operation at low
output loads while synchronized to an external clock, but
instead will pulse skip to maintain regulation. The LT8616
may be synchronized over a 200kHz to 3MHz range. The
RT resistor should be chosen to set the LT8616 switching
frequency to 20% below the lowest synchronization input.
For example, if the synchronization signal will be 500kHz
and higher, the RT should be selected for 400kHz.
The slope compensation is set by the RT value, while the
minimum slope compensation required to avoid subhar-
monic oscillations is established by the inductor size,
input voltage, and output voltage. Since the synchroniza-
tion frequency will not change the slopes of the inductor
current waveform, if the inductor is large enough to avoid
subharmonic oscillations at the frequency set by RT, then
the slope compensation will be sufficient for all synchro-
nization frequencies.
The duty cycle of the SYNC signal can be used to set
the relative phasing of the two channels for minimizing
input ripple.
The LT8616 does not operate in forced continuous mode
regardless of the SYNC signal. Never leave the SYNC/
MODE pin floating.
2ms/DIV
TR/SS
500mV/DIV
VOUT
2V/DIV
8616 F05
TEMPERATURE (°C)
–50
–15
PG THRESHOLD RELATIVE TO FB (%)
10
5
0
–5
–10
15
75 100 125 150–25 0 25
8616 F06
50
PG HIGH FALLING
PG LOW RISING
PG HIGH RISING
PG LOW FALLING
LT8616
19
8616f
For more information www.linear.com/LT8616
applicaTions inForMaTion
Figure 7. Reverse VIN Protection for
Tw o Independent Input Voltages
Pulse-Skipping Mode
Pulse-skipping mode is activated by applying logic high
(above 2.4V) or an external clock to the SYNC/MODE pin.
While in pulse-skipping mode, the oscillator operates
continuously and SW transitions are aligned to the clock.
During light loads, switch pulses are skipped to regulate
the output and the quiescent current per channel will be
several hundred µA. Full switching frequency is reached
at lower output load than in Burst Mode operation.
Shorted and Reversed Input Protection
The LT8616 will tolerate a shorted output. The bottom
switch current is monitored such that if inductor current
is beyond safe levels, turn on of the top switch will be
delayed until the inductor current falls to safe levels. A
fault condition of one channel will not affect the operation
of the other.
There is another situation to consider in systems where the
output will be held high when the input to the LT8616 is
absent. This may occur in battery charging applications or
in battery-backup systems where a battery or some other
supply is OR-ed with channel 1's output. If the VIN1 pin is
allowed to float and either EN/UV pin is held high (either
by a logic signal or because it is tied to VIN1), then the
LT8616’s internal circuitry will pull its quiescent current
through its SW1 pin. This is acceptable if the system can
tolerate current draw in this state. If both EN/UV pins
are grounded the SW1 pin current will drop to near 1µA.
However, if the VIN1 pin is grounded while channel 1
output is held high, regardless of EN/UV1, parasitic body
diodes inside the LT8616 can pull current from the output
through the SW1 pin and the VIN1 pin, damaging the IC
VIN2 is not connected to the shared internal supply and
will not draw any current if left floating. If both VIN1 and
VIN2 are floating, regardless of EN/UV pins states, no-load
will be present at the output of channel 2. However, if the
VIN2 pin is grounded while channel 2 output is held high,
parasitic body diodes inside the LT8616 can pull current
from the output through the SW2 pin and the VIN2 pin,
damaging the IC
Figure 7 shows a connection of the VIN and EN/UV pins
that will allow the LT8616 to run only when the input
voltage is present and that protects against a shorted or
reversed input.
D1
LT8616
GND
D2
8616 F07
VIN2
VIN2
VIN1 VIN1
EN/UV1 EN/UV2
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 8 shows the
recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT8616’s VIN pins, GND pins, and the input
capacitors (CIN1 and CIN2). The loop formed by the input
capacitor should be as small as possible. When using a
physically large input capacitor the resulting loop may
become too large in which case using a small case/value
capacitor placed close to the VIN and GND pins plus a larger
capacitor further away is preferred. These components,
along with the inductor and output capacitor, should be
placed on the same side of the circuit board, and their
connections should be made on that layer. Place a local,
unbroken ground plane under the application circuit on
the layer closest to the surface layer. The SW and BOOST
nodes should be as small as possible. Finally, keep the FB
and RT nodes small so that the ground traces will shield
them from the SW and BOOST nodes. The exposed pad acts
as a heat sink and is connected electrically to ground. The
exposed pad of the TSSOP package is the only electrical
connection to ground and must be soldered to ground. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT8616 to additional ground planes within the circuit
board and on the bottom side.
LT8616
20
8616f
For more information www.linear.com/LT8616
applicaTions inForMaTion
Figure 8. Recommended Layout
High Temperature Considerations
For higher ambient temperatures, care should be taken in
the layout of the PCB to ensure good heat sinking of the
LT8616. The exposed pad on the bottom of the package
must be soldered to a ground plane. This ground should
be tied to large copper layers below with thermal vias;
these layers will spread heat dissipated by the LT8616.
Placing additional vias can reduce thermal resistance
further. The maximum load current should be derated
as the ambient temperature approaches the maximum
junction rating. Power dissipation within the LT8616 can
be estimated by calculating the total power loss from an
efficiency measurement and subtracting the inductor loss.
The die temperature is calculated by multiplying the LT8616
power dissipation by the thermal resistance from junction
to ambient. The LT8616 will stop switching and indicate
a fault condition if safe junction temperature is exceeded.
Open Pins and Shorting Neighboring Pins
The LT8616 in TSSOP package is designed to tolerate
faults to each pin. Output voltages will stay at or below
regulation if adjacent pins are shorted or a pin is left float-
ing. See Table 4 for pin fault behavior when the LT8616
in the TSSOP package is connected in the application
shown on Figure 9.
BOOST1VIN1
EN/UV1
INTVCC
RT
PG1
PG2
SYNC/MODE
TR/SS1
TR/SS2
SW1
LT8616
GND
FB1
BIAS
BOOST2
SW2
8616 F09
FB2
0.1µF
0.1µF
VOUT1
5V, 1.5A
10pF
2 × 47µF
4.7µF 1M
5V, 3.3V, 700KHZ STEP-DOWN CONVERTER
200k
VIN
12V
VOUT2
3.3V, 2.5A
10nF
10µH
4.7µH
1M
316k
1M
1M
187k
56.2k
VIN2
EN/UV2
4.7µF 5.6pF 47µF
1µF
Figure 9. See Table 4 for Open and Short Pin Behavior of this Application in the TSSOP Package
8616 F08
R2
NOTE: CVCC IS BELOW THE PACKAGE ON THE BACK SIDE
R1
L1
L2
R4
R3
CFF1
CBST1
CBST2
CFF2
CIN2
CIN1
RT
RPG1
COUT1
COUT2
RPG2
1
2
3
4
5
6
7
8
9
10
11
12
13
14
TOP VIEW
FE PACKAGE
28-LEAD PLASTIC TSSOP
28
27
26
25
24
23
22
21
20
19
18
17
16
15
EN/UV2
PG2
SW2
SW2
SW2
BOOST2
NC
BOOST1
SW1
SW1
PG1
TR/SS1
FB1
FB1
TR/SS2
FB2
FB2
NC
VIN2
NC
BIAS
INTVCC
NC
VIN1
NC
SYNC/MODE
EN/UV1
RT
29
GND
LT8616
21
8616f
For more information www.linear.com/LT8616
applicaTions inForMaTion
Table 4. LT8616xFE Pin Fault Behavior For Circuit In Figure 9
LT8616 Pin Float Short to Next Pin
EN/UV2 1 Part May Be On or Off Part May Be On or Off
PG2 2 No Change No Change
SW2 3 No Change No Change
SW2 4 No Change No Change
SW2 5 No Change OUT2 Below Regulation
BOOST2 6 OUT2 Below Regulation No Change
NC 7 No Change No Change
BOOST1 8 OUT1 Below Regulation OUT1 Below Regulation
SW1 9 No Change No Change
SW1 10 No Change No Change
PG1 11 No Change No Change
TR/SS1 12 No Change No Change
FB1 13 No Change OUT1 Below Regulation
FB1 14 No Change
RT 15 Switching Frequency Reduces CH1, CH2 Off
EN/UV1 16 Part May Be On or Off CH1, CH2 Off
SYNC/MODE 17 No Change No Change
NC 18 No Change No Change
VIN1 19 CH1, CH2 Off No Change
NC 20 No Change No Change
INTVCC 21 OUT1, OUT2 Below Regulation No Change
BIAS 22 No Change No Change
NC 23 No Change No Change
VIN2 24 CH2 Off No Change
NC 25 No Change No Change
FB2 26 No Change No Change
FB2 27 No Change OUT2 Below Regulation
TR/SS2 28 No Change
LT8616
22
8616f
For more information www.linear.com/LT8616
Typical applicaTions
BOOST1VIN1
EN/UV1
INTVCC
RT
PG1
PG2
SYNC/MODE
TR/SS1
TR/SS2
SW1
LT8616
GND
FB1
BIAS
BOOST2
SW2
8616 TA02
FB2
0.1µF
0.1µF
VOUT1
5V, 1.5A
10pF
100µF
6.3V, 1210
4.7µF
VIN
5.8V TO 42V
VOUT2
2.5V, 2.5A
10nF
3.3µH
IHLP-2525CZ-01
IHLP-2020BZ-01
1.5µH
1M
464k
1M
1M
187k
14.7k
VIN2
EN/UV2
4.7µF 5.6pF 47µF
10V, 1210
1µF
5V, 2.5V, 2.05MHz Step-Down Converter
BOOST1VIN1
EN/UV1
INTVCC
RT
PG1
PG2
SYNC/MODE
TR/SS1
TR/SS2
SW1
LT8616
GND
FB1
BIAS
BOOST2
SW2
8616 TA03
FB2
0.1µF
0.1µF
XFL4020-102ME
XFL4020-472ME
VOUT1
3.3V, 0.8A
2 × 100µF
6.3V, 1210
4.7µF
VIN
4.2V TO 42V
VOUT1
VOUT2
0.79V, 2.5A
10nF
4.7µH
1µH
1M
1M 316k
3.3V
1.5A
0.7A
37.4k
VIN2
EN/UV2
1M
4.7µF 5.6pF 47µF
10V,
1210
1µF
10nF
3.3V, 0.79V, 1MHz 2-Stage Step-Down Converter, Sequenced Start-Up
LT8616
23
8616f
For more information www.linear.com/LT8616
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE28 (EB) TSSOP REV K 0913
0.09 – 0.20
(.0035 – .0079)
0° – 8°
0.25
REF
0.50 – 0.75
(.020 – .030)
4.30 – 4.50*
(.169 – .177)
1 3 4 5678 9 10 11 12 13 14
192022 21 151618 17
9.60 – 9.80*
(.378 – .386)
4.75
(.187)
2.74
(.108)
28 27 26 2524 23
1.20
(.047)
MAX
0.05 – 0.15
(.002 – .006)
0.65
(.0256)
BSC 0.195 – 0.30
(.0077 – .0118)
TYP
2
RECOMMENDED SOLDER PAD LAYOUT
EXPOSED
PAD HEAT SINK
ON BOTTOM OF
PACKAGE
0.45 ±0.05
0.65 BSC
4.50 ±0.10
6.60 ±0.10
1.05 ±0.10
4.75
(.187)
2.74
(.108)
MILLIMETERS
(INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN
3. DRAWING NOT TO SCALE
SEE NOTE 4
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
6.40
(.252)
BSC
FE Package
28-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663 Rev K)
Exposed Pad Variation EB
LT8616
24
8616f
For more information www.linear.com/LT8616
LINEAR TECHNOLOGY CORPORATION 2015
LT 0415 • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8616
relaTeD parTs
Typical applicaTion
PART NUMBER DESCRIPTION COMMENTS
LT8609 42V, 2A, 95% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 2.5µA, ISD = <1µA,
MSOP-10E Package
LT8610A/AB 42V, 3.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, MSOP-10E Package
LT8610AC 42V, 3.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 2.5µA, ISD = <1µA,
MSOP-10E Package
LT8610 42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, MSOP-10E Package
LT8611 42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA and Input/Output
Current Limit/Monitor
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, 3 × 5 QFN-24 Package
LT8620 65V, 2.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 65V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, 3 × 5 QFN-24 Package
LT8614 42V, 4A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, 3 × 5 QFN-18 Package
LT8612 42V, 6A, 96% Efficiency, 2.2MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 3.0µA,
ISD = <1µA, 3 × 6 QFN-28 Package
LT8640 42V, 6A, 96% Efficiency, 3MHz Synchronous MicroPower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA,
ISD = <1µA, 3 × 4 QFN-18 Package
LT8602 42V, Quad Output (2.5A+1.5A+1.5A+1.5A) 95% Efficiency,
2.2MHz Synchronous MicroPower Step-Down DC/DC
Converter with IQ = 25µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 25µA, ISD = <1µA,
6 × 6 QFN-40 Package
BOOST1VIN1
EN/UV1
INTVCC
RT
PG1
PG2
SYNC/MODE
TR/SS1
TR/SS2
SW1
LT8616
GND
FB1
BIAS
BOOST2
SW2
8616 TA04
FB2
0.1µF
0.1µF
VOUT1
5V, 1.5A
4.7pF
47µF
10V, 1210
2.2µF
VIN
5.8V TO 42V
VOUT2
3.3V, 2.5A
10nF
3.3µH
XAL4030-332ME
XFL4020-222ME
2.2µH
1M
316k
1M
1M
187k
14.7k
VIN2
EN/UV2
2.2µF
4.7pF 22µF
10V, 1210
1µF
10nF
0.1µF
0.1µF
0.1µF
0.1µF
5V, 3.3V, 2.05MHz Step-Down Converter