LMV791MKX - Texas Instruments High-Performance Analog

LMV791,LMV792

LMV791/LMV792 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers

with Shutdown

Literature Number: SNOSAG6E

June 23, 2008

LMV791/LMV792

17 MHz, Low Noise, CMOS Input, 1.8V Operational

Amplifiers with Shutdown

General Description

The LMV791 (Single) and the LMV792 (Dual) low noise,

CMOS input operational amplifiers offer a low input voltage

noise density of 5.8 nV/ while consuming only 1.15 mA

(LMV791) of quiescent current. The LMV791 and LMV792 are

unity gain stable op amps and have gain bandwidth of 17

MHz. The LMV791/ LMV792 have a supply voltage range of

1.8V to 5.5V and can operate from a single supply. The

LMV791/LMV792 each feature a rail-to-rail output stage ca-

pable of driving a 600Ω load and sourcing as much as 60 mA

of current.

The LMV791 family provides optimal performance in low volt-

age and low noise systems. A CMOS input stage, with typical

input bias currents in the range of a few femtoAmperes, and

an input common mode voltage range which includes ground

make the LMV791 and the LMV792 ideal for low power sensor

applications. The LMV791 family has a built-in enable feature

which can be used to optimize power dissipation in low power

applications.

The LMV791/LMV792 are manufactured using National’s ad-

vanced VIP50 process and are offered in a 6-pin TSOT23 and

a 10-pin MSOP package respectively.

Features

(Typical 5V supply, unless otherwise noted)

■Input referred voltage noise 5.8 nV/√Hz

■Input bias current 100 fA

■Unity gain bandwidth 17 MHz

■Supply current per channel enable mode

—LMV791 1.15 mA

—LMV792 1.30 mA

■Supply current per channel in shutdown mode 0.02 µA

■Rail-to-rail output swing

—@ 10 kΩ load 25 mV from rail

—@ 2 kΩ load 45 mV from rail

■Guaranteed 2.5V and 5.0V performance

■Total harmonic distortion 0.01% @1 kHz, 600Ω

■Temperature range −40°C to 125°C

Applications

■Photodiode amplifiers

■Active filters and buffers

■Low noise signal processing

■Medical Instrumentation

■Sensor interface applications

Typical Application

20116869

Photodiode Transimpedance Amplifier 20116839

Input Referred Voltage Noise vs. Frequency

LMV791/LMV792 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers with Shutdown

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

ESD Tolerance (Note 2)

Human Body Model 2000V

Machine Model 200V

Charge-Device Model 1000V

VIN Differential ±0.3V

Supply Voltage (V+ – V−)6.0V

Input/Output Pin Voltage V+ +0.3V, V− −0.3V

Storage Temperature Range −65°C to 150°C

Junction Temperature (Note 3) +150°C

Soldering Information

Infrared or Convection (20 sec) 235°C

Wave Soldering Lead Temp (10 sec) 260°C

Operating Ratings (Note 1)

Temperature Range (Note 3) −40°C to 125°C

Supply Voltage (V+ – V−)

−40°C ≤ TA ≤ 125°C 2.0V to 5.5V

0°C ≤ TA ≤ 125°C 1.8V to 5.5V

Package Thermal Resistance (θJA (Note 3))

6-Pin TSOT23 170°C/W

10-Pin MSOP 236°C/W

2.5V Electrical Characteristics

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO, VEN = V+. Boldface limits

apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)

Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift LMV791 (Note 6) −1.0 μV/°C

LMV792 (Note 6) −1.8

IBInput Bias Current VCM = 1.0V

(Notes 7, 8)

−40°C ≤ TA ≤ 85 °C 0.05 1

25 pA

−40°C ≤ TA ≤ 85 °C 0.05 1

100

IOS Input Offset Current VCM = 1.0V

(Note 8)

10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 1.4V 80

94 dB

PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

1.5

1.5 V

AVOL Open Loop Voltage Gain VOUT = 0.15V to 2.2V,

RLOAD = 2 kΩ to V+/2

LMV791 85

LMV792 82

VOUT = 0.15V to 2.2V,

RLOAD = 10 kΩ to V+/2

110

VOUT Output Voltage Swing High RLOAD = 2 kΩ to V+/2 25 75

mV from

either rail

RLOAD = 10 kΩ to V+/2 20 65

Output Voltage Swing Low RLOAD = 2 kΩ to V+/2 30 75

RLOAD = 10 kΩ to V+/2 15 65

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LMV791/LMV792

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)

Units

IOUT Output Current Sourcing to V−

VIN = 200 mV (Note 9)

Sinking to V+

VIN = –200 mV (Note 9)

ISSupply Current per Amplifier Enable Mode

VEN ≥ 2.1V

LMV791 0.95 1.30

1.65 mA

LMV792

per channel

1.1 1.50

1.85

Shutdown Mode, VEN < 0.4

per channel

0.02 1

5μA

SR Slew Rate AV = +1, Rising (10% to 90%) 8.5 V/μs

AV = +1, Falling (90% to 10%) 10.5

GBW Gain Bandwidth 14 MHz

enInput Referred Voltage Noise Density f = 1 kHz 6.2 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

ton Turn-on Time 140 ns

toff Turn-off Time 1000 ns

VEN Enable Pin Voltage Range Enable Mode 2.1 2 to 2.5 V

Shutdown Mode 0 to 0.5 0.4

IEN Enable Pin Input Current Enable Mode VEN = 2.5V (Note 7) 1.5 3 μA

Shutdown Mode VEN = 0V (Note 7) 0.003 0.1

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RLOAD = 600Ω 0.01 %

5V Electrical Characteristics

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO, VEN = V+. Boldface limits

apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)

Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift LMV791 (Note 6) −1.0 μV/°C

LMV792 (Note 6) −1.8

IBInput Bias Current VCM = 2.0V

(Notes 7, 8)

−40°C ≤ TA ≤ 85°C 0.1 1

25 pA

−40°C ≤ TA ≤ 125°C 0.1 1

100

IOS Input Offset Current VCM = 2.0V

(Note 8)

10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 3.7V 80

100 dB

PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

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LMV791/LMV792

AVOL Open Loop Voltage Gain VOUT = 0.3V to 4.7V,

RLOAD = 2 kΩ to V+/2

LMV791 85

LMV792 82

VOUT = 0.3V to 4.7V,

RLOAD = 10 kΩ to V+/2

110

VOUT Output Voltage Swing High RLOAD = 2 kΩ to V+/2 35 75

mV from

either rail

RLOAD = 10 kΩ to V+/2 25 65

Output Voltage Swing Low RLOAD = 2 kΩ to V+/2 LMV791 42 75

LMV792 45 80

RLOAD = 10 kΩ to V+/2 20 65

IOUT Output Current Sourcing to V−

VIN = 200 mV (Note 9)

Sinking to V+

VIN = –200 mV (Note 9)

ISSupply Current per Amplifier Enable Mode

VEN ≥ 4.6V

LMV791 1.15 1.40

1.75 mA

LMV792

per channel

1.30 1.70

2.05

Shutdown Mode (VEN ≤ 0.4V) 0.14 1

5μA

SR Slew Rate AV = +1, Rising (10% to 90%) 6.0 9.5 V/μs

AV = +1, Falling (90% to 10%) 7.5 11.5

GBW Gain Bandwidth 17 MHz

enInput Referred Voltage Noise Density f = 1 kHz 5.8 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

ton Turn-on Time 110 ns

toff Turn-off Time 800 ns

VEN Enable Pin Voltage Range Enable Mode 4.6 4.5 to 5 V

Shutdown Mode 0 to 0.5 0.4

IEN Enable Pin Input Current Enable Mode VEN = 5.0V

(Note 7)

5.6 10

μA

Shutdown Mode VEN = 0V

(Note 7)

0.005 0.2

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RLOAD = 600Ω 0.01 %

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics

Tables.

Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF

Note 3: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is

PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.

Note 4: Typical values represent the parametric norm at the time of characterization.

Note 5: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the statistical quality

control (SQC) method.

Note 6: Offset voltage average drift is determined by dividing the change in VOS by temperature change.

Note 7: Positive current corresponds to current flowing into the device.

Note 8: This parameter is guaranteed by design and/or characterization and is not tested in production.

Note 9: The short circuit test is a momentary test, the short circuit duration is 1.5 ms.

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LMV791/LMV792

Connection Diagrams

6-Pin TSOT23

20116801

Top View

10-Pin MSOP

20116802

Top View

Ordering Information

Package Part Number Package Marking Transport Media NSC Drawing

6-Pin TSOT23 LMV791MK AS1A 1k Units Tape and Reel MK06A

LMV791MKX 3k Units Tape and Reel

10-Pin MSOP LMV792MM AX2A 1k Units Tape and Reel MUB10A

LMV792MMX 3.5k Units Tape and Reel

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LMV791/LMV792

Typical Performance Characteristics Unless otherwise specified, TA = 25°C, V– = 0, V+ = Supply Voltage

= 5V, VCM = V+/2, VEN = V+.

Supply Current vs. Supply Voltage (LMV791)

20116805

Supply Current vs. Supply Voltage (LMV792)

20116881

Supply Current vs. Supply Voltage in Shutdown Mode

20116806

VOS vs. VCM

20116809

VOS vs. VCM

20116851

VOS vs. VCM

20116811

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LMV791/LMV792

VOS vs. Supply Voltage

20116812

Slew Rate vs. Supply Voltage

20116829

Supply Current vs. Enable Pin Voltage (LMV791)

20116808

Supply Current vs. Enable Pin Voltage(LMV791)

20116807

Supply Current vs. Enable Pin Voltage (LMV792)

20116882

Supply Current vs. Enable Pin Voltage (LMV792)

20116883

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LMV791/LMV792

Input Bias Current vs. VCM

20116862

Input Bias Current vs. VCM

20116887

Sourcing Current vs. Supply Voltage

20116820

Sinking Current vs. Supply Voltage

20116819

Sourcing Current vs. Output Voltage

20116850

Sinking Current vs. Output Voltage

20116854

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LMV791/LMV792

Positive Output Swing vs. Supply Voltage

20116817

Negative Output Swing vs. Supply Voltage

20116815

Positive Output Swing vs. Supply Voltage

20116816

Negative Output Swing vs. Supply Voltage

20116814

Positive Output Swing vs. Supply Voltage

20116818

Negative Output Swing vs. Supply Voltage

20116813

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LMV791/LMV792

Input Referred Voltage Noise vs. Frequency

20116839

Time Domain Voltage Noise

20116831

Overshoot and Undershoot vs. CLOAD

20116830

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20116826

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20116804

THD+N vs. Frequency

20116874

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LMV791/LMV792

THD+N vs. Frequency

20116875

Open Loop Gain and Phase with Capacitive Load

20116841

Open Loop Gain and Phase with Resistive Load

20116873

Closed Loop Output Impedance vs. Frequency

20116832

Crosstalk Rejection

20116880

Small Signal Transient Response, AV = +1

20116838

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LMV791/LMV792

Large Signal Transient Response, AV = +1

20116837

Small Signal Transient Response, AV = +1

20116833

Large Signal Transient Response, AV = +1

20116834

Phase Margin vs. Capacitive Load (Stability)

20116845

Phase Margin vs. Capacitive Load (Stability)

20116846

Positive PSRR vs. Frequency

20116827

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LMV791/LMV792

Negative PSRR vs. Frequency

20116828

CMRR vs. Frequency

20116856

Input Common Mode Capacitance vs. VCM

20116876

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LMV791/LMV792

Application Information

ADVANTAGES OF THE LMV791/LMV792

Wide Bandwidth at Low Supply Current

The LMV791 and LMV792 are high performance op amps that

provide a unity gain bandwidth of 17 MHz while drawing a low

supply current of 1.15 mA. This makes them ideal for provid-

ing wideband amplification in portable applications. The en-

able and shutdown feature can also be used to design more

power efficient systems that offer wide bandwidth and high

performance while consuming less average power.

Low Input Referred Noise and Low Input Bias Current

The LMV791/LMV792 have a very low input referred voltage

noise density (5.8 nV/ at 1 kHz). A CMOS input stage en-

sures a small input bias current (100 fA) and low input referred

current noise (0.01 pA/ ). This is very helpful in maintain-

ing signal fidelity, and makes the LMV791 and LMV792 ideal

for audio and sensor based applications.

Low Supply Voltage

The LMV791 and the LMV792 have performance guaranteed

at 2.5V and 5V supply. The LMV791 family is guaranteed to

be operational at all supply voltages between 2.0V and 5.5V,

for ambient temperatures ranging from −40°C to 125°C, thus

utilizing the entire battery lifetime. The LMV791 and LMV792

are also guaranteed to be operational at 1.8V supply voltage,

for temperatures between 0°C and 125°C. This makes the

LMV791 family ideal for usage in low-voltage commercial ap-

plications.

RRO and Ground Sensing

Rail-to-rail output swing provides maximum possible dynamic

range at the output. This is particularly important when oper-

ating at low supply voltages. An innovative positive feedback

scheme is used to boost the current drive capability of the

output stage. This allows the LMV791 and the LMV792 to

source more than 40 mA of current at 1.8V supply. This also

limits the performance of the LMV791 family as comparators,

and hence the usage of the LMV791 and the LMV792 in an

open-loop configuration is not recommended. The input com-

mon-mode range includes the negative supply rail which

allows direct sensing at ground in single supply operation.

Enable and Shutdown Features

The LMV791 family is ideal for battery powered systems. With

a low supply current of 1.15 mA and a shutdown current of

140 nA typically, the LMV791 and LMV792 allow the designer

to maximize battery life. The enable pin of the LMV791 and

the LMV792 allows the op amp to be turned off and reduce

its supply current to less than 1 μA. To power on the op amp

the enable pin should be higher than V+ - 0.5V, where V+ is

the positive supply. To disable the op amp, the enable pin

voltage should be less than V− + 0.5V, where V− is the neg-

ative supply.

Small Size

The small footprint of the LMV791 and the LMV792 package

saves space on printed circuit boards, and enables the design

of smaller electronic products, such as cellular phones,

pagers, or other portable systems. Long traces between the

signal source and the opamp make the signal path suscepti-

ble to noise. By using a physically smaller LMV791 and

LMV792 package, the opamp can be placed closer to the sig-

nal source, reducing noise pickup and increasing signal in-

tegrity.

CAPACITIVE LOAD TOLERANCE

The LMV791 and LMV792 can directly drive 120 pF in unity-

gain without oscillation. The unity-gain follower is the most

sensitive configuration to capacitive loading. Direct capacitive

loading reduces the phase margin of amplifiers. The combi-

nation of the amplifier’s output impedance and the capacitive

load induces phase lag. This results in either an under-

damped pulse response or oscillation. To drive a heavier

capacitive load, the circuit in Figure 1 can be used.

In Figure 1, the isolation resistor RISO and the load capacitor

CL form a pole to increase stability by adding more phase

margin to the overall system. The desired performance de-

pends on the value of RISO. The bigger the RISO resistor value,

the more stable VOUT will be. Increased RISO would, however,

result in a reduced output swing and short circuit current.

20116861

FIGURE 1. Isolation of CL to Improve Stability

INPUT CAPACITANCE AND FEEDBACK CIRCUIT

ELEMENTS

The LMV791 family has a very low input bias current (100 fA)

and a low 1/f noise corner frequency (400 Hz), which makes

it ideal for sensor applications. However, to obtain this per-

formance a large CMOS input stage is used, which adds to

the input capacitance of the op-amp, CIN. Though this does

not affect the DC and low frequency performance, at higher

frequencies the input capacitance interacts with the input and

the feedback impedances to create a pole, which results in

lower phase margin and gain peaking. This can be controlled

by being selective in the use of feedback resistors, as well as

by using a feedback capacitance, CF. For example, in the in-

verting amplifier shown in Figure 2, if CIN and CF are ignored

and the open loop gain of the op amp is considered infinite

then the gain of the circuit is −R2/R1. An op amp, however,

usually has a dominant pole, which causes its gain to drop

with frequency. Hence, this gain is only valid for DC and low

frequency. To understand the effect of the input capacitance

coupled with the non-ideal gain of the op amp, the circuit

needs to be analyzed in the frequency domain using a

Laplace transform.

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LMV791/LMV792

20116864

FIGURE 2. Inverting Amplifier

For simplicity, the op amp is modelled as an ideal integrator

with a unity gain frequency of A0 . Hence, its transfer function

(or gain) in the frequency domain is A0/s. Solving the circuit

equations in the frequency domain, ignoring CF for the mo-

ment, results in an expression for the gain shown in Equation

(1)

It can be inferred from the denominator of the transfer function

that it has two poles, whose expressions can be obtained by

solving for the roots of the denominator and are shown in

Equation 2.

(2)

Equation 2 shows that as the values of R1 and R2 are in-

creased, the magnitude of the poles, and hence the band-

width of the amplifier, is reduced. This theory is verified by

using different values of R1 and R2 in the circuit shown in

Figure 1 and by comparing their frequency responses. In Fig-

ure 3 the frequency responses for three different values of

R1 and R2 are shown. When both R1 and R2 are 1 kΩ, the

response is flattest and widest; whereas, it narrows and peaks

significantly when both their values are changed to 10 kΩ or

30 kΩ. So it is advisable to use lower values of R1 and R2 to

obtain a wider and flatter response. Lower resistances also

help in high sensitivity circuits since they add less noise.

20116859

FIGURE 3. Gain Peaking Caused by Large R1, R2

A way of reducing the gain peaking is by adding a feedback

capacitance CF in parallel with R2. This introduces another

pole in the system and prevents the formation of pairs of com-

plex conjugate poles which cause the gain to peak. Figure 4

shows the effect of CF on the frequency response of the cir-

cuit. Adding a capacitance of 2 pF removes the peak, while a

capacitance of 5 pF creates a much lower pole and reduces

the bandwidth excessively.

20116860

FIGURE 4. Gain Peaking Eliminated by CF

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LMV791/LMV792

AUDIO PREAMPLIFIER WITH BANDPASS FILTERING

With low input referred voltage noise, low supply voltage and

low supply current, and a low harmonic distortion, the

LMV791 family is ideal for audio applications. Its wide unity

gain bandwidth allows it to provide large gain for a wide range

of frequencies and it can be used to design a preamplifier to

drive a load of as low as 600Ω with less than 0.01% distortion.

Two amplifier circuits are shown in Figure 5 and Figure 6.

Figure 5 is an inverting amplifier, with a 10 kΩ feedback re-

sistor, R2, and a 1kΩ input resistor, R1, and hence provides a

gain of −10. Figure 6 is a non-inverting amplifier, using the

same values of R1and R2, and provides a gain of 11. In either

of these circuits, the coupling capacitor CC1 decides the lower

frequency at which the circuit starts providing gain, while the

feedback capacitor CF decides the frequency at which the

gain starts dropping off. Figure 7 shows the frequency re-

sponse of the inverting amplifier with different values of CF.

20116865

FIGURE 5. Inverting Audio Preamplifier

20116866

FIGURE 6. Non-inverting Audio Preamplifier

20116858

FIGURE 7. Frequency Response of the Inverting Audio

Preamplifier

TRANSIMPEDANCE AMPLIFIER

CMOS input op amps are often used in transimpedance ap-

plications as they have an extremely high input impedance.

A transimpedance amplifier converts a small input current into

a voltage. This current is usually generated by a photodiode.

The transimpedance gain, measured as the ratio of the output

voltage to the input current, is expected to be large and wide-

band. Since the circuit deals with currents in the range of a

few nA, low noise performance is essential. The LMV791/

LMV792 are CMOS input op amps providing wide bandwidth

and low noise performance, and are hence ideal for tran-

simpedance applications.

Usually, a transimpedance amplifier is designed on the basis

of the current source driving the input. A photodiode is a very

common capacitive current source, which requires tran-

simpedance gain for transforming its miniscule current into

easily detectable voltages. The photodiode and amplifier’s

gain are selected with respect to the speed and accuracy re-

quired of the circuit. A faster circuit would require a photodi-

ode with lesser capacitance and a faster amplifier. A more

sensitive circuit would require a sensitive photodiode and a

high gain. A typical transimpedance amplifier is shown in Fig-

ure 8. The output voltage of the amplifier is given by the

equation VOUT = −IINRF. Since the output swing of the amplifier

is limited, RF should be selected such that all possible values

of IIN can be detected.

The LMV791/LMV792 have a large gain-bandwidth product

(17 MHz), which enables high gains at wide bandwidths. A

rail-to-rail output swing at 5.5V supply allows detection and

amplification of a wide range of input currents. A CMOS input

stage with negligible input current noise and low input voltage

noise allows the LMV791/LMV792 to provide high fidelity am-

plification for wide bandwidths. These properties make the

LMV791/LMV792 ideal for systems requiring wide-band tran-

simpedance amplification.

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LMV791/LMV792

20116869

FIGURE 8. Photodiode Transimpedance Amplifier

As mentioned earlier, the following parameters are used to

design a transimpedance amplifier: the amplifier gain-band-

width product, A0; the amplifier input capacitance, CCM; the

photodiode capacitance, CD; the transimpedance gain re-

quired, RF; and the amplifier output swing. Once a feasible

RF is selected using the amplifier output swing, these num-

bers can be used to design an amplifier with the desired

transimpedance gain and a maximally flat frequency re-

sponse.

An essential component for obtaining a maximally flat re-

sponse is the feedback capacitor, CF. The capacitance seen

at the input of the amplifier, CIN, combined with the feedback

capacitor, RF, generate a phase lag which causes gain-peak-

ing and can destabilize the circuit. CIN is usually just the sum

of CD and CCM. The feedback capacitor CF creates a pole,

fP in the noise gain of the circuit, which neutralizes the zero in

the noise gain, fZ, created by the combination of RF and CIN.

If properly positioned, the noise gain pole created by CF can

ensure that the slope of the gain remains at 20 dB/decade till

the unity gain frequency of the amplifier is reached, thus en-

suring stability. As shown in Figure 9, fP is positioned such

that it coincides with the point where the noise gain intersects

the op amp’s open loop gain. In this case, fP is also the overall

3 dB frequency of the transimpedance amplifier. The value of

CF needed to make it so is given by Equation 3. A larger value

of CF causes excessive reduction of bandwidth, while a small-

er value fails to prevent gain peaking and instability.

(3)

20116884

FIGURE 9. CF Selection for Stability

Calculating CF from Equation 3 can sometimes return unrea-

sonably small values (<1 pF), especially for high speed ap-

plications. In these cases, its often more practical to use the

circuit shown in Figure 10 in order to allow more reasonable

values. In this circuit, the capacitance CF′ is (1+ RB/RA) time

the effective feedback capacitance, CF. A larger capacitor can

now be used in this circuit to obtain a smaller effective ca-

pacitance.

For example, if a CF of 0.5 pF is needed, while only a 5 pF

capacitor is available, RB and RA can be selected such that

RB/RA = 9. This would convert a CF′ of 5 pF into a CF of

0.5 pF. This relationship holds as long as RA << RF.

20116871

FIGURE 10. Obtaining Small CF from large CF′

LMV791 AS A TRANSIMPEDANCE AMPLIFIER

The LMV791 was used to design a number of amplifiers with

varying transimpedance gains and source capacitances. The

gains, bandwidths and feedback capacitances of the circuits

created are summarized in Table 1. The frequency responses

are presented in Figure 11 and Figure 12. The feedback ca-

pacitances are slightly different from the formula in Equation

3, since the parasitic capacitance of the board and the feed-

back resistor RF had to be accounted for.

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LMV791/LMV792

TABLE 1.

Transimpedance, ATI CIN CF3 dB Frequency

470000 50 pF 1.5 pF 350 kHz

470000 100 pF 2.0 pF 250 kHz

470000 200 pF 3.0 pF 150 kHz

47000 50 pF 4.5 pF 1.5 MHz

47000 100 pF 6.0 pF 1 MHz

47000 200 pF 9.0 pF 700 kHz

20116877

FIGURE 11. Frequency Response for ATI = 470000

20116878

FIGURE 12. Frequency Response for ATI = 47000

HIGH GAIN WIDEBAND TRANSIMPEDANCE AMPLIFIER

USING THE LMV792

The LMV792, dual, low noise, wide bandwidth, CMOS input

op amp IC can be used for compact, robust and integrated

solutions for sensing and amplifying wide-band signals ob-

tained from sensitive photodiodes. One of the two op amps

available can be used to obtain transimpedance gain while

the other can be used for amplifying the output voltage to fur-

ther enhance the transimpedance gain. The wide bandwidth

of the op amps (17 MHz) ensures that they are capable of

providing high gain for a wide range of frequencies. The low

input referred noise (5.8 nV/ ) allows the amplifier to de-

liver an output with a high SNR (signal to noise ratio). The

small MSOP-10 footprint saves space on printed circuit

boards and allows ease of design in portable products.

The circuit shown in Figure 13, has the first op amp acting as

a transimpedance amplifier with a gain of 47000, while the

second stage provides a voltage gain of 10. This provides a

total transimpedance gain of 470000 with a −3 dB bandwidth

of about 1.5 MHz, for a total input capacitance of 50 pF. The

frequency response for the circuit is shown in Figure 14

20116886

FIGURE 13. 1.5 MHz Transimpedance Amplifier, with ATI

= 470000

20116879

FIGURE 14. 1.5 MHz Transimpedance Amplifier

Frequency Response

www.national.com 18

LMV791/LMV792

SENSOR INTERFACES

The low input bias current and low input referred noise of the

LMV791 and LMV792 make them ideal for sensor interfaces.

These circuits are required to sense voltages of the order of

a few μV, and currents amounting to less than a nA, and

hence the op amp needs to have low voltage noise and low

input bias current. Typical applications include infra-red (IR)

thermometry, thermocouple amplifiers and pH electrode

buffers. Figure 15 is an example of a typical circuit used for

measuring IR radiation intensity, often used for estimating the

temperature of an object from a distance. The IR sensor gen-

erates a voltage proportional to I, which is the intensity of the

IR radiation falling on it. As shown in Figure 15, K is the con-

stant of proportionality relating the voltage across the IR

sensor (VIN) to the radiation intensity, I. The resistances RA

and RB are selected to provide a high gain to amplify this volt-

age, while CF is added to filter out the high frequency noise.

20116872

FIGURE 15. IR Radiation Sensor

19 www.national.com

LMV791/LMV792

Physical Dimensions inches (millimeters) unless otherwise noted

6-Pin TSOT23

NS Package Number MK06A

10-Pin MSOP

NS package Number MUB10A

www.national.com 20

LMV791/LMV792

Notes

21 www.national.com

LMV791/LMV792

Notes

LMV791/LMV792 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers with Shutdown

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