LM48555TLX/NOPB - National Semiconductor

March 2007

LM48555

Ceramic Speaker Driver

General Description

The LM48555 is an audio power amplifier designed to drive

ceramic speakers in applications such as cell phones, smart

phones, PDAs and other portable devices. The LM48555 pro-

duces 15.7VP-P with less than 1% THD+N while operating

from a 3.2V power supply. The LM48555 features a low power

shutdown mode, and differential inputs for improved noise

rejection.

The LM48555 includes advanced click and pop suppression

that eliminates audible turn-on and turn-off transients. Addi-

tionally, the integrated boost regulator features a soft start

function that minimizes transient current during power-up

Boomer audio power amplifiers were designed specifically to

provide high quality output power with a minimal number of

external components. The LM48555 does not require boot-

strap capacitors, or snubber circuits.

The LM48555 is unity-gain stable and uses external gain-set-

ting resistors.

Key Specifications

■ IDDQ (Boost Converter + Amplifier)

at VDD = 5V 7.5mA (typ)

■ Output Voltage Swing

VDD = 3.2V, THD ≤ 1% 15.7VP-P (typ)

■ Power Supply Rejection Ratio

f = 217Hz 80dB (typ)

Features

■Fully differential amplifier

■Externally configurable gain

■Soft start function

■Low power shutdown mode

■Under voltage lockout

Applications

■Mobile phones

■PDA's

■Digital cameras

Typical Application

300099c9

* CF+ and CF- are optional. Refer to “Selecting Input and Feedback Capacitor and Resistor for Audio Amplifier” section.

FIGURE 1. Typical Audio Amplifier Application Circuit

Boomer® is a registered trademark of National Semiconductor Corporation.

LM48555 Ceramic Speaker Driver

Connection Diagrams

LM48555TL Bumps Down View

300099c8

Top View

Order Number LM48555TL

See NS Package Number TLA12Z1A

LM48555TL Marking Drawing

30009956

Top View

X = One digit date code

V = Die traceability

G = Boomer Family

I4 = LM48555TL

TLA12 Package View (Bumps Up)

30009931

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LM48555

Absolute Maximum Ratings (Notes 1, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (VDD)9.5V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJA (Note 10) 114°C/W

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX (Note 10) −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 2.7V ≤ VDD ≤ 6.5V

Electrical Characteristics (Notes 1, 2)

The following specifications apply for VDD = 3.2V and the conditions shown in “Typical Audio Amplifier Application Circuit” (see

Figure 1), unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions

LM48555 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

IDD

Quiescent Power Supply Current

in Boosted Ringer Mode

VIN = 0V, No Load

VDD = 5.0V 7.5 mA

VDD = 3.6V 10 mA

VDD = 3.2V 12 15 mA (max)

ISD Shutdown Current SD = GND (Note 9) 0.1 1 µA (max)

VLH Logic High Threshold Voltage 1.2 V (min)

VLL Logic Low Threshold Voltage 0.4 V (max)

RPULLDOWN Pulldown Resistor on SD pin 80 53 kΩ (min)

TWU Wake-up Time CSS = 0.1μF100 ms

VAMP

Boost Converter Output Voltage Voltage on

VAMP Pin 88.5

7.5

V (max)

V (min)

VOUT Output Voltage Swing THD = 1% (max); f = 1kHz 15.7 15 VP-P (min)

THD+N Total Harmonic Distortion + Noise VOUT = 14VP-P, f = 1kHz 0.05 0.5 % (max)

εOS Output Noise A-Weighted Filter, VIN = 0V 70 µV

PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 217Hz,

AV = 20dB 80 72 dB (min)

ISW Switch Current Limit 2 A

VOS Output Offset Voltage 0.5 4.5 mV (max)

CMRR Common Mode Rejection Ratio Input referred 70 65 dB (min)

UVLO Under-Voltage Lock Out 2.5 2.6 V (max)

RDS(ON) Switch ON resistance 0.3 Ω

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LM48555

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions

which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters

where no limit is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower.

Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25°C and represent the parametric norm.

Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured in a normal room environment. The SD pin should be driven as close as possible to GND for minimum shutdown current.

Note 10: The value for θJA is measured with the LM48555 mounted on a 3” x 1.5” (76.2mm x 3.81mm) four layer board. The copper thickness for all four layers

is 0.5oz (roughly 0.18mm).

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LM48555

Typical Performance Characteristics

THD+N vs Frequency

VO = 4.95VRMS, VDD = 3.2V,

ZL = 1μF + 20Ω

30009950

THD+N vs Frequency

VO = 4.95VRMS, VDD = 4.2V,

ZL = 1μF + 20Ω

30009951

THD+N vs Frequency

VO = 4.95VRMS, VDD = 5V,

ZL = 1μF + 20Ω

30009952

THD+N vs Output Voltage Swing

VDD = 3.2V, ZL = 1μF + 20Ω

30009964

THD+N vs Output Voltage Swing

VDD = 4.2V, ZL = 1μF + 20Ω

30009965

THD+N vs Output Voltage Swing

VDD = 5V, ZL = 1μF + 20Ω

30009969

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LM48555

CMRR vs Frequency

VDD = 3.2V, ZL = 1μF + 20Ω

VIN = 100mVP-P

30009957

CMRR vs Frequency

VDD = 4.2V, ZL = 1μF + 20Ω

VIN = 100mVP-P

30009958

CMRR vs Frequency

VDD = 5V, ZL = 1μF + 20Ω

VIN = 100mVP-P

30009959

PSRR vs Frequency

VDD = 3.2V, ZL = 1μF + 20Ω

VRIPPLE = 200mVP-P

30009961

PSRR vs Frequency

VDD = 4.2V, ZL = 1μF + 20Ω

VRIPPLE = 200mVP-P

30009962

PSRR vs Frequency

VDD = 5V, ZL = 1μF + 20Ω

VRIPPLE = 200mVP-P

30009963

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LM48555

Inductor Current vs Output Voltage Swing

ZL = 1μF + 20Ω, f = 1kHz,

VDD = 3V, 4.2V, 5V

30009960

Supply Current vs Supply Voltage

30009967

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LM48555

Application Information

CHARACTERISTICS OF CERAMIC SPEAKERS

Because of their ultra-thin profile piezoelectric ceramic speak-

ers are ideal for portable applications. Piezoelectric materials

have high dielectric constants and their component electrical

property is like a capacitor. Therefore, piezoelectric ceramic

speakers essentially represent capacitive loads over frequen-

cy. Because these speakers are capacitive rather than resis-

tive, they require less current than traditional moving coil

speakers. However, ceramic speakers require high driving

voltages (approximately 15VP-P). To achieve these high out-

put voltages in battery operated applications, the LM48555

integrates a boost converter with an audio amplifier. High

quality piezoelectric ceramic speakers are manufactured by

TayioYuden (www.t-yuden.com) and muRata

(www.murata.com). Tayio Yuden's MLS-A Series Ceramic

Speaker and Murata's piezoelectric speaker VSL series are

recommended.

DIFFERENTIAL AMPLIFIER EXPLANATION

The LM48555 includes a fully differential audio amplifier that

features differential input and output stages. Internally this is

accomplished by two circuits: a differential amplifier and a

common mode feedback amplifier that adjusts the output volt-

ages so that the average value remains VDD/2. When setting

the differential gain, the amplifier can be considered to have

"halves". Each half uses an input and feedback resistor (RIN_

and RF_) to set its respective closed-loop gain (see Figure 1).

With RIN+ = RIN- and RF+ = RF-, the gain is set at -RF/RIN

for each half. This results in a differential gain of

AVD = -RF/RIN (1)

It is extremely important to match the input resistors, as well

as the feedback resistors to each other for best amplifier per-

formance. A differential amplifier works in a manner where the

difference between the two input signals is amplified. In most

applications, this would require input signals that are 180° out

of phase with each other. The LM48555 can be used, how-

ever, as a single-ended input amplifier while still retaining its

fully differential benefits. In fact, completely unrelated signals

may be placed at the input pins. The LM48555 simply ampli-

fies the difference between them.

The LM48555 provides what is known as a "bridged mode"

output (bridge-tied-load, BTL). This results in output signals

at OUT+ and OUT- that are 180° out of phase with respect to

each other. Bridged mode operation is different from the tra-

ditional single-ended amplifier configuration that connects the

load between the amplifier output and ground. A bridged am-

plifier design has advantages over the single-ended configu-

ration: it provides differential drive to the load, thus doubling

maximum possible output swing for a specific supply voltage.

Up to four times the output power is possible compared with

a single-ended amplifier under the same conditions.

A bridged configuration, such as the one used in the

LM48555, also creates a second advantage over single-end-

ed amplifiers. Since the differential outputs, OUT+ and OUT-,

are biased at half-supply, no net DC voltage exists across the

load. This assumes that the input resistor pair and the feed-

back resistor pair are properly matched. BTL configuration

eliminates the output coupling capacitor required in single

supply, single-ended amplifier configurations. If an output

coupling capacitor is not used in a single-ended output con-

figuration, the half-supply bias across the load would result in

both increased internal IC power dissipation as well as per-

manent loudspeaker damage.

BOOST CONVERTER POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch

FET means the maximum output current will be determined

by power dissipation within the LM48555 FET switch. The

switch power dissipation from ON-time conduction is calcu-

lated by Equation 2.

PD(SWITCH) = DC x (IINDUCTOR(AVE))2 x RDS(ON) (W) (2)

where DC is the duty cycle.

There will be some switching losses in addition to the power

loss calculated in Eqaution 3, so some derating needs to be

applied when calculating IC power dissipation. See “Maxi-

mum Power Dissipation” section.

MAXIMUM AMPLIFIER POWER DISSIPATION

Power dissipation is a major concern when designing a suc-

cessful amplifier, whether the amplifier is bridged or single-

ended. A direct consequence of the increased power

delivered to the load by a bridge amplifier is an increase in

internal power dissipation. Since the amplifier portion of the

LM48555 has two operational amplifiers, the maximum inter-

nal power dissipation is 4 times that of a single-ended ampli-

fier. The maximum power dissipation for a given BTL

application can be derived from Equation 3.

PDMAX(AMP) = (2VDD2) / (π2RO) (W) (3)

MAXIMUM TOTAL POWER DISSIPATION

The total power dissipation for the LM48555 can be calculated

by adding Equation 2 and Equation 3 together to establish

Equation 4:

PDMAX(TOTAL) = (2VDD2) / (π2EFF2RO) (W) (4)

where

EFF = Efficiency of boost converter

The result from Equation 4 must not be greater than the power

dissipation that results from Equation 5:

PDMAX = (TJMAX - TA) / θJA (W) (5)

For the TLA12Z1A, θJA = 114°C/W. TJMAX = 150°C for the

LM48555. Depending on the ambient temperature, TA, of the

system surroundings, Equation 5 can be used to find the

maximum internal power dissipation supported by the IC

packaging. If the result of Equation 4 is greater than that of

Equation 5, then either the supply voltage must be decreased,

the load impedance increased or TA reduced. For typical ap-

plications, power dissipation is not an issue. Power dissipa-

tion is a function of output power and thus, if typical operation

is not around the maximum power dissipation point, the am-

bient temperature may be increased accordingly.

STARTUP SEQUENCE

Correct startup sequencing is important for optimal device

performance. Using the correct startup sequence will improve

click and pop performance as well as avoid transients that

could reduce battery life. The device should be in Shutdown

mode when the supply voltage is applied. Once the supply

voltage has been supplied the device can be released from

Shutdown mode.

SHUTDOWN FUNCTION

In many applications, a microcontroller or microprocessor

output is used to control the shutdown circuitry to provide a

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LM48555

quick, smooth transition into shutdown. Another solution is to

use a single-pole, single-throw switch connected between

VDD and Shutdown pins.

BOOTSTRAP PIN

The bootstrap pin provides a voltage supply for the internal

switch driver. Connecting the bootstrap pin to VAMP (See

Figure 1) allows for a higher voltage to drive the gate of the

switch thereby reducing the RDS(ON). This configuration is

necessary in applications with heavier loads. The bootstrap

pin can be connected to VDD when driving lighter loads to im-

prove device performance (IDD, THD+N, Noise, etc.).

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using

integrated power amplifiers, and switching DC-DC convert-

ers, is critical for optimizing device and system performance.

Consideration to component values must be used to maxi-

mize overall system quality. The best capacitors for use with

the switching converter portion of the LM48555 are multi-layer

ceramic capacitors. They have the lowest ESR (equivalent

series resistance) and highest resonance frequency, which

makes them optimum for high frequency switching convert-

ers. When selecting a ceramic capacitor, only X5R and X7R

dielectric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor. High-quality ceramic capacitors

can be obtained from Taiyo-Yuden and Murata.

POWER SUPPLY BYPASSING

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor location on both V1 and VDD pins should be as close

to the device as possible.

SELECTING INPUT AND FEEDBACK CAPACITORS AND

RESISTOR FOR AUDIO AMPLIFIER

Special care must be taken to match the values of the feed-

back resistors (RF+ and RF-) to each other as well as match-

ing the input resistors (RIN+ and RIN-) to each other (see

Figure 1). Because of the balanced nature of differential am-

plifiers, resistor matching differences can result in net DC

currents across the load. This DC current can increase power

consumption, internal IC power dissipation, reduce PSRR,

and possibly damage the loudspeaker. To achieve best per-

formance with minimum component count, it is highly recom-

mended that both the feedback and input resistors match to

1% tolerance or better.

The input coupling capacitors, CIN, forms a first order high

pass filter which limits low frequency response. This value

should be chosen based on needed frequency response.

High value input capacitors are both expensive and space

hungry in portable designs. A certain value capacitor is need-

ed to couple in low frequencies without severe attenuation.

Ceramic speakers used in portable systems, whether internal

or external, have little ability to reproduce signals below

100Hz to 150Hz. Thus, using a high value input capacitor may

not increase actual system performance. In addition to sys-

tem cost and size, click and pop performance is affected by

the value of the input coupling capacitor, CIN. A high value

input coupling capacitor requires more charge to reach its

quiescent DC voltage (nominally 1/2 VDD). This charge comes

from the output via the feedback and is apt to create pops

upon device enable. Thus, by minimizing the capacitor value

based on desired low frequency response, turn-on pops can

be minimized.

The LM48555 is unity-gain stable which gives the designer

maximum system flexibility. However, to drive ceramic speak-

ers, a typical application requires a closed-loop differential

gain of 10V/V. In this case, feedback capacitors (CF+, CF-)

may be needed as shown in Figure 1 to bandwidth limit the

amplifier. If the available input signal is bandwith limited, then

capacitors CF+ and CF- can be eliminated. These feedback

capacitors create a low pass filter that eliminates possible

high frequency noise. Care should be taken when calculating

the -3dB frequency (from equation 6) because an incorrect

combination of RF and CF will cause rolloff before the desired

frequency.

f–3dB = 1 / 2πRF*CF (6)

SELECTING OUTPUT CAPACITOR (CO) FOR BOOST

CONVERTER

A single 4.7μF to 10μF ceramic capacitor will provide suffi-

cient output capacitance for most applications. If larger

amounts of capacitance are desired for improved line support

and transient response, tantalum capacitors can be used.

Aluminum electrolytics with ultra low ESR such as Sanyo Os-

con can be used. Typical electrolytic capacitors are not suit-

able for switching frequencies above 500 kHz because of

significant ringing and temperature rise due to self-heating

from ripple current. An output capacitor with excessive ESR

can also reduce phase margin and cause instability. In gen-

eral, if electrolytics are used, it is recommended that they be

paralleled with ceramic capacitors to reduce ringing, switch-

ing losses, and output voltage ripple.

SELECTING A POWER SUPPLY BYPASS CAPACITOR

A supply bypass capacitor is required to serve as an energy

reservoir for the current which must flow into the coil each time

the switch turns on. This capacitor must have extremely low

ESR, so ceramic capacitors are the best choice. A nominal

value of 4.7μF is recommended, but larger values can be

used. Since this capacitor reduces the amount of voltage rip-

ple seen at the input pin, it also reduces the amount of EMI

passed back along that line to other circuitry.

SELECTING A SOFT-START CAPACITOR (CSS)

The soft-start function charges the boost converter reference

voltage slowly. This allows the output of the boost converter

to ramp up slowly thus limiting the transient current at startup.

Selecting a soft-start capacitor (CSS) value presents a trade

off between the wake-up time and the startup transient cur-

rent. Using a larger capacitor value will increase wake-up time

and decrease startup transient current while the apposite ef-

fect happens with a smaller capacitor value. A general guide-

line is to use a capacitor value 1000 times smaller than the

output capacitance of the boost converter (CO). A 0.1uF soft-

start capacitor is recommended for a typical application.

SELECTING DIODES

The external diode used in Figure 1 should be a Schottky

diode. A 20V diode such as the MBR0520 from Fairchild

Semiconductor or ON Semiconductor is recommended. The

MBR05XX series of diodes are designed to handle a maxi-

mum average current of 0.5A. For applications exceeding

0.5A average but less than 1A, a Microsemi UPS5817 can be

used.

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LM48555

OUTPUT VOLTAGE OF BOOST CONVERTER

The output voltage is set using two internal resistors. The

output voltage of the boost converter is set to 8V (typ).

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined by equation

Duty Cycle = (VAMP+VDIODE -VDD)/(VAMP+VDIODE-VSW) (7)

This applies for continuous mode operation.

INDUCTANCE VALUE

Inductor value involves trade-offs in performance. Larger in-

ductors reduce inductor ripple current, which typically means

less output voltage ripple (for a given size of output capacitor).

Larger inductors also mean more load power can be delivered

because the energy stored during each switching cycle is:

E = L/2 x IP2(8)

Where “lp” is the peak inductor current. The LM48555 will limit

its switch current based on peak current. With IP fixed, in-

creasing L will increase the maximum amount of power avail-

able to the load. Conversely, using too little inductance may

limit the amount of load current which can be drawn from the

output. Best performance is usually obtained when the con-

verter is operated in “continuous” mode at the load current

range of interest, typically giving better load regulation and

less output ripple. Continuous operation is defined as not al-

lowing the inductor current to drop to zero during the cycle.

Boost converters shift over to discontinuous operation if the

load is reduced far enough, but a larger inductor stays con-

tinuous over a wider load current range.

INDUCTOR SUPPLIERS

The recommended inductors for the LM48555 are the Taiyo-

Yuden NR4012, NR3010, and CBC3225 series and Murata's

LQH3NPN series. When selecting an inductor, the continu-

ous current rating must be high enough to avoid saturation at

peak currents. A suitable core type must be used to minimize

core (switching) losses, and wire power losses must be con-

sidered when selecting the current rating.

CALCULATING OUTPUT CURRENT OF BOOST

CONVERTER (IAMP)

The load current of the boost converter is related to the av-

erage inductor current by the relation:

IAMP = IINDUCTOR(AVE) x (1 - DC) (A) (9)

Where "DC" is the duty cycle of the application.

The switch current can be found by:

ISW = IINDUCTOR(AVE) + 1/2 (IRIPPLE) (A) (10)

Inductor ripple current is dependent on inductance, duty cy-

cle, supply voltage and frequency:

IRIPPLE = DC x (VDD-VSW) / (f x L) (A) (11)

where f = switching frequency = 1MHz

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

IAMP(max) = (1–DC)x[ISW(max)–DC(V-VSW)]/2fL (A) (12)

The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-off switching

losses of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET "ON" voltage (referred to as VSW in

equations 9 thru 12) is dependent on load current. A good

approximation can be obtained by multiplying the on resis-

tance (RDS(ON) of the FET times the average inductor current.

The maximum peak switch current the device can deliver is

dependent on duty cycle.

EVALUATION BOARD AND PCB LAYOUT GUIDELINES

For information on the LM48555 demo board and PCB layout

guidelines refer to Application Notes (AN-1611).

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LM48555

Revision History

Rev Date Description

0.1 03/15/07 Initial Web Release

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LM48555

Physical Dimensions inches (millimeters) unless otherwise noted

Thin micro SMD

Order Number LM48555TL

NS Package Number TLA12Z1A

X1 = 1.463±0.03mm X2 = 1.970±0.03mm X3 = 0.600±0.075mm

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LM48555

Notes

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LM48555

Notes

LM48555 Ceramic Speaker Driver

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