MP2307DN - Monolithic Power Systems, Inc.

MP2307

3A, 23V, 340KHz Synchronous Rectified

Step-Down Converter

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The Future of Analog IC Technology

DESCRIPTION

The MP2307 is a monolithic synchronous buck

regulator. The device integrates 100m

MOSFETS that provide 3A of continuous load

current over a wide operating input voltage of

4.75V to 23V. Current mode control provides

fast transient response and cycle-by-cycle

current limit.

An adjustable soft-start prevents inrush current

at turn-on and in shutdown mode, the supply

current drops below 1µA.

This device, available in an 8-pin SOIC

package, provides a very compact system

solution with minimal reliance on external

components.

EVALUATION BOARD REFERENCE

Board Number Dimensions

EV2307DN-00A 2.0”X x 1.5”Y x 0.5”Z

FEATURES

• 3A Continuous Output Current

4A Peak Output Current

• Wide 4.75V to 23V Operating Input Range

• Integrated 100m Power MOSFET Switches

• Output Adjustable from 0.925V to 20V

• Up to 95% Efficiency

• Programmable Soft-Start

• Stable with Low ESR Ceramic Output Capacitors

• Fixed 340KHz Frequency

• Cycle-by-Cycle Over Current Protection

• Input Under Voltage Lockout

• Thermally Enhanced 8-Pin SOIC Package

APPLICATIONS

• Distributed Power Systems

• Networking Systems

• FPGA, DSP, ASIC Power Supplies

• Green Electronics/Appliances

• Notebook Computers

“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of

Monolithic Power Systems, Inc.

TYPICAL APPLICATION

INPUT

4.75V to 23V

OUTPUT

3.3V

3.9nF

10nF

MP2307

BSIN

GND COMP

MP2307_TAC01

100

EFFICIENCY (%)

0.1 1.0 10

LOAD CURRENT (A)

MP2307_EC01

Efficiency vs

Load Current

VIN = 5V

VIN = 23V

VIN = 12V

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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PACKAGE REFERENCE

GND

COMP

TOP VIEW

MP2307_PD01_SOIC8N

EXPOSED PAD

ON BACKSIDE

Part Number* Package Temperature

MP2307DN SOIC8N

(Exposed Pad) –40° to +85°C

* For Tape & Reel, add suffix –Z (eg. MP2307DN–Z)

For Lead Free, add suffix –LF (eg. MP2307DN–LF–Z)

ABSOLUTE MAXIMUM RATINGS (1)

Supply Voltage VIN....................... –0.3V to +26V

Switch Voltage VSW ................. –1V to VIN + 0.3V

Boost Voltage VBS ..........VSW – 0.3V to VSW + 6V

All Other Pins................................. –0.3V to +6V

Junction Temperature...............................150°C

Lead Temperature ....................................260°C

Storage Temperature .............–65°C to +150°C

Recommended Operating Conditions (2)

Input Voltage VIN............................ 4.75V to 23V

Output Voltage VOUT .................... 0.925V to 20V

Ambient Operating Temp .............. –40°C to +85°C

Thermal Resistance (3) θJA θJC

SOIC8N .................................. 50 ...... 10... °C/W

Maximum Power Dissipation Operating

(TA=25°C)

SOIC8N(4), POUT ....................... .......... ........ 2W

Notes:

1) Exceeding these ratings may damage the device.

2) The device is not guaranteed to function outside of its operating conditions.

3) Measured on approximately 1” square of 1 oz copper.

4) Derating 20mW/°C at TA > 25°C

ELECTRICAL CHARACTERISTICS

VIN = 12V, TA = +25°C, unless otherwise noted.

Parameter Symbol Condition Min Typ Max Units

Shutdown Supply Current VEN = 0V 0.3 3.0 µA

Supply Current VEN = 2.0V, VFB = 1.0V 1.3 1.5 mA

Feedback Voltage VFB 4.75V ≤ VIN ≤ 23V 0.900 0.925 0.950 V

Feedback Overvoltage Threshold 1.1 V

Error Amplifier Voltage Gain (5) A

EA 400 V/V

Error Amplifier Transconductance GEA ∆IC = ±10µA 820 µA/V

High-Side Switch On-Resistance (5) R

DS(ON)1 100 m

Low-Side Switch On-Resistance (5) R

DS(ON)2 100 m

High-Side Switch Leakage Current VEN = 0V, VSW = 0V 0 10 µA

Upper Switch Current Limit Minimum Duty Cycle 4.0 5.8 A

Lower Switch Current Limit From Drain to Source 0.9 A

COMP to Current Sense

Transconductance GCS 5.2 A/V

Oscillation Frequency Fosc1 300 340 380 KHz

Short Circuit Oscillation Frequency Fosc2 V

FB = 0V 110 KHz

Maximum Duty Cycle DMAX V

FB = 1.0V 90 %

Minimum On Time (5) T

ON 220 ns

EN Shutdown Threshold Voltage VEN Rising 1.1 1.5 2.0 V

EN Shutdown Threshold Voltage

Hysterisis 220 mV

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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ELECTRICAL CHARACTERISTICS (continued)

VIN = 12V, TA = +25°C, unless otherwise noted.

Parameter Symbol Condition Min Typ Max Units

EN Lockout Threshold Voltage 2.2 2.5 2.7 V

EN Lockout Hysterisis 210 mV

Input Under Voltage Lockout

Threshold V

IN Rising 3.80 4.05 4.40 V

Input Under Voltage Lockout

Threshold Hysteresis 210 mV

Soft-Start Current VSS = 0V 6 µA

Soft-Start Period CSS = 0.1µF 15 ms

Thermal Shutdown (5) 160 °C

Note:

5) Guaranteed by design, not tested.

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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TYPICAL PERFORMANCE CHARACTERISTICS

C1 = 2 x 10µF, C2 = 2 x 22µF, L= 10µH, CSS= 0.1µF, TA = +25°C, unless otherwise noted.

20mV/div.

OUT

20mV/div.

10V/div.

1A/div.

MP2307-TPC01

Steady State Test

Waveforms

= 12V, V

OUT

= 3.3V, I

OUT

= 0A

200mV/div.

OUT

20MV/div.

V/div.

2A/div.

MP2307-TPC02

Steady State Test

Waveforms

= 12V, V

OUT

= 3.3V, I

OUT

= 3A

5V/div.

OUT

2V/div.

10V/div.

1A/div.

2ms/div.

MP2307-TPC03

Startup through

Enable Waveforms

= 12V, V

OUT

= 3.3V, No Load

5V/div.

OUT

2V/div.

10V/div.

2A/div.

2ms/div.

MP2307-TP04

Startup Through

Enable Waveforms

= 12V, V

OUT

= 3.3V,

OUT

= 3A (Resistance Load)

5V/div.

OUT

2V/div.

10V/div.

1A/div.

2ms/div.

MP2307-TPC05

Shutdown Through

Enable Waveforms

= 12V, V

OUT

= 3.3V, No Load

OUT

2V/div.

10V/div.

5V/div.

2A/div.

MP2307-TPC06

Shutdown Through

Enable Waveforms

= 12V, V

OUT

= 3.3V,

OUT

= 3A (Resistance Load)

OUT

200mV/div.

1A/div.

LOAD

1A/div.

MP2307 -TPC07

Load Transient Test

Waveforms

= 12V, V

OUT

= 3.3V,

OUT

= 1A to 2A step

OUT

2V/div.

2A/div.

MP2307-TPC08

Short Circuit Test

Waveforms

= 12V, V

OUT

= 3.3V

OUT

2V/div.

2A/div.

MP2307-TPC09

Short Circuit Recovery

Waveforms

= 12V, V

OUT

= 3.3V

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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PIN FUNCTIONS

Pin # Name Description

1 BS

High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET

switch. Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch.

2 IN

Power Input. IN supplies the power to the IC, as well as the step-down converter switches.

Drive IN with a 4.75V to 23V power source. Bypass IN to GND with a suitably large capacitor

to eliminate noise on the input to the IC. See Input Capacitor.

3 SW

Power Switching Output. SW is the switching node that supplies power to the output. Connect

the output LC filter from SW to the output load. Note that a capacitor is required from SW to

BS to power the high-side switch.

4 GND Ground (Connect the exposed pad to Pin 4).

5 FB

Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive

voltage divider connected to it from the output voltage. The feedback threshold is 0.925V. See

Setting the Output Voltage.

6 COMP

Compensation Node. COMP is used to compensate the regulation control loop. Connect a

series RC network from COMP to GND. In some cases, an additional capacitor from COMP to

GND is required. See Compensation Co m ponents.

7 EN

Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on

the regulator; low to turn it off. Attach to IN with a 100k pull up resistor for automatic startup.

8 SS

Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND

to set the soft-start period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the

soft-start feature, leave SS unconnected.

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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OPERATION

FUNCTIONAL DESCRIPTION

The MP2307 regulates input voltages from

4.75V to 23V down to an output voltage as low

as 0.925V, and supplies up to 3A of load

current.

The MP2307 uses current-mode control to

regulate the output voltage. The output voltage

is measured at FB through a resistive voltage

divider and amplified through the internal

transconductance error amplifier. The voltage at

the COMP pin is compared to the switch current

(measured internally) to control the output

voltage.

The converter uses internal N-Channel

MOSFET switches to step-down the input

voltage to the regulated output voltage. Since

the high side MOSFET requires a gate voltage

greater than the input voltage, a boost capacitor

connected between SW and BS is needed to

drive the high side gate. The boost capacitor is

charged from the internal 5V rail when SW is low.

When the FB pin voltage exceeds 20% of the

nominal regulation value of 0.925V, the over

voltage comparator is tripped and the COMP

pin and the SS pin are discharged to GND,

forcing the high-side switch off.

MP2307_BD01

LOCKOUT

COMPARATOR

INTERNAL

REGULATORS

ERROR

AMPLIFIER

1.2V

OVP

RAMP

CLK

0.925V

0.3V

CURRENT

COMPARATOR

CURRENT

SENSE

AMPLIFIER

1.1V

SHUTDOWN

COMPARATOR

COMP 6

SS 8

FB 5

GND

OSCILLATOR

110/340KHz

OVP

IN < 4.10V

EN OK

Zener

1.5V

2.5V +

Figure 1—Functional Block Diagram

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

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APPLICATIONS INFORMATION

COMPONENT SELECTION

Setting the Output Voltage

The output voltage is set using a resistive

voltage divider connected from the output

voltage to FB. The voltage divider divides the

output voltage down to the feedback voltage by

the ratio:

2R1R

VV OUTFB +

Thus the output voltage is:

2R1R

925.0VOUT

×=

R2 can be as high as 100k, but a typical value

is 10k. Using the typical value for R2, R1 is

determined by:

)925.0V(81.101R OUT −×= (k)

For example, for a 3.3V output voltage, R2 is

10k, and R1 is 26.1k. Table 1 lists

recommended resistance values of R1 and R2

for standard output voltages.

Table 1—Recommended Resistance Values

VOUT R1 R2

1.8V 9.53k 10k

2.5V 16.9k 10k

3.3V 26.1k 10k

5V 44.2k 10k

12V 121k 10k

Inductor

The inductor is required to supply constant

current to the load while being driven by the

switched input voltage. A larger value inductor

will result in less ripple current that will in turn

result in lower output ripple voltage. However,

the larger value inductor will have a larger

physical size, higher series resistance, and/or

lower saturation current. A good rule for

determining inductance is to allow the peak-to-

peak ripple current to be approximately 30% of

the maximum switch current limit. Also, make

sure that the peak inductor current is below the

maximum switch current limit.

The inductance value can be calculated by:

⎟

⎠

⎞

⎜

⎝

⎛−×

∆×

OUT

Where VOUT is the output voltage, VIN is the

input voltage, fS is the switching frequency, and

IL is the peak-to-peak inductor ripple current.

Choose an inductor that will not saturate under

the maximum inductor peak current, calculated

by:

⎟

⎠

⎞

⎜

⎝

⎛−×

××

OUT

LOADLP V

Lf2

Where ILOAD is the load current.

The choice of which style inductor to use mainly

depends on the price vs. size requirements and

any EMI constraints.

Optional Schottky Diode

During the transition between the high-side

switch and low-side switch, the body diode of

the low-side power MOSFET conducts the

inductor current. The forward voltage of this

body diode is high. An optional Schottky diode

may be paralleled between the SW pin and

GND pin to improve overall efficiency. Table 2

lists example Schottky diodes and their

Manufacturers.

Table 2—Diode Selection Guide

Part Number Voltage/Current

Rating Vendor

B130 30V, 1A Diodes, Inc.

SK13 30V, 1A Diodes, Inc.

MBRS130 30V, 1A

International

Rectifier

Input Capacitor

The input current to the step-down converter is

discontinuous, therefore a capacitor is required

to supply the AC current while maintaining the

DC input voltage. Use low ESR capacitors for

the best performance. Ceramic capacitors are

preferred, but tantalum or low-ESR electrolytic

capacitors will also suffice. Choose X5R or

X7R dielectrics when using ceramic capacitors.

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Since the input capacitor (C1) absorbs the input

switching current, it requires an adequate ripple

current rating. The RMS current in the input

capacitor can be estimated by:

⎟

⎠

⎞

⎜

⎝

⎛

×−×=

OUT

LOAD1C V

The worst-case condition occurs at VIN = 2VOUT,

where IC1 = ILOAD/2. For simplification, use an

input capacitor with a RMS current rating

greater than half of the maximum load current.

The input capacitor can be electrolytic, tantalum

or ceramic. When using electrolytic or tantalum

capacitors, a small, high quality ceramic

capacitor, i.e. 0.1µF, should be placed as close

to the IC as possible. When using ceramic

capacitors, make sure that they have enough

capacitance to provide sufficient charge to

prevent excessive voltage ripple at input. The

input voltage ripple for low ESR capacitors can

be estimated by:

⎟

⎠

⎞

⎜

⎝

⎛−××

=∆

OUT

LOAD

IN V

f1C

Where C1 is the input capacitance value.

Output Capacitor

The output capacitor (C2) is required to

maintain the DC output voltage. Ceramic,

tantalum, or low ESR electrolytic capacitors are

recommended. Low ESR capacitors are

preferred to keep the output voltage ripple low.

The output voltage ripple can be estimated by:

⎟

⎠

⎞

⎜

⎝

⎛

××

+×

⎟

⎠

⎞

⎜

⎝

⎛−×

=∆ 2Cf8

ESR

OUT

Where C2 is the output capacitance value and

RESR is the equivalent series resistance (ESR)

value of the output capacitor.

When using ceramic capacitors, the impedance

at the switching frequency is dominated by the

capacitance which is the main cause for the

output voltage ripple. For simplification, the

output voltage ripple can be estimated by:

⎟

⎠

⎞

⎜

⎝

⎛−×

×××

OUT

OUT V

2CLf8

V

When using tantalum or electrolytic capacitors,

the ESR dominates the impedance at the

switching frequency. For simplification, the

output ripple can be approximated to:

ESR

OUT

OUT R

V×

⎟

⎠

⎞

⎜

⎝

⎛−×

The characteristics of the output capacitor also

affect the stability of the regulation system. The

MP2307 can be optimized for a wide range of

capacitance and ESR values.

Compensation Components

MP2307 employs current mode control for easy

compensation and fast transient response. The

system stability and transient response are

controlled through the COMP pin. COMP is the

output of the internal transconductance error

amplifier. A series capacitor-resistor

combination sets a pole-zero combination to

govern the characteristics of the control system.

The DC gain of the voltage feedback loop is

given by:

OUT

EACSLOADVDC V

AGRA ×××=

Where VFB is the feedback voltage (0.925V),

AVEA is the error amplifier voltage gain, GCS is

the current sense transconductance and RLOAD

is the load resistor value.

The system has two poles of importance. One

is due to the compensation capacitor (C3) and

the output resistor of the error amplifier, and the

other is due to the output capacitor and the load

resistor. These poles are located at:

VEA

1P A3C2

f××π

LOAD

2P R2C2

f××π

Where GEA is the error amplifier transconductance.

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The system has one zero of importance, due to the

compensation capacitor (C3) and the compensation

resistor (R3). This zero is located at:

3R3C2

f1Z ××π

The system may have another zero of

importance, if the output capacitor has a large

capacitance and/or a high ESR value. The zero,

due to the ESR and capacitance of the output

capacitor, is located at:

ESR

ESR R2C2

f××π

In this case, a third pole set by the

compensation capacitor (C6) and the

compensation resistor (R3) is used to

compensate the effect of the ESR zero on the

loop gain. This pole is located at:

3R6C2

f3P ××π

The goal of compensation design is to shape

the converter transfer function to get a desired

loop gain. The system crossover frequency

where the feedback loop has the unity gain is

important. Lower crossover frequencies result

in slower line and load transient responses,

while higher crossover frequencies could cause

system instability. A good standard is to set the

crossover frequency below one-tenth of the

switching frequency.

To optimize the compensation components, the

following procedure can be used.

1. Choose the compensation resistor (R3) to set

the desired crossover frequency.

Determine R3 by the following equation:

OUT

CSEA

OUT

CSEA

f1.02C2

f2C2

3R ×

×××π

<×

××π

Where fC is the desired crossover frequency

which is typically below one tenth of the

switching frequency.

2. Choose the compensation capacitor (C3) to

achieve the desired phase margin. For

applications with typical inductor values, setting

the compensation zero (fZ1) below one-forth of

the crossover frequency provides sufficient

phase margin.

Determine C3 by the following equation:

f3R2

3C ××π

Where R3 is the compensation resistor.

3. Determine if the second compensation

capacitor (C6) is required. It is required if the

ESR zero of the output capacitor is located at

less than half of the switching frequency, or the

following relationship is valid:

R2C2

ESR

××π

If this is the case, then add the second

compensation capacitor (C6) to set the pole fP3

at the location of the ESR zero. Determine C6

by the equation:

R2C

6C ESR

PCB Layout Guide

PCB layout is very important to achieve stable

operation. It is highly recommended to duplicate

EVB layout for optimum performance.

If change is necessary, please follow these

guidelines and take Figure2 for reference.

1) Keep the path of switching current short

and minimize the loop area formed by Input

cap., high-side MOSFET and low-side

MOSFET.

2) Bypass ceramic capacitors are suggested

to be put close to the Vin Pin.

3) Ensure all feedback connections are short

and direct. Place the feedback resistors

and compensation components as close to

the chip as possible.

4) ROUT SW away from sensitive analog areas

such as FB.

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5) Connect IN, SW, and especially GND

respectively to a large copper area to cool

the chip to improve thermal performance

and long-term reliability.

TOP Layer

Bottom Layer

Figure 2―PCB Layout (Double Layer)

External Bootstrap Diode

An external bootstrap diode may enhance the

efficiency of the regulator, the applicable

conditions of external BS diode are:

z VOUT is 5V or 3.3V; and

z Duty cycle is high: D=

OUT

V>65%

In these cases, an external BS diode is

recommended from the output of the voltage

regulator to BS pin, as shown in Figure3

MP2307

BST

5V or 3.3V

OUT

External BST Diode

IN4148

Figure 3—Add Optional External Bootstrap

Diode to Enhance Efficiency

The recommended external BS diode is IN4148,

and the BS cap is 0.1~1µF.

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TYPICAL APPLICATION CIRCUIT

INPUT

4.75V to 23V

OUTPUT

3.3V

3.9nF

B130

(optional)

10nF

MP2307

BSIN

GND COMP

(optional)

MP2307_F03

Figure 4—MP2307 with 3.3V Output, 22uF/6.3V Ceramic Output Capacitor

MP2307 – 3A, 23V, 340KHz SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER

NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third

party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not

assume any legal responsibility for any said applications.

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PACKAGE INFORMATION

SOIC8N (EXPOSED PAD)

SEE DETAIL "A"

0.0075(0.19)

0.0098(0.25)

0.050(1.27)

BSC

0.013(0.33)

0.020(0.51)

SEATING PLANE

0.000(0.00)

0.006(0.15)

0.051(1.30)

0.067(1.70)

TOP VIEW

FRONT VIEW

SIDE VIEW

BOTTOM VIEW

NOTE:

1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN

BRACKET IS IN MILLIMETERS.

2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,

PROTRUSIONS OR GATE BURRS.

3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH

OR PROTRUSIONS.

4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)

SHALL BE 0.004" INCHES MAX.

5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.

6) DRAWING IS NOT TO SCALE.

0.089(2.26)

0.101(2.56)

0.124(3.15)

0.136(3.45)

RECOMMENDED LAND PATTERN

0.213(5.40)

0.063(1.60)

0.050(1.27)

0.024(0.61)

0.103(2.62)

0.138(3.51)

0.150(3.80)

0.157(4.00)

PIN 1 ID

0.189(4.80)

0.197(5.00)

0.228(5.80)

0.244(6.20)

0.016(0.41)

0.050(1.27)

0o-8o

DETAIL "A"

0.010(0.25)

0.020(0.50) x 45o

0.010(0.25) BSC

GAUGE PLANE