High Performance Video Op Amp
Data Sheet
AD811
FEATURES
High speed
140 MHz bandwidth (3 dB, G = +1)
120 MHz bandwidth (3 dB, G = +2)
35 MHz bandwidth (0.1 dB, G = +2)
2500 V/µs slew rate
25 ns settling time to 0.1% (for a 2 V step)
65 ns settling time to 0.01% (for a 10 V step)
Excellent video performance (RL =150 Ω)
0.01% differential gain, 0.01° differential phase
Voltage noise of 1.9 nV/√Hz
Low distortion: THD = −74 dB at 10 MHz
Excellent dc precision: 3 mV max input offset voltage
Flexible operation
Specified for ±5 V and ±15 V operation
±2.3 V output swing into a 75 Ω load (VS = ±5 V)
APPLICATIONS
Video crosspoint switchers, multimedia broadcast systems
HDTV compatible systems
Video line drivers, distribution amplifiers
ADC/DAC buffers
DC restoration circuits
Medical
Ultrasound
PET
Gamma
Counter applications
MIL-STD-883B parts available
CONNECTION DIAGRAMS
Figure 1. 8-Lead Plastic (N-8), CERDIP (Q-8), SOIC_N (R-8)
Figure 2. 16-Lead SOIC_W (RW-16)
Figure 3. 20-Terminal LCC (E-20-1)
GENERAL DESCRIPTION
A wideband current feedback operational amplifier, the AD811
is optimized for broadcast-quality video systems. The −3 dB
bandwidth of 120 MHz at a gain of +2 and the differential gain
and phase of 0.01% and 0.01° (RL = 150 Ω) make the AD811 an
excellent choice for all video systems. The AD811 is designed to
meet a stringent 0.1 dB gain flatness specification to a bandwidth of
35 MHz (G = +2) in addition to low differential gain and phase
errors. This performance is achieved whether driving one or two
back-terminated 75 Ω cables, with a low power supply current
of 16.5 mA. Furthermore, the AD811 is specified over a power
supply range of ±4.5 V to ±18 V.
The AD811 is also excellent for pulsed applications where
transient response is critical. It can achieve a maximum slew
rate of greater than 2500 V/µs with a settling time of less than
25 ns to 0.1% on a 2 V step and 65 ns to 0.01% on a 10 V step.
The AD811 is ideal as an ADC or DAC buffer in data acquisition
systems due to its low distortion up to 10 MHz and its wide unity
gain bandwidth. Because the AD811 is a current feedback ampli-
fier, this bandwidth can be maintained over a wide range of
gains. The AD811 also offers low voltage and current noise of
1.9 nV/√Hz and 20 pA/√Hz, respectively, and excellent dc
accuracy for wide dynamic range applications.
1
2
3
4
8
7
6
5
AD811
NC
–IN
+IN
–V
S
+V
S
NC
OUTPUT
NC
NC = NO CONNECT
00866-E-001
NC 1
NC 2
–IN 3
NC 4
NC
16
NC
15
+VS
14
NC
13
+IN 5OUTPUT
12
NC 6NC
11
–VS7NC
10
NC 8NC
9
NOTES
1. NC = NO CO NNE C T. DO NOT CONNE CT TO THIS PIN.
AD811
TOP VIEW
(No t t o Scal e)
00866-E-002
4
NC
5
NC
6
–IN
7
NC
8
+IN
18
NC
17
NC
16
+V
S
15
NC
14
OUTPUT
19
NC
20
NC
1
NC
2
NC
3
NC
13
NC
12
NC
11
NC
10
NC
9
–V
S
TOP VIEW
(No t t o Scal e)
00866-E-003
AD811
NOTES
1. NC = NO CO NNE C T. DO NOT CONNE CT TO THIS PIN.
Rev. G Document Feedback
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 ©2015 Analog Devices, Inc. All rights reserved.
Technical Support www.analog.com
AD811 Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications ....................................................................................... 1
Connection Diagrams ...................................................................... 1
General Description ......................................................................... 1
Revision History ............................................................................... 2
Specifications ..................................................................................... 3
Absolute Maximum Ratings ............................................................ 5
Maximum Power Dissipation ..................................................... 5
Metalization Photograph ............................................................. 5
ESD Caution .................................................................................. 5
Typical Performance Characteristics ..............................................6
Applications Information .............................................................. 12
General Design Considerations ............................................... 12
Achieving the Flattest Gain Response at High Frequency .... 12
Operation as a Video Line Driver ............................................ 14
An 80 MHz Voltage-Controlled Amplifier Circuit................ 15
A Video Keyer Circuit ............................................................... 16
Outline Dimensions ....................................................................... 18
Ordering Guide .......................................................................... 20
REVISION HISTORY
4/15—Rev. F to Rev. G
Changes to Figure 25 ........................................................................ 9
Changes to Ordering Guide .......................................................... 20
2/14—Rev. E to Rev. F
Changes to R-8 Package, RW-16 Package, and E-20-1 Package;
Deleted R-20 Package .................................................... Throughout
Changes to Applications Section .................................................... 1
Removed Figure 4; Renumbered Sequentially .............................. 1
Moved Figure 4 and Figure 5 .......................................................... 6
Changes to An 80 MHz Voltage-Controlled Amplifier
Circuit Section ................................................................................ 15
Updated Outline Dimensions, Removed Figure 54 ................... 18
Changes to Ordering Guide .......................................................... 19
7/04—Rev. D to Rev. E
Updated Format .................................................................. Universal
Change to Maximum Power Dissipation Section ........................ 7
Changes to Ordering Guide .......................................................... 20
Updated Outline Dimensions ....................................................... 20
Rev. G | Page 2 of 20
Data Sheet AD811
SPECIFICATIONS
At TA = +25°C, VS = ±15 V dc, RLOAD = 150 Ω, unless otherwise noted.
Table 1.
AD811J/AD811A1 AD811S2
Parameter Conditions VS Min Typ Max Min Typ Max Unit
DYNAMIC PERFORMANCE
Small Signal Bandwidth (No Peaking)
−3 dB
G = +1 RFB = 562 Ω ±15 V 140 140 MHz
G = +2 RFB = 649 Ω ±15 V 120 120 MHz
G = +2 RFB = 562 Ω ±15 V 80 80 MHz
G = +10 RFB = 511 Ω ±15 V 100 100 MHz
0.1 dB Flat
G = +2 RFB = 562 Ω ±15 25 25 MHz
RFB = 649 Ω ±15 35 35 MHz
Full Power Bandwidth3 VOUT = 20 V p-p ±15 40 40 MHz
Slew Rate VOUT = 4 V p-p ±15 400 400 V/µs
VOUT = 20 V p-p ±15 2500 2500 V/µs
Settling Time to 0.1% 10 V Step, AV = − 1 ±15 50 50 ns
Settling Time to 0.01% 10 V Step, AV = − 1 ±15 65 65 ns
Settling Time to 0.1% 2 V Step, AV = − 1 ±15 25 25 ns
Rise Time, Fall Time RFB = 649, AV = +2 ±15 3.5 3.5 ns
Differential Gain f = 3.58 MHz ±15 0.01 0.01 %
Differential Phase f = 3.58 MHz ±15 0.01 0.01 Degree
THD at fC = 10 MHz VOUT = 2 V p-p, AV = +2 ±15 −74 −74 dBc
Third-Order Intercept
4
At f
C
= 10 MHz
±15
36
36
dBm
±15 43 43 dBm
INPUT OFFSET VOLTAGE ±5 V, ±15 V 0.5 3 0.5 3 mV
TMIN to TMAX 5 5 mV
Offset Voltage Drift 5 5 µV/°C
INPUT BIAS CURRENT
−Input ±5 V, ±15 V 2 5 2 5 µA
T
MIN
to T
MAX
15
30
µA
+Input ±5 V, ±1 5 V 2 10 2 10 µA
TMIN to TMAX 20 25 µA
TRANSRESISTANCE TMIN to TMAX
VOUT = ±10 V
RL = ∞ ±15 V 0.75 1.5 0.75 1.5 MΩ
RL = 200 Ω ±15 V 0.5 0.75 0.5 0.75 MΩ
VOUT = ±2.5 V
RL = 150 Ω ±5 V 0.25 0.4 0.125 0.4 MΩ
COMMON-MODE REJECTION
VOS (vs. Common Mode)
T
MIN
to T
MAX
V
CM
= ±2.5 V
±5 V
56
60
50
60
dB
TMIN to TMAX VCM = ±10 V ±15 V 60 66 56 66 dB
Input Current (vs. Common Mode) TMIN to TMAX 1 3 1 3 µA/V
POWER SUPPLY REJECTION VS = ±4.5 V to ±18 V
VOS TMIN to TMAX 60 70 60 70 dB
+Input Current TMIN to TMAX 0.3 2 0.3 2 µA/V
−Input Current TMIN to TMAX 0.4 2 0.4 2 µA/V
Rev. G | Page 3 of 20
AD811 Data Sheet
AD811J/AD811A1 AD811S2
Parameter Conditions VS Min Typ Max Min Typ Max Unit
INPUT VOLTAGE NOISE f = 1 kHz 1.9 1.9 nV/√Hz
INPUT CURRENT NOISE f = 1 kHz 20 20 pA/√Hz
OUTPUT CHARACTERISTICS
Voltage Swing, Useful Operating
Range5
±5 V ±2.9 ±2.9 V
±15 V ±12 ±12 V
Output Current TJ = 25°C 100 100 mA
Short-Circuit Current 150 150 mA
Output Resistance
(Open Loop at 5 MHz)
9
9
Ω
INPUT CHARACTERISTIC
+Input Resistance 1.5 1.5 MΩ
−Input Resistance 14 14 Ω
Input Capacitance +Input 7.5 7.5 pF
Common-Mode Voltage Range ±5 V ±3 ±3 V
±15 V ±13 ±13 V
POWER SUPPLY
Operating Range ±4.5 ±18 ±4.5 ±18 V
Quiescent Current ±5 V 14.5 16.0 14.5 16.0 mA
±15 V 16.5 18.0 16.5 18.0 mA
TRANSISTOR COUNT Number of Transistors 40 40
1 The AD811JR is specified with ±5 V power supplies only, with operation up to ±12 V.
2 See the Analog Devices military data sheet for 883B tested specifications.
3 FPBW = slew rate/(2 π VPEAK).
4 Output power level, tested at a closed-loop gain of two.
5 Useful operating range is defined as the output voltage at which linearity begins to degrade.
Rev. G | Page 4 of 20
Data Sheet AD811
Rev. G | Page 5 of 20
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Rating
Supply Voltage ±18 V
AD811JR Grade Only ±12 V
Internal Power Dissipation Observe Derating Curves
8-Lead PDIP Package θJA = 90°C/ W
8-Lead CERDIP Package θJA = 110°C/W
8-Lead SOIC_N Package θJA = 155°C/W
16-Lead SOIC_W Package θJA = 85°C/W
20-Lead LCC Package θJA = 70°C/W
Output Short-Circuit Duration Observe Derating Curves
Common-Mode Input Voltage ±VS
Differential Input Voltage ±6 V
Storage Temperature Range (Q, E) −65°C to +150°C
Storage Temperature Range (N, R) −65°C to +125°C
Operating Temperature Range
AD811J 0°C to +70°C
AD811A −40°C to +85°C
AD811S −55°C to +125°C
Lead Temperature Range
(Soldering 60 sec)
300°C
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
MAXIMUM POWER DISSIPATION
The maximum power that can be safely dissipated by the AD811 is
limited by the associated rise in junction temperature. For the
plastic packages, the maximum safe junction temperature is 145°C.
For the CERDIP and LCC packages, the maximum junction tem-
perature is 175°C. If these maximums are exceeded momentarily,
proper circuit operation is restored as soon as the die tempera-
ture is reduced. Leaving the device in the overheated condition
for an extended period can result in device burnout. To ensure
proper operation, it is important to observe the derating curves
in Figure 21 and Figure 24.
While the AD811 is internally short-circuit protected, this may
not be sufficient to guarantee that the maximum junction tem-
perature is not exceeded under all conditions. An important
example is when the amplifier is driving a reverse-terminated
75 Ω cable and the cable’s far end is shorted to a power supply.
With power supplies of ±12 V (or less) at an ambient temperature
of +25°C or less, and the cable shorted to a supply rail, the
amplifier is not destroyed, even if this condition persists for
an extended period.
METALIZATION PHOTOGRAPH
Contact the factory for the latest dimensions.
Figure 4. Metalization Photograph
Dimensions Shown in Inches and (Millimeters)
ESD CAUTION
0.0618
(1.57)
0.098 (2.49)
+INPUT
–
INPUT V–
V+ VOUT
AD811
6
7
4
3
2
00866-E-007
AD811 Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
Figure 5. Differential Gain and Phase
Figure 6. Frequency Response
Figure 7. Input Common-Mode Voltage Range vs. Supply Voltage
Figure 8. Output Voltage Swing vs. Resistive Load
Figure 9. Input Bias Current vs. Junction Temperature
Figure 10. Output Voltage Swing vs. Supply Voltage
0
0.02
0.04
0.06
0.08
0.10
0.12
0.14
0.16
0.18
0.20
DIFFERENTIAL PHASE (DEGREES)
0
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
0.09
0.10
DIFFERENTIAL GAIN (%)
5 6 7 8 9 10 11 12 13 14 15
SUPPLY VOLTAGE (±V)
00866-E-005
RF = 649Ω
FC = 3.58MHz
100 IRE
MODULATED RAMP
RL = 150Ω
PHASE
GAIN
–6
–3
0
3
6
9
12
GAIN (dB)
FREQUENCY (MHz)
110 100
00866-E-006
V
S
= ±15V
V
S
= ±5V
G = +2
R
L
= 150Ω
R
G
= R
FB
COMMON-MODE VOLTAGE RANGE (±V)
0
5
10
15
20
SUPPLY VOLTAGE (±V)
5
0 10 15 20
00866-E-008
T
A
= 25°C
0
5
10
15
20
25
30
35
OUTPUT VOLTAGE (V p-p)
LOAD RESISTANCE (Ω)
10 1k100 10k
00866-E-009
V
S
= ±15V
V
S
= ±5V
–30
–25
–20
–15
–10
–5
MASTER CLOCK FREQUENCY (MHz)
0
5
10
–60 –40 –20 0 20 40 60 80 100 120 140
JUNCTION TEMPERATURE (°C)
00866-E-010
NONINVERTING INPUT
±5 TO ±15V
INVERTING INPUT
V
S
= ±15V
V
S
= ±5V
MAGNITUDE OF THE OUTPUT VOLTAGE (±V)
0
5
10
15
20
SUPPLY VOLTAGE (±V)
50 10 15 20
00866-E-011
T
A
= 25°C
NO LOAD
R
L
= 150Ω
Rev. G | Page 6 of 20
Data Sheet AD811
Figure 11. Quiescent Supply Current vs. Junction Temperature
Figure 12. Input Offset Voltage vs. Junction Temperature
Figure 13. Short-Circuit Current vs. Junction Temperature
Figure 14. Closed-Loop Output Resistance vs. Frequency
Figure 15. Rise Time and Overshoot vs. Value of Feedback Resistor, RFB
Figure 16. Transresistance vs. Junction Temperature
3
6
9
12
15
18
21
QUIESCENT SUPPLY CURRENT (mA)
–60 –40 –20 0 20 40 60 80 100 120 140
JUNCTION TEMPERATURE (°C)
00866-E-012
V
S
= ±15V
V
S
= ±5V
–10
–8
–6
–4
–2
0
2
4
6
8
10
INPUT OFFSET VOLTAGE (mV)
–60 –40 –20 0 20 40 60 80 100 120 140
JUNCTION TEMPERATURE (°C)
00866-E-013
V
S
= ±15V
V
S
= ±5V
SHORT-CIRCUIT CURRENT (mA)
50
100
150
200
250
–60 –40 –20 0 20 40 60 80 100 120 140
JUNCTION TEMPERATURE (°C)
00866-E-014
V
S
= ±15V
V
S
= ±5V
CLOSED-LOOP OUTPUT RESISTANCE (Ω)
0.01
1
0.1
10
FREQUENCY (Hz)
100k10k 1M 10M 100M
00866-E-015
V
S
= ±15V
V
S
= ±5V
GAIN = –2
R
FB
= 649Ω
–20
0
20
40
60
100
OVERSHOOT (%)
0
2
4
6
8
10
RISE TIME (ns)
0.8 1.00.4 0.6 1.2 1.4 1.6
VALUE OF FEEDBACK RESISTOR [R
FB
] (kΩ)
00866-E-016
OVERSHOOT
RISE TIME
V
S
= ±15V
V
O
= 1V p-p
R
L
= 150Ω
GAIN = +2
TRANSRESISTANCE (MΩ)
0
0.5
1.0
1.5
2.0
–60 –40 –20 0 20 40 60 80 100 120 140
JUNCTION TEMPERATURE (°C)
00866-E-017
V
S
= ±15V
R
L
= 200Ω
V
OUT
= ±10V
V
S
= ±5V
R
L
= 150Ω
V
OUT
= ±2.5V
Rev. G | Page 7 of 20
AD811 Data Sheet
Figure 17. Input Noise vs. Frequency
Figure 18. −3 dB Bandwidth and Peaking vs. Value of RFB
Figure 19. Common-Mode Rejection Ratio vs. Frequency
Figure 20. Power Supply Rejection Ration vs. Frequency
Figure 21. Maximum Power Dissipation vs. Temperature for Plastic Packages
Figure 22. Large Signal Frequency Response
NOISE CURRENT (pA/ Hz)
1
10
100
NOISE VOLTAGE (nV/ Hz)
1
10
100
FREQUENCY (Hz)
10010 1k 10k 100k
00866-E-018
NONINVERTING CURRENT V
S
= ±5V TO ±15V
INVERTING CURRENT V
S
= ±5V TO ±15V
VOLTAGE NOISE V
S
= ±15V
VOLTAGE NOISE V
S
= ±5V
0
2
4
6
8
10
PEAKING (dB)
0
40
80
120
160
200
–3dB BANDWIDTH (MHz)
0.8 1.00.4 0.6 1.2 1.4 1.6
VALUE OF FEEDBACK RESISTOR [R
FB
] (kΩ)
00866-E-019
PEAKING
BANDWIDTH V
S
= ±15V
V
O
= 1V p-p
R
L
= 150Ω
GAIN = +2
30
40
50
60
70
80
CMRR (dB)
90
100
110
FREQUENCY (Hz)
10k1k 100k 1M 10M
00866-E-020
V
S
= ±15V
V
S
= ±5V
649Ω
150Ω150Ω
649Ω
V
IN
V
OUT
5
10
20
30
40
50
PSRR (dB)
60
70
80
FREQUENCY (Hz)
10k1k 100k 1M 10M
00866-E-021
V
S
= ±15V
R
F
= 649Ω
A
V
= +2
V
S
= ±5V
CURVES ARE FOR WORST
CASE CONDITION WHERE
ONE SUPPLY IS VARIED
WHILE THE OTHER IS
HELD CONSTANT.
TOTAL POWER DISSIPATION (W)
0.5
1.0
1.5
2.0
2.5
–50 –30–40 –20 –10 020 6010 30 40 50 70 80 90
AMBIENT TEMPERATURE ( °C)
00866-E-022
T
J
MAX = –145°C
16-L E AD S OIC
8-L E AD S OIC
8-L E AD P DIP
0
5
10
15
20
25
OUTPUT VOLTAGE (V p-p)
FREQUENCY (Hz)
100k 10M1M 100M
00866-E-023
V
S
= ±15V
V
S
= ±5V
GAIN = +10
OUTPUT LEVEL
FOR 3% THD
Rev. G | Page 8 of 20
Data Sheet AD811
Figure 23. Harmonic Distortion vs. Frequency
Figure 24. Maximum Power Dissipation vs.
Temperature for Hermetic Packages
Figure 25. Noninverting Amplifier Connection
Figure 26. Small Signal Pulse Response, Gain = +1
Figure 27. Small Signal Pulse Response, Gain = +10
Figure 28. Closed-Loop Gain vs. Frequency, Gain = +1
HARMONIC DISTORTION (dBc)
–130
–110
–90
�–70
–50
FREQUENCY (Hz)
10k1k 100k 1M 10M
00866-E-024
R
L
= 100Ω
V
OUT
= 2V p-p
GAIN = +2
±5V SUPPLIES
±15V SUPPLIES
SECOND HARMONIC
THIRD HARMONIC
SECOND
HARMONIC THIRD HARMONIC
0.4
0.8
0.6
1.2
2.0
2.4
2.8
3.0
3.2
3.4
1.6
1.0
1.8
2.2
2.6
1.4
TOTAL POWER DISSIPATION (W)
–60 –40 –20 0 20 40 60 80 100 120 140
AMBIENT TEMPERATURE (°C)
00866-E-025
T
J
MAX = –175°C
20-LEAD LCC
8-LEAD CERDIP
VOUT TO
TEKTRONIX
P6201 F ET
PROBE
0.1µF
0.1µF
RL
50Ω
AD811
+VS
+
RFB
VIN
–VS
–
HP8130
PULSE
GENERATOR
RG2
3
7
4
6
00866-026
00866-E-027
10
90
100
0%
1V
1V10ns
V
IN
V
OUT
00866-E-028
10
90
100
0%
1V
100mV 10ns
V
IN
V
OUT
–12
–9
–6
–3
0
3
6
9
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-029
V
S
= ±15V
R
FB
= 750Ω
V
S
= ±5V
R
FB
= 619Ω
G = +1
R
L
= 150Ω
R
G
= ∞
Rev. G | Page 9 of 20
AD811 Data Sheet
Figure 29. Closed-Loop Gain vs. Frequency, Gain = +10
Figure 30. Large Signal Pulse Response, Gain = +10
Figure 31. Inverting Amplifier Connection
Figure 32. Small Signal Pulse Response, Gain = −1
Figure 33. Small Signal Pulse Response, Gain = −10
Figure 34. Closed-Loop Gain vs. Frequency, Gain = −1
8
11
14
17
20
23
26
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-030
V
S
= ±15V
R
FB
= 511Ω
V
S
= ±5V
R
FB
= 442Ω
G = +1
R
L
= 150Ω
00866-E-031
10
90
100
0%
10V
1V20ns
V
IN
V
OUT
00866-E-032
HP8130
PULSE
GENERATOR
V
IN
R
G
R
FB
R
L
–V
S
+V
S
AD811
+
–
7
6
4
3
2
0.1µF
0.1µF
V
OUT
TO
TEKTRONIX
P6201 FET
PROBE
00866-E-033
10
90
100
0%
1V
1V 10ns
V
IN
V
OUT
00866-E-034
10
90
100
0%
1V
100mV 10ns
V
IN
V
OUT
–12
–9
–6
–3
0
3
6
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-035
V
S
= ±15V
R
FB
= 590Ω
V
S
= ±5V
R
FB
= 562Ω
G = –1
R
L
= 150Ω
Rev. G | Page 10 of 20
Data Sheet AD811
Figure 35. Closed-Loop Gain vs. Frequency, Gain = −10
Figure 36. Large Signal Pulse Response, Gain = −10
8
11
14
17
20
23
26
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-036
V
S
= ±15V
R
FB
= 511Ω
V
S
= ±5V
R
FB
= 442Ω
G = –1
R
L
= 150Ω
00866-E-037
10
90
100
0%
10V
1V 20ns
V
IN
V
OUT
Rev. G | Page 11 of 20
AD811 Data Sheet
APPLICATIONS INFORMATION
GENERAL DESIGN CONSIDERATIONS
The AD811 is a current feedback amplifier optimized for use in
high performance video and data acquisition applications.
Because it uses a current feedback architecture, its closed-loop
−3 dB bandwidth is dependent on the magnitude of the
feedback resistor. The desired closed-loop gain and bandwidth
are obtained by varying the feedback resistor (RFB) to tune the
bandwidth and by varying the gain resistor (RG) to obtain the
correct gain. Table 3 contains recommended resistor values for
a variety of useful closed-loop gains and supply voltages.
Table 3. −3 dB Bandwidth vs. Closed-Loop Gain and
Resistance Values
VS = ±15 V
Closed-Loop Gain RFB RG −3 dB BW (MHz)
+1 750 Ω 140
+2 649 Ω 649 Ω 120
+10 511 Ω 56.2 Ω 100
−1 590 Ω 590 Ω 115
−10 511 Ω 51.1 Ω 95
VS = ±5 V
Closed-Loop Gain RFB RG −3 dB BW (MHz)
+1 619 Ω 80
+2 562 Ω 562 Ω 80
+10 442 Ω 48.7 Ω 65
−1 562 Ω 562 Ω 75
−10 442 Ω 44.2 Ω 65
VS = ±10 V
Closed-Loop Gain RFB RG −3 dB BW (MHz)
+1 649 Ω 105
+2 590 Ω 590 Ω 105
+10 499 Ω 49.9 Ω 80
−1 590 Ω 590 Ω 105
−10 499 Ω 49.9 Ω 80
Figure 17 and Figure 18 illustrate the relationship between the
feedback resistor and the frequency and time domain response
characteristics for a closed-loop gain of +2. (The response at
other gains is similar.)
The 3 dB bandwidth is somewhat dependent on the power
supply voltage. As the supply voltage is decreased, for example,
the magnitude of the internal junction capacitances is increased,
causing a reduction in closed-loop bandwidth. To compensate
for this, smaller values of feedback resistor are used at lower
supply voltages.
ACHIEVING THE FLATTEST GAIN RESPONSE AT
HIGH FREQUENCY
Achieving and maintaining gain flatness of better than 0.1 dB at
frequencies above 10 MHz requires careful consideration of
several issues.
Choice of Feedback and Gain Resistors
Because of the previously mentioned relationship between the
3 dB bandwidth and the feedback resistor, the fine scale gain
flatness varies, to some extent, with feedback resistor tolerance.
Therefore, it is recommended that resistors with a 1% tolerance
be used if it is desired to maintain flatness over a wide range of
production lots. In addition, resistors of different construction
have different associated parasitic capacitance and inductance.
Metal film resistors were used for the bulk of the character-
ization for this data sheet. It is possible that values other than
those indicated are optimal for other resistor types.
Printed Circuit Board Layout Considerations
As is expected for a wideband amplifier, PC board parasitics can
affect the overall closed-loop performance. Of concern are stray
capacitances at the output and the inverting input nodes. If a
ground plane is used on the same side of the board as the signal
traces, a space (3/16" is plenty) should be left around the signal
lines to minimize coupling. Additionally, signal lines connecting
the feedback and gain resistors should be short enough so that
their associated inductance does not cause high frequency gain
errors. Line lengths less than 1/4" are recommended.
Quality of Coaxial Cable
Optimum flatness when driving a coax cable is possible only
when the driven cable is terminated at each end with a resistor
matching its characteristic impedance. If the coax is ideal, then
the resulting flatness is not affected by the length of the cable.
While outstanding results can be achieved using inexpensive
cables, note that some variation in flatness due to varying cable
lengths may occur.
Power Supply Bypassing
Adequate power supply bypassing can be critical when optimiz-
ing the performance of a high frequency circuit. Inductance in
the power supply leads can form resonant circuits that produce
peaking in the amplifier’s response. In addition, if large current
transients must be delivered to the load, then bypass capacitors
(typically greater than 1 µF) are required to provide the best
settling time and lowest distortion. Although the recommended
0.1 µF power supply bypass capacitors are sufficient in many
applications, more elaborate bypassing (such as using two
paralleled capacitors) may be required in some cases.
Rev. G | Page 12 of 20
Data Sheet AD811
Driving Capacitive Loads
The feedback and gain resistor values in Table 3 result in very
flat closed-loop responses in applications where the load
capacitances are below 10 pF. Capacitances greater than this
result in increased peaking and overshoot, although not
necessarily in a sustained oscillation.
There are at least two very effective ways to compensate for this
effect. One way is to increase the magnitude of the feedback
resistor, which lowers the 3 dB frequency. The other method is
to include a small resistor in series with the output of the ampli-
fier to isolate it from the load capacitance. The results of these
two techniques are illustrated in Figure 38. Using a 1.5 kΩ
feedback resistor, the output ripple is less than 0.5 dB when
driving 100 pF. The main disadvantage of this method is that it
sacrifices a little bit of gain flatness for increased capacitive load
drive capability. With the second method, using a series resistor,
the loss of flatness does not occur.
Figure 37. Recommended Connection for Driving a Large Capacitive Load
Figure 38. Performance Comparison of Two Methods
for Driving a Capacitive Load
Figure 39. Recommended Value of Series Resistor vs.
the Amount of Capacitive Load
Figure 39 shows recommended resistor values for different load
capacitances. Refer again to Figure 38 for an example of the
results of this method. Note that it may be necessary to adjust
the gain setting resistor, RG, to correct for the attenuation which
results due to the divider formed by the series resistor, RS, and
the load resistance.
Applications that require driving a large load capacitance at a
high slew rate are often limited by the output current available
from the driving amplifier. For example, an amplifier limited to
25 mA output current cannot drive a 500 pF load at a slew rate
greater than 50 V/µs. However, because of the 100 mA output
current of the AD811, a slew rate of 200 V/µs is achievable
when driving the same 500 pF capacitor, as shown in Figure 40.
Figure 40. Output Waveform of an AD811 Driving a 500 pF Load.
Gain = +2, RFB = 649 Ω, RS = 15 Ω, RS = 10 kΩ
AD811
+
–
7
6
4
2
3
R
S
(OPTIONAL)
C
L
R
L
V
OUT
–V
S
+V
S
R
FB
R
G
R
T
V
IN
0.1µF
0.1µF
00866-E-038
–6
–3
0
3
6
9
12
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-039
V
S
= ±15V
C
L
= 100pF
R
L
= 10kΩ
GAIN = +2
R
FB
= 1.5kΩ
R
S
= 0
R
FB
= 649Ω
R
S
= 30Ω
0
10
20
30
40
50
60
70
80
90
100
VALUE OF R
S
(Ω)
LOAD CAPACITANCE (pF)
10 100 1000
00866-E-040
GAIN = +2
V
S
= ±15V
R
S
VALUE SPECIFIED
IS FOR FLATTEST
FREQUENCY RESPONSE
00866-E-041
10
90
100
0%
5V
2V 100ns
V
IN
V
OUT
Rev. G | Page 13 of 20
AD811 Data Sheet
OPERATION AS A VIDEO LINE DRIVER
The AD811 has been designed to offer outstanding performance
at closed-loop gains of +1 or greater, while driving multiple
reverse-terminated video loads. The lowest differential gain and
phase errors are obtained when using ±15 V power supplies.
With ±12 V supplies, there is an insignificant increase in these
errors and a slight improvement in gain flatness. Due to power
dissipation considerations, ±12 V supplies are recommended
for optimum video performance. Excellent performance can be
achieved at much lower supplies as well.
The closed-loop gain versus the frequency at different supply
voltages is shown in Figure 42. Figure 43 is an oscilloscope
photograph of an AD811 line driver’s pulse response with
±15 V supplies. The differential gain and phase error versus the
supply are plotted in Figure 44 and Figure 45, respectively.
Another important consideration when driving multiple cables
is the high frequency isolation between the outputs of the
cables. Due to its low output impedance, the AD811 achieves
better than 40 dB of output-to-output isolation at 5 MHz
driving back-terminated 75 Ω cables.
Figure 41. A Video Line Driver Operating at a Gain of +2
Figure 42. Closed-Loop Gain vs. Frequency, Gain = +2
Figure 43. Small Signal Pulse Response, Gain = +2, VS = ±15 V
Figure 44. Differential Gain Error vs. Supply Voltage for
the Video Line Driver of Figure 41
Figure 45. Differential Phase Error vs. Supply Voltage for
the Video Line Driver of Figure 41
VIN
–VS
+VS
0.1µF
0.1µF
AD811
VOUT No. 2
VOUT No. 1
75Ω CABLE
75Ω CABLE
75Ω CABLE
75Ω
75Ω
649Ω649Ω
75Ω75Ω
75Ω
+
–7
6
4
3
2
00866-E-042
–6
–3
0
3
6
9
12
GAIN (dB)
FREQUENCY (MHz)
1 10 100
00866-E-043
V
S
= ±15V
R
FB
= 649Ω
V
S
= ±5V
R
FB
= 562Ω
G = +2
R
L
= 150Ω
R
G
= R
FB
00866-E-044
10
90
100
0%
1V
1V10ns
V
IN
V
OUT
0
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
0.09
0.10
DIFFERENTIAL GAIN (%)
5 6 7 8 9 10 11 12 13 14 15
SUPPLY VOLTAGE (V)
00866-E-045
RF = 649Ω
FC = 3.58MHz
100 IRE
MODULATED RAMP
a. DRIVING A SINGLE, BACK-
TERMINATED, 75Ω COAX CABLE
b. DRIVING TWO PARALLEL, BACK-
TERMINATED, COAX CABLES
a
b
0
0.02
0.04
0.06
0.08
0.10
0.12
0.14
0.16
0.18
0.20
DIFFERENTIAL PHASE (DEGREES)
5 6 7 8 9 10 11 12 13 14 15
SUPPLY VOLTAGE (V)
00866-E-046
RF = 649Ω
FC = 3.58MHz
100 IRE
MODULATED RAMP
a. DRIVING A SINGLE, BACK-
TERMINATED, 75Ω COAX CABLE
b. DRIVING TWO PARALLEL, BACK-
TERMINATED, COAX CABLES
a
b
Rev. G | Page 14 of 20
Data Sheet AD811
Rev. G | Page 15 of 20
AN 80 MHZ VOLTAGE-CONTROLLED AMPLIFIER
CIRCUIT
The voltage-controlled amplifier (VCA) circuit of Figure 46 shows
the AD811 being used with the AD834, a 500 MHz, 4-quadrant
multiplier. The AD834 multiplies the signal input by the dc control
voltage, VG. The AD834 outputs are in the form of differential
currents from a pair of open collectors, ensuring that the full
bandwidth of the multiplier (which exceeds 500 MHz) is
available for certain applications. Here, the AD811 op amp
provides a buffered, single-ended, ground-referenced output.
Using feedback resistors R8 and R9 of 511 Ω, the overall gain
ranges from −70 dB for VG = 0 V to +12 V (a numerical gain of
+4) when VG = 1 V. The overall transfer function of the VCA is
VOUT = 4 (X1 − X2)(Y1 − Y2), which reduces to VOUT = 4 VG VIN
using the labeling conventions shown in Figure 46. The circuit’s
−3 dB bandwidth of 80 MHz is maintained essentially constant—
that is, independent of gain. The response can be maintained
flat to within ±0.1 dB from dc to 40 MHz at full gain with the
addition of an optional capacitor of about 0.3 pF across the
feedback resistor R8. The circuit produces a full-scale output of
±4 V for a ±1 V input and can drive a reverse-terminated load
of 50 Ω or 75 Ω to ±2 V.
The gain can be increased to 20 dB (×10) by raising R8 and R9
to 1.27 kΩ, with a corresponding decrease in −3 dB bandwidth
to approximately 25 MHz. The maximum output voltage under
these conditions is increased to ±9 V using ±12 V supplies.
The gain-control input voltage, VG, may be a positive or negative
ground-referenced voltage, or fully differential, depending on
the choice of connections at Pin 7 and Pin 8. A positive value of
VG results in an overall noninverting response. Reversing the sign
of VG simply causes the sign of the overall response to invert. In
fact, although this circuit has been classified as a voltage-controlled
amplifier, it is also quite useful as a general-purpose, four-quadrant
multiplier, with good load driving capabilities and fully
symmetrical responses from the X and Y inputs.
The AD811 and AD834 can both be operated from power supply
voltages of ±5 V. While it is not necessary to power them from
the same supplies, the common-mode voltage at W1 and W2
must be biased within the common-mode range of the input
stage of the AD811. To achieve the lowest differential gain and
phase errors, it is recommended that the AD811 be operated
from power supply voltages of ±10 V or greater. This VCA
circuit operates from a ±12 V dual power supply.
Figure 46. An 80 MHz Voltage-Controlled Amplifier
X2 X1 +V
S
W1
Y1 Y2 W2
–V
S
U1
AD834 U3
AD811
R4
182
R5
182
R1 100
R2 100
R3
249
R6
294
R7
294
R9*
R8*
R
L
FB
C2
0.1F
C1
0.1F
–12V
V
OUT
FB +12V
V
G
V
IN
*R8 = R9 = 511 FOR 4 GAIN
R8 = R9 = 1.27k FOR 10 GAIN
+
–
1234
8765
7
6
4
3
2
+
–
00866-E-047
AD811 Data Sheet
A VIDEO KEYER CIRCUIT
By using two AD834 multipliers, an AD811, and a 1 V dc source,
a special form of a two-input VCA circuit called a video keyer
can be assembled. Keying is the term used in reference to blending
two or more video sources under the control of a third signal or
signals to create such special effects as dissolves and overlays.
The circuit shown in Figure 47 is a two-input keyer, with video
inputs VA and VB, and a control input VG. The transfer function
(with VOUT at the load) is given by
VOUT = GVA + (1 − G)VB
where G is a dimensionless variable (actually, just the gain of the
A signal path) that ranges from 0 when VG = 0 to 1 when VG =
1 V. Thus, VOUT varies continuously between VA and VB as G
varies from 0 to 1.
Circuit operation is straightforward. Consider first the signal path
through U1, which handles video input VA. Its gain is clearly 0
when VG = 0, and the scaling chosen ensures that it has a unity
value when VG = 1 V; this takes care of the first term of the transfer
function. On the other hand, the VG input to U2 is taken to the
inverting input X2 while X1 is biased at an accurate 1 V. Thus,
when VG = 0, the response to video input VB is already at its
full-scale value of unity, whereas when VG = 1 V, the differential
input X1 − X2 is 0. This generates the second term.
The bias currents required at the output of the multipliers are
provided by R8 and R9. A dc level-shifting network comprising
R10/R12 and R11/R13 ensures that the input nodes of the
AD811 are positioned at a voltage within its common-mode
range. At high frequencies, C1 and C2 bypass R10 and R11,
respectively. R14 is included to lower the HF loop gain and is
needed because the voltage-to-current conversion in the
AD834s, via the Y2 inputs, results in an effective value of the
feedback resistance of 250 Ω; this is only about half the value
required for optimum flatness in the AD811’s response. (Note
that this resistance is unaffected by G: when G = +1, all the
feedback is via U1, while when G = 0 it is all via U2). R14
reduces the fractional amount of output current from the
multipliers into the current-summing inverting input of the
AD811 by sharing it with R8. This resistor can be used to adjust
the bandwidth and damping factor to best suit the application.
Figure 47. A Practical Video Keyer Circuit
U3
AD811
7
6
4
3
2
+
–
X2 X1 +VSW1
Y1 Y2 W2
–VS
U1
AD834
1234
8765
X2 X1 +VSW1
Y1 Y2 W2
–VS
U1
AD834
1234
8765
R8
29.4Ω
29.4Ω
R9
R12
6.98kΩ
6.98kΩ
R13
R10
2.49kΩ
C3
FB
VOUT
FB
VGR6
226Ω
R7
45.3Ω
+5V
–5V
+5V
U4
AD589
R5
113Ω
(0 TO +1V dc)
V
A (±1V FS)
–5V
R4
1.02kΩ
R3
100Ω
R2
174Ω
R1
1.87kΩ
VB (±1V FS)
+5V –5V
C1
0.1µF
0.1µF
0.1µF0.1µF
R14
SEE TEXT
+5V
–5V
C4
C2
R11
2.49kΩ
LOAD
GND
LOAD
GND
200Ω
TO Y2
TO PIN 6
AD811
SETUP FOR DRIVING
REVERSE-TERMINATED LOAD
ZO
ZO
200Ω
VOUT
INSET
00866-E-048
Rev. G | Page 16 of 20
Data Sheet AD811
To generate the 1 V dc needed for the 1 − G term, an AD589
reference supplies 1.225 V ± 25 mV to a voltage divider consisting
of resistors R2 through R4. Potentiometer R3 should be adjusted
to provide exactly 1 V at the X1 input.
In this case, an arrangement is shown using dual supplies of ±5 V
for both the AD834 and the AD811. Also, the overall gain is
arranged to be unity at the load when it is driven from a reverse-
terminated 75 Ω line. This means that the dual VCA has to operate
at a maximum gain of +2, rather than +4 as in the VCA circuit
of Figure 46. However, this cannot be achieved by lowering the
feedback resistor because below a critical value (not much less
than 500 Ω) the peaking of the AD811 may be unacceptable.
This is because the dominant pole in the open-loop ac response
of a current feedback amplifier is controlled by this feedback
resistor. It would be possible to operate at a gain of ×4 and then
attenuate the signal at the output. Instead, the signals have been
attenuated by 6 dB at the input to the AD811; this is the
function of R8 through R11.
Figure 48 is a plot of the ac response of the feedback keyer when
driving a reverse-terminated 50 Ω cable. Output noise and
adjacent channel feedthrough, with either channel fully off and
the other fully on, is about −50 dB to 10 MHz. The feedthrough
at 100 MHz is limited primarily by board layout. For VG = 1 V,
the −3 dB bandwidth is 15 MHz when using a 137 Ω resistor for
R14 and 70 MHz with R14 = 49.9 Ω. For more information on
the design and operation of the VCA and video keyer circuits,
refer to the AN-216 Application Note, Video VCAs and Keyers:
Using the AD834 and AD811 by Brunner, Clarke, and Gilbert,
available on the Analog Devices, Inc. website at www.analog.com.
Figure 48. A Plot of the AC Response of the Video Keyer
–90
–80
–70
–60
–50
–40
–30
–20
–10
0
10
CLOSED-LOOP GAIN (dB)
FREQUENCY (Hz)
100k10k 1M 10M 100M
00866-E-049
R14 = 49.9Ω
R14 = 137Ω
GAIN
ADJACENT CHANNEL
FEEDTHROUGH
Rev. G | Page 17 of 20
AD811 Data Sheet
Rev. G | Page 18 of 20
OUTLINE DIMENSIONS
Figure 49. 8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
Dimensions shown in inches and (millimeters)
Figure 50. 8-Lead Ceramic Dual In-Line Package [CERDIP]
(Q-8)
Dimensions shown in inches and (millimeters)
COM P L IANT TO JEDE C S TANDARDS MS- 001
CON T RO LL ING DIME NS I ONS ARE IN I NCH ES ; MIL L IME T E R DI M E NS ION S
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REF E RE NCE O NLY AND ARE NO T APP RO P RIATE F O R US E IN DESIG N.
COR NE R LEADS MAY BE CONFIGURED AS WHOL E OR HALF LEADS.
070606-A
0.02 2 (0. 56 )
0.01 8 (0. 46 )
0.01 4 (0. 36 )
SEATING
PLANE
0.015
(0.38)
MIN
0.21 0 ( 5.33)
MAX
0.15 0 ( 3.81)
0.13 0 ( 3.30)
0.115 (2.92)
0.07 0 ( 1.78)
0.06 0 ( 1.52)
0.04 5 ( 1.14)
8
14
5
0.28 0 (7.11)
0.25 0 (6. 35 )
0.24 0 (6. 10 )
0.100 (2.54)
BSC
0.40 0 ( 10.16)
0.36 5 ( 9.27)
0.35 5 ( 9.02)
0.060 (1.52)
MAX
0.430 (10.92)
MAX
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
0.32 5 ( 8.26)
0.31 0 ( 7.87)
0.30 0 ( 7.62)
0.19 5 ( 4.95)
0.13 0 ( 3.30)
0.115 (2.92)
0.01 5 ( 0.38)
GAUGE
PLANE
0.00 5 ( 0.13)
MIN
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
0.310 (7.87)
0.220 (5.59)
0.005 (0.13)
MIN 0.055 (1.40)
MAX
0.100 (2.54) BSC
15°
0°
0.320 (8.13)
0.290 (7.37)
0.015 (0.38)
0.008 (0.20)
SEATING
PLANE
0.200 (5.08)
MAX
0.405 (10.29) MAX
0.150 (3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36) 0.070 (1.78)
0.030 (0.76)
0.060 (1.52)
0.015 (0.38)
14
58
Data Sheet AD811
Rev. G | Page 19 of 20
Figure 51. 8-Lead Standard Small Outline Package [SOIC_N]
Narrow Body (R-8)
Dimensions shown in millimeters and (inches)
Figure 52. 20-Terminal Ceramic Leadless Chip Carrier [LCC]
(E-20-1)
Dimensions shown in inches and (millimeters)
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
COMPLIANT TO JEDEC STANDARDS MS-012-AA
012407-A
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
0.50 (0.0196)
0.25 (0.0099) 45°
8°
0°
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
4
1
85
5.00(0.1968)
4.80(0.1890)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
6.20 (0.2441)
5.80 (0.2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
CONTRO LL ING D IME NS IONS ARE IN INCHES ; MILLIMETER DIME NS IO N S
(IN PARENTHESES) ARE ROUNDED-OFF I NCH EQUIVALENTS FOR
REF ERENCE ONLY AND ARE NO T APPR OPRIATE F OR US E IN DE SIGN.
1
20 4
9
8
13
19
14
3
18
BOTTOM
VIEW
0.02 8 ( 0.71)
0.02 2 ( 0.56)
45° TYP
0.01 5 ( 0.38)
MIN
0.055 (1.40)
0.045 (1.14)
0.050 (1. 27)
BSC
0.07 5 ( 1.91)
REF
0.0 11 (0.28)
0.00 7 ( 0.18)
R TYP
0.09 5 ( 2.41)
0.07 5 ( 1.90)
0.100 (2.54) REF
0.20 0 ( 5.08)
REF
0.150 (3. 81)
BSC
0.075 (1.91)
REF
0.35 8 ( 9.09)
0.34 2 ( 8.69)
SQ
0.358
(9.09)
MAX
SQ
0.10 0 ( 2.54)
0.06 4 ( 1.63)
0.08 8 ( 2.24)
0.05 4 ( 1.37)
022106-A
AD811 Data Sheet
Rev. G | Page 20 of 20
Figure 53. 16-Lead Standard Small Outline Package [SOIC_W]
Wide Body (RW-16)
Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model1 Temperature Range Package Description Package Option
AD811ANZ −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
AD811AR-16 −40°C to +85°C 16-Lead Standard Small Outline Package [SOIC_W] RW-16
AD811ARZ-16 −40°C to +85°C 16-Lead Standard Small Outline Package [SOIC_W] RW-16
AD811ARZ-16-REEL −40°C to +85°C 16-Lead Standard Small Outline Package [SOIC_W] RW-16
AD811ARZ-16-REEL7 −40°C to +85°C 16-Lead Standard Small Outline Package [SOIC_W] RW-16
AD811JR-EBZ 8-Lead SOIC Evaluation Board
AD811JRZ 0°C to +70°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
AD811JRZ-REEL7 0°C to +70°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
AD811SQ/883B −55°C to +125°C 8-Lead Ceramic Dual In-Line Package [CERDIP] Q-8
AD811SE/883B −55°C to +125°C 20-Terminal Ceramic Leadless Chip Carrier [LCC] E-20-1
AD811ACHIPS −40°C to +85°C DIE
AD811SCHIPS −55°C to +125°C DIE
1 Z = RoHS Compliant Part.
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
COMPLIANT TO JEDEC STANDARDS MS-013-AA
10.50 (0.4134)
10.10 (0.3976)
0.30 (0.0118)
0.10 (0.0039)
2.65 (0.1043)
2.35 (0.0925)
10.65 (0.4193)
10.00 (0.3937)
7.60 (0.2992)
7.40 (0.2913)
0.75(0.0295)
0.25(0.0098)
45°
1.27 (0.0500)
0.40 (0.0157)
C
OPLANARITY
0.10 0.33 (0.0130)
0.20 (0.0079)
0.51 (0.0201)
0.31 (0.0122)
SEATING
PLANE
8°
0°
16 9
8
1
1.27 (0.0500)
BSC
03-27-2007-B
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D00866-0-4/15(G)
