General Description
MAX1846/MAX1847 high-efficiency PWM inverting
controllers allow designers to implement compact, low-
noise, negative-output DC-DC converters for telecom
and networking applications. Both devices operate
from +3V to +16.5V input and generate -500mV to
-200V output. To minimize switching noise, both devices
feature a current-mode, constant-frequency PWM control
scheme. The operating frequency can be set from 100kHz
to 500kHz through a resistor.
The MAX1846 is available in an ultra-compact 10-pin
µMAX® package. Operation at high frequency, com-
patibility with ceramic capacitors, and inverting topol-
ogy without transformers allow for a compact design.
Compatibility with electrolytic capacitors and flexibility to
operate down to 100kHz allow users to minimize the cost
of external components. The high-current output drivers
are designed to drive a P-channel MOSFET and allow the
converter to deliver up to 30W.
The MAX1847 features clock synchronization and shut-
down functions. The MAX1847 can also be configured to
operate as an inverting flyback controller with an N-channel
MOSFET and a transformer to deliver up to 70W. The
MAX1847 is available in a 16-pin QSOP package.
Current-mode control simplifies compensation and
provides good transient response. Accurate current-mode
control and over current protection are achieved through
low-side current sensing.
Applications
●Cellular Base Stations
●Networking Equipment
●Optical Networking Equipment
●SLIC Supplies
●CO DSL Line Driver Supplies
●Industrial Power Supplies
●Servers
●VOIP Supplies
Features
●90% Efficiency
●+3.0V to +16.5V Input Range
●-500mV to -200V Output
●Drives High-Side P-Channel MOSFET
●100kHz to 500kHz Switching Frequency
●Current-Mode, PWM Control
●Internal Soft-Start
●Electrolytic or Ceramic Output Capacitor
●The MAX1847 also offers:
Synchronization to External Clock Shutdown
N-Channel Inverting Flyback Option
µMAX is a registered trademark of Maxim Integrated Products, Inc.
Pin Configurations appear at end of data sheet.
19-2091; Rev 4; 7/16
+Denotes a lead(Pb)-free/RoHS-compliant package.
PART TEMP RANGE PIN-PACKAGE
MAX1846EUB -40°C to +85°C 10 µMAX
MAX1846EUB+ -40°C to +105°C 10 µMAX
MAX1847EEE -40°C to +85°C 16 QSOP
MAX1847EEE+ -40°C to +85°C 16 QSOP
MAX1846
MAX1847
VL IN
COMP
FREQ
REF
GND FB
PGND
CS
EXT
P
POSITIVE
VIN
NEGATIVE
VOUT
MAX1846/MAX1847 High-Efficiency, Current-Mode,
Inverting PWM Controller
Typical Operating Circuit
Ordering Information
EVALUATION KIT AVAILABLE
IN, SHDN to GND .................................................-0.3V to +20V
PGND to GND ......................................................-0.3V to +0.3V
VL to PGND for VIN ≤ 5.7V ........................-0.3V to (VIN + 0.3V)
VL to PGND for VIN > 5.7V .....................................-0.3V to +6V
EXT to PGND .............................................-0.3V to (VIN + 0.3V)
REF, COMP to GND ....................................-0.3V to (VL + 0.3V)
CS, FB, FREQ, POL, SYNC to GND ......................-0.3V to +6V
Continuous Power Dissipation (TA = +70°C)
10-Pin µMAX (derate 5.6mW/°C above +70°C) ..........444mW
16-Pin QSOP (derate 8.3mW/°C above +70°C)..........696mW
Operating Temperature Range ......................... -40°C to +105°C
Junction Temperature ......................................................+150°C
Storage Temperature Range ............................ -65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow)
Lead(Pb)-free...............................................................+260°C
Containing lead(Pb) ..................................................... +240°C
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C,
unless otherwise noted.)
PARAMETER CONDITIONS MIN TYP MAX UNITS
PWM CONTROLLER
Operating Input Voltage Range 3.0 16.5 V
UVLO Threshold
VIN rising -40°C to ~+85°C 2.8 2.95
V-40°C to ~+105°C 2.8 2.96
VIN falling -40°C to ~+85°C 2.6 2.74
-40°C to ~+105°C 2.59 2.74
UVLO Hysteresis 60 mV
FB Threshold No load -12 0 12 mV
FB Input Current VFB = -0.1V -50 -6 50 nA
Load Regulation CCOMP = 0.068µF, VOUT = -48V,
IOUT = 20mA to 200mA (Note 1) -1 0 %
Line Regulation CCOMP = 0.068µF, VOUT = -48V,
VIN = +8V to +16.5V, IOUT = 100mA 0.04 %
Current-Limit Threshold 85 100 115 mV
CS Input Current CS = GND 10 20 µA
Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V 0.75 1.2 mA
Shutdown Supply Current SHDN = GND, VIN = +3.0V to +16.5V
VIN = +3.0V to +16.5V 10 25 µA
REFERENCE AND VL REGULATOR
REF Output Voltage IREF = 50µA 1.236 1.25 1.264 V
REF Load Regulation IREF = 0 to 500µA -2 -15 mV
VL Output Voltage IVL = 100µA 3.85 4.25 4.65 V
VL Load Regulation IVL = 0.1mA to 2.0mA -20 -60 mV
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Electrical Characteristics
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Absolute Maximum Ratings
Note 1: Production test correlates to operating conditions.
Note 2: Guaranteed by design and characterization.
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C,
unless otherwise noted.)
PARAMETER CONDITIONS MIN TYP MAX UNITS
OSCILLATOR
Oscillator Frequency
RFREQ = 500kΩ ±1% 88 100 112
kHzRFREQ = 147kΩ ±1% 255 300 345
RFREQ = 76.8kΩ ±1% 500
Maximum Duty Cycle
RFREQ = 500kΩ ±1% 93 96 98
%RFREQ = 147kΩ ±1% 85 88 92
RFREQ = 76.8kΩ ±1% 80
SYNC Input Signal Duty-Cycle
Range 7 93 %
Minimum SYNC Input Logic-Low
Pulse Width 50 200 ns
SYNC Input Rise/Fall Time (Note 2) 200 ns
SYNC Input Frequency Range 100 550 kHz
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage 2.0 V
POL, SYNC, SHDN Input Low
Voltage 0.45 V
POL, SYNC Input Current POL, SYNC = GND or VL 20 40 µA
SHDN Input Current VSHDN = +5V or GND -12 -4 0 µA
VSHDN = +16.5V 1.5 6
SOFT-START
Soft-Start Clock Cycles 1024
Soft-Start Levels 64
EXT OUTPUT
EXT Sink/Source Current VIN = +5V, VEXT forced to +2.5V 1 A
EXT On-Resistance EXT high or low, tested with 100mA load, VIN = +5V 3 7.5 Ω
EXT high or low, tested with 100mA load, VIN = +3V 5 12
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Electrical Characteristics (continued)
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER CONDITIONS MIN MAX UNITS
PWM CONTROLLER
Operating Input Voltage Range 3.0 16.5 V
UVLO Threshold VIN rising 2.95 V
VIN falling 2.6
FB Threshold No load -20 +20 mV
FB Input Current VFB = -0.1V -40°C to ~+85°C -50 +50 nA
-40°C to ~+105°C -150 +150
Load Regulation CCOMP = 0.068µF, VOUT = -48V,
IOUT= 20mA to 200mA (Note 1) -2 0 %
Current Limit Threshold 85 115 mV
CS Input Current CS = GND 20 µA
Supply Current VFB = -0.1V, VIN = +3.0V to +16.5V 1.2 mA
Shutdown Supply Current SHDN = GND, VIN = +3.0V to +16.5V 25 µA
REFERENCE AND VL REGULATOR
REF Output Voltage IREF = 50µA 1.225 1.275 V
REF Load Regulation IREF = 0 to 500µA -15 mV
VL Output Voltage IVL = 100µA 3.85 4.65 V
VL Load Regulation IVL = 0.1mA to 2.0mA -60 mV
OSCILLATOR
Oscillator Frequency RFREQ = 500kΩ ±1% 84 116 kHz
RFREQ = 147kΩ ±1% 255 345
Maximum Duty Cycle RFREQ = 500kΩ ±1% 93 98 %
RFREQ = 147kΩ ±1% 84 93
SYNC Input Signal Duty-Cycle
Range 7 93 %
Minimum SYNC Input Logic Low
Pulse Width 200 ns
SYNC Input Rise/Fall Time (Note 2) 200 ns
SYNC Input Frequency Range 100 550 kHz
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage 2.0 V
POL, SYNC, SHDN Input Low
Voltage 0.45 V
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Electrical Characteristics
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1°F, TA = +25°C, unless otherwise
noted.)
Note 3: Parameters to -40°C are guaranteed by design and characterization.
(VSHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER CONDITIONS MIN MAX UNITS
POL, SYNC Input Current POL, SYNC = GND or VL 40 µA
SHDN Input Current VSHDN = +5V or GND -12 0 µA
VSHDN = +16.5V 6
EXT OUTPUT
EXT On-Resistance
EXT high or low, IEXT =
100mA, VIN = +5V
-40°C to ~+85°C 7.5
Ω
-40°C to ~+105°C 8.75
EXT high or low, IEXT = 100mA, VIN = +3V 12
1.238
1.246
1.242
1.254
1.250
1.258
1.262
-40 20 40-20
06
080 100
REFERENCE VOLTAGE
vs. TEMPERATURE
MAX1846/7 toc06
TEMPERATURE (C)
VREF (V)
0
0.4
0.2
0.8
0.6
1.0
1.2
1.4
1.6
0462810 12 14 16
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
MAX1846/7 toc05
VIN (V)
IIN (mA)
VFB = -0.1V
-12.10
-12.04
-12.06
-12.08
-12.02
-12.00
-11.98
-11.96
-11.94
-11.92
-11.90
0 200100 300 400 500 600
OUTPUT VOLTAGE LOAD REGULATION
MAX1846/7 toc04
LOAD CURRENT (mA)
OUTPUT VOLTAGE (V)
APPLICATION CIRCUIT B VIN = 5V
90
0
1 100010010
EFFICIENCY vs. LOAD CURRENT
30
10
70
50
100
40
20
80
60
MAX1846/7 toc03
LOAD CURRENT (mA)
EFFICIENCY (%)
VIN = 12V
VIN = 16.5V
APPLICATION CIRCUIT C VOUT = -48V
100
0
110 100 1000 10,000
EFFICIENCY vs. LOAD CURRENT
20
MAX1846/7 toc02
LOAD CURRENT (mA)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 5V
VIN = 3.3V
VOUT = -12V
APPLICATION CIRCUIT B
VIN = 3V
100
0
110 100 1000 10,000
EFFICIENCY vs. LOAD CURRENT
20
MAX1846/7 toc01
LOAD CURRENT (mA)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 5V
VIN = 16.5V
VOUT = -5V
APPLICATION CIRCUIT A
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Typical Operating Characteristics
Electrical Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1°F, TA = +25°C, unless otherwise
noted.)
IL
0
5V/div
1A/div
5V/div
VOUT
EXITING SHUTDOWN
MAX1846/7 toc15
APPLICATION CIRCUIT B
1ms/div
SHDN
0
40
20
60
120
140
100
80
160
0 2000 4000 6000 8000 10,000
EXT RISE/FALL TIME
vs. CAPACITANCE
MAX1846/7 toc14
CAPACITANCE (pF)
TIME (ns)
RISE TIME
FALL TIME
VIN = 12V
SWITCHING FREQUENCY
vs. TEMPERATURE
MAX1846/7 toc13
294
295
297
296
300
301
299
298
302
FREQUENCY (kHz)
-40 020-20 40 60 80 100
TEMPERATURE (C)
RFREQ = 147k 1%
0
100
300
200
400
500
0 200100 300 400 500 600
SWITCHING FREQUENCY
vs. RFREQ
MAX1846/7 toc12
RFREQ (k)
fOSC (kHz)
0
4
2
8
6
12
10
14
-40 020-20 40 60 80 100
OPERATING CURRENT
vs. TEMPERATURE
MAX1846/7 toc11
TEMPERATURE (C)
OPERATING CURRENT (mA)
A
B
C
A: VIN = 3V, VOUT = -12V
B: VIN = 5V, VOUT = -5V
C: VIN = 16.5V, VOUT = -5V
APPLICATION CIRCUIT A
SHUTDOWN SUPPLY CURRENT
vs. TEMPERATURE
MAX1846/7 toc10
0
2
6
4
12
14
10
8
16
SHUTDOWN SUPPLY CURRENT (A)
-40 020-20 40 60 80 100
TEMPERATURE (C)
VIN = 10V
VIN = 16.5V
VIN = 3V
4.22
4.23
4.24
4.25
4.26
4.27
0 0.8 1.00.4 0.60.2 1.2 1.4 1.6 1.8 2.0
VL LOAD REGULATION
MAX1846/7 toc09
IVL (mA)
VL (V)
4.100
4.180
4.140
4.260
4.220
4.300
4.340
-40 20 40-20
06
080 100
VL VOLTAGE
vs. TEMPERATURE
MAX1846/7 toc08
TEMPERATURE (C)
VL (V)
IVL = 0
1.240
1.245
1.255
1.250
1.260
0 100 200 300 400 500
REFERENCE LOAD REGULATION
MAX1846/7 toc07
IREF (A)
VREF (V)
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1°F, TA = +25°C, unless otherwise
noted.)
IL
500mA/div
200mV/div
ILOAD
V
OUT
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc20
APPLICATION CIRCUIT C
400s/div
ILOAD = 4mA to 100mA
IL1A/div
500mV/div
ILOAD
VOUT
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc19
APPLICATION CIRCUIT B
2ms/div
ILOAD = 10mA to 400mA
IL
LX
1A/div
10V/div
100mV/div
VOUT
LIGHT-LOAD SWITCHING
WAVEFORM
MAX1846/7 toc18
APPLICATION CIRCUIT B
1s/div
ILOAD = 50mA
IL
LX
1A/div
10V/div
100mV/div
V
OUT
HEAVY-LOAD SWITCHING
WAVEFORM
MAX1846/7 toc17
APPLICATION CIRCUIT B
1s/div
ILOAD = 600mA
IL
0
5V/div
1A/div
5V/div
VOUT
ENTERING SHUTDOWN
MAX1846/7 toc16
APPLICATION CIRCUIT B
1ms/div
SHDN
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Typical Operating Characteristics (continued)
PIN NAME FUNCTION
MAX1846 MAX1847
— 1 POL
Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external
PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS low-
side FET in transformer-based applications.
1 2 VL VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND.
2 3 FREQ
Oscillator Frequency Set Input. Connect a resistor (RFREQ) from FREQ to GND to set the
internal oscillator frequency from 100kHz (RFREQ = 500kW) to 500kHz (RFREQ = 76.8kW).
RFREQ is still required if an external clock is used at SYNC. See Setting the Operating
Frequency section.
3 4 COMP Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network
from COMP to GND for loop compensation. See Design Procedure.
4 5 REF 1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic
capacitor from REF to GND.
5 6 FB Feedback Input. Connect FB to the center of a resistor-divider connected between the
output and REF. The FB threshold is 0.
— 7, 9 N.C. No Connection
— 8 SHDN Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect
to IN for normal operation.
6 10, 11 GND Analog Ground. Connect to PGND.
7 12 PGND Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND.
8 13 CS Positive Current-Sense Input. Connect a current-sense resistor (RCS) between CS and
PGND.
9 14 EXT External MOSFET Gate-Driver Output. EXT swings from IN to PGND.
10 15 IN Power-Supply Input
— 16 SYNC
Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set
the internal oscillator frequency with RFREQ. Drive SYNC with a logic-level clock input
signal to externally set the converter’s operating frequency. DC-DC conversion cycles
initiate on the rising edge of the input clock signal. Note that when driving SYNC with an
external signal, RFREQ must still be connected to FREQ.
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Pin Description
MAX1847
REF FB
GND
CS
EXT
PGND
FREQ
COMP
VL IN
0.22µF
150kΩ
0.1µF
R2
10.0kΩ
1%
R1
95.3kΩ
1%
0.02Ω
1W
10µH
DO5022P-103
CMSH5-40
47µF
16V
FDS6375
3 x 22µF
10V
POL
SYNC
SHDN
10kΩ
47µF
16V
22kΩ
1
2
16
8
3
4
5
10, 11
6
12
13
14
15
7, 9
N.C.
VIN
+3V to +5.5V
0.47µFVOUT
-12V AT 400mA
SANYO
16TPB47M
1200pF
220pF
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Typical Application Circuit
MAX1846
MAX1847
STARTUP
CIRCUITRY
CONTROL
CIRCUITRY
VL
REGULATOR
OSCILLATOR
REFERENCE
SLOPE
COMP
UNDER-
VOLTAGE
LOCK OUT
SOFT-START
IN
MAX1847 ONLY
MAX1847 ONLY
POL
FREQ
COMP
FB
REF
SYNC
SHDN
ERROR
AMPLIFIER
CURRENT-
SENSE
AMPLIFIER
GND
ERROR
COMPARATOR
EXT DRIVER
EXT
VL
CS
PGND
PGND
GM
X3.3
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Functional Diagram
Detailed Description
The MAX1846/MAX1847 current-mode PWM controllers
use an inverting topology that is ideal for generating
output voltages from -500mV to -200V. Features include
shutdown, adjustable internal operating frequency or
synchronization to an external clock, soft-start, adjustable
current limit, and a wide (+3V to +16.5V) input range.
PWM Controller
The architecture of the MAX1846/MAX1847 current-mode
PWM controller is a BiCMOS multi-input system that
simultaneously processes the output-error signal, the
current-sense signal, and a slope-compensation ramp
(Functional Diagram). Slope compensation prevents sub-
harmonic oscillation, a potential result in current-mode
regulators operating at greater than 50% duty cycle. The
controller uses fixed-frequency, current-mode operation
where the duty ratio is set by the input-to-output voltage
ratio. The current-mode feedback loop regulates peak
inductor current as a function of the output error signal.
Internal Regulator
The MAX1846/MAX1847 incorporate an internal low-
dropout regulator (LDO). This LDO has a 4.25V output
and powers all MAX1846/MAX1847 functions (excluding
EXT) for the primary purpose of stabilizing the perfor-
mance of the IC over a wide input voltage range (+3V to
+16.5V). The input to this regulator is connected to IN,
and the dropout voltage is typically 100mV, so that when
VIN is less than 4.35V, VL is typically VIN minus 100mV.
When the LDO is in dropout, the MAX1846/MAX1847 still
operate with VIN as low as 3V. For best performance, it is
recommended to connect VL to IN when the input supply
is less than 4.5V.
Undervoltage Lockout
The MAX1846/MAX1847 have an undervoltage lockout
circuit that monitors the voltage at VL. If VL falls below
the UVLO threshold (2.8V typ), the control logic turns
the P-channel FET off (EXT high impedance). The rest
of the IC circuitry is still powered and operating. When
VL increases to 60mV above the UVLO threshold, the IC
resumes operation from a start up condition (soft-start).
Soft-Start
The MAX1846/MAX1847 feature a “digital” soft-start
that is preset and requires no external capacitor. Upon
startup, the FB threshold decrements from the refer-
ence voltage to 0 in 64 steps over 1024 cycles of fOSC
or fSYNC. See the Typical Operating Characteristics for
a scope picture of the soft-start operation. Soft-start is
implemented: 1) when power is first applied to the IC,
2) when exiting shutdown with power already applied, and
3) when exiting undervoltage lockout.
Shutdown (MAX1847 only)
The MAX1847 shuts down to reduce the supply current
to 10µA when SHDN is low. In this mode, the internal
reference, error amplifier, comparators, and biasing
circuitry turn off. The EXT output becomes high imped-
ance and the external pullup resistor connected to EXT
pulls VEXT to VIN, turning off the P-channel MOSFET.
When in shutdown mode, the converter's output goes to 0.
Frequency Synchronization
(MAX1847 only)
The MAX1847 is capable of synchronizing its switching
frequency with an external clock source. Drive SYNC
with a logic-level clock input signal to synchronize the
MAX1847. A switching cycle starts on the rising edge
of the signal applied to SYNC. Note that the frequen-
cy of the signal applied to SYNC must be higher than
the default frequency set by RFREQ. This frequency
is required so that the internal clock does not start a
switching cycle prematurely. If SYNC is inactive for an
entire clock cycle of the internal oscillator, the internal
oscillator takes over the switching operation. Choose
RFREQ such that fOSC = 0.9 5 fSYNC.
EXT Polarity (MAX1847 only)
The MAX1847 features an option to utilize an N-channel
MOSFET configuration, rather than the typical p-channel
MOSFET configuration (Figure 1). In order to drive the
different polarities of these MOSFETs, the MAX1847
is capable of reversing the phase of EXT by 180
degrees. When driving a P-channel MOSFET, connect
POL to GND. When driving an n-channel MOSFET,
connect POL to VL. These POL connections ensure the
proper polarity for EXT. For design guidance in regard to
this application, refer to the MAX1856 data sheet.
Design Procedure
Initial Specications
In order to start the design procedure, a few parameters
must be identified: the minimum input voltage expect-
ed (VIN(MIN)), the maximum input voltage expected
(VIN(MAX)), the desired output voltage (VOUT), and the
expected maximum load current (ILOAD).
Calculate the Equivalent Load Resistance
This is a simple calculation used to shorten the verifica-
tion equations:
RLOAD = VOUT / ILOAD
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Calculate the Duty Cycle
The duty cycle is the ratio of the on-time of the MOSFET
switch to the oscillator period. It is determined by the
ratio of the input voltage to the output voltage. Since
the input voltage typically has a range of operation, a
minimum (DMIN) and maximum (DMAX) duty cycle is
calculated by:
OUT D
MIN IN(MAX) SW LIM OUT D
VV
D
V VVV V
−
−−
+
=
−+
OUT D
MAX IN(MIN) SW LIM OUT D
VV
D
V VVV V
−
−−−
+
=+
where VD is the forward drop across the output diode,
VSW is the drop across the external FET when on,
and VLIM is the current-limit threshold. To begin with,
assume VD = 0.5V for a Schottky diode, VSW = 100mV,
and VLIM = 100mV. Remember that VOUT is negative
when using this formula.
Setting the Output Voltage
The output voltage is set using two external resistors to
form a resistive-divider to FB between the output and
REF (refer to R1 and R2 in Figure 1). VREF is nominally
1.25V and the regulation voltage for FB is nominally 0.
The load presented to the reference by the feedback
resistors must be less than 500µA to guarantee that
VREF is in regulation (see Electrical Characteristics
Table). Conversely, the current through the feedback
resistors must be large enough so that the leakage
current of the FB input (50nA) is insignificant. Therefore,
select R2 so that IR2 is between 50µA and 250µA.
IR2 = VREF / R2
where VREF = 1.25V. A typical value for R2 is 10kW.
Once R2 is selected, calculate R1 with the following
equation:
R1 = R2 x (-VOUT / VREF)
Setting the Operating Frequency
The MAX1846/MAX1847 are capable of operating at
switching frequencies from 100kHz to 500kHz. Choice
of operating frequency depends on a number of factors:
1) Noise considerations may dictate setting (or
synchronizing) fOSC above or below a certain
frequency or band of frequencies, particularly in RF
applications.
Figure 1. Using an N-Channel MOSFET (MAX1847 only)
REF
COMP
0.033µF270kΩ
SYNC
150kΩ
GND
FREQ
VL
VIN
+12V
VOUT
-48V AT 100mA
12µF
100V
12µF
25V
2
14
13
12
6
5
3
4
0.05Ω
0.5W
1800pF
15
0.47µF
0.1µF
10, 11
EXT
PGND
IN
CS
FB
N.C.
MAX1847
10.0kΩ
1%
383kΩ
1%
7, 9
POL
1
8
16
VP1-0190
12.2µH1:4
CMR1U-02
470Ω
100pF
100V
SHDN
IRLL2705
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
2) Higher frequencies allow the use of smaller value
(hence smaller size) inductors and capacitors.
3) Higher frequencies consume more operating power
both to operate the IC and to charge and discharge
the gate at the external FET, which tends to reduce the
efficiency at light loads.
4) Higher frequencies may exhibit lower overall efficiency
due to more transition losses in the FET; however, this
shortcoming can often be nullified by trading some
of the inductor and capacitor size benefits for lower-
resistance components.
5) High-duty-cycle applications may require lower fre-
quencies to accommodate the controller minimum
off-time of 0.4µs. Calculate the maximum oscillator
frequency with the following formula:
IN(MIN) SW LIM
OSC(MAX) IN(MIN) SW LIM OUT D
OFF(MIN)
V VV
fV VVV V
1
t
−−
−−−
=+
×
Remember that VOUT is negative when using this formula.
When running at the maximum oscillator frequency
(fOSCILLATOR) and maximum duty cycle (DMAX), do
not exceed the minimum value of DMAX stated in the
Electrical Characteristics table. For designs that exceed
the DMAX and fOSC(MAX), an autotransformer can reduce
the duty cycle and allow higher operating frequencies.
The oscillator frequency is set by a resistor, RFREQ,
which is connected from FREQ to GND. The relation-
ship between fOSC (in Hz) and RFREQ (in W) is slightly
nonlinear, as illustrated in the Typical Operating
Characteristics. Choose the resistor value from the graph
and check the oscillator frequency using the following
formula:
( ) ( ) ( )
( )
2
7 11 19
OSC
FREQ FREQ
1
f
5.21 10 1.92 10 R 4.86 10 R
−− −
−
=
× +× × × ×
External Synchronization (MAX1847 only)
The SYNC input provides external-clock synchroniza-
tion (if desired). When SYNC is driven with an exter-
nal clock, the frequency of the clock directly sets the
MAX1847's switching frequency. A rising clock edge on
SYNC is interpreted as a synchronization input. If the
sync signal is lost, the internal oscillator takes over at
the end of the last cycle, and the frequency is returned
to the rate set by RFREQ. Choose RFREQ such that
fOSC = 0.9 x fSYNC.
Choosing Inductance Value
The inductance value determines the operation of the
current-mode regulator. Except for low-current applica-
tions, most circuits are more efficient and economical
operating in continuous mode, which refers to continu-
ous current in the inductor. In continuous mode there is
a trade-off between efficiency and transient response.
Higher inductance means lower inductor ripple current,
lower peak current, lower switching losses, and, there-
fore, higher efficiency. Lower inductance means higher
inductor ripple current and faster transient response. A
reasonable compromise is to choose the ratio of inductor
ripple current to average continuous current at mini-
mum duty cycle to be 0.4. Calculate the inductor ripple
with the following formula:
( )
( )
RIPPLE
LOAD(MAX) IN(MAX) SW LIM OUT D
IN(MAX) SW LIM
I
0.4 I V V V V V
V VV
−−−
−−
=
×× +
Then calculate an inductance value:
L = (VIN(MAX) / IRIPPLE) x (DMIN / fOSC)
Choose the closest standard value. Once again, remem-
ber that VOUT is negative when using this formula.
Determining Peak Inductor Current
The peak inductor current required for a particular output
is:
ILPEAK = ILDC + (ILPP / 2)
where ILDC is the average DC inductor current and ILPP
is the inductor peak-to-peak ripple current. The ILDC and
ILPP terms are determined as follows:
( )
( )
()
LOAD
LDC MAX
SW LIM MAX
IN MIN
LPP OSC
I
I1 D
V V V x D
IL x f
=−
−−
=
where L is the selected inductance value. The
saturation rating of the selected inductor should meet
or exceed the calculated value for ILPEAK, although
most coil types can be operated up to 20% over their
saturation rating without difficulty. In addition to the sat-
uration criteria, the inductor should have as low a series
resistance as possible. For continuous inductor current,
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MAX1846/MAX1847 High-Efciency, Current-Mode,
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the power loss in the inductor resistance (PLR) is
approximated by:
2
LOAD
LR L
MAX
I
P ~R x
I D
−
where RL is the inductor series resistance.
Once the peak inductor current is calculated, the
current sense resistor, RCS, is determined by:
RCS = 85mV / ILPEAK
For high peak inductor currents (>1A), Kelvin-sensing
connections should be used to connect CS and PGND
to RCS. Connect PGND and GND together at the ground
side of RCS. A lowpass filter between RCS and CS may
be required to prevent switching noise from tripping the
current-sense comparator at heavy loads. Connect a
100W resistor between CS and the high side of RCS, and
connect a 1000pF capacitor between CS and GND.
Checking Slope-Compensation Stability
In a current-mode regulator, the cycle-by-cycle
stability is dependent on slope compensation to prevent
subharmonic oscillation at duty cycles greater than
50%. For the MAX1846/MAX1847, the internal slope
compensation is optimized for a minimum inductor value
(LMIN) with respect to duty cycle. For duty cycles greater
then 50%, check stability by calculating LMIN using the
following equation:
( )
( ) ( )
MIN IN(MIN) CS S
MAX MAX
L V xR /M
x 2xD 1 / 1 D
−−
=
where VIN(MIN) is the minimum expected input voltage,
Ms is the Slope Compensation Ramp (41 mV/µs) and
DMAX is the maximum expected duty cycle. If LMIN is
larger than L, increase the value of L to the next standard
value that is larger than LMIN to ensure slope compensa-
tion stability.
Choosing the Inductor Core
Choosing the most cost-effective inductor usually requires
optimizing the field and flux with size. With higher output
voltages the inductor may require many turns, and this
can drive the cost up. Choosing an inductor value at LMIN
can provide a good solution if discontinuous inductor
current can be tolerated. Powdered iron cores can pro-
vide the most economical solution but are larger in size
than ferrite.
Power MOSFET Selection
The MAX1846/MAX1847 drive a wide variety of P-channel
power MOSFETs (PFETs). The best performance,
especially with input voltages below 5V, is achieved with
low-threshold PFETs that specify on-resistance with
a gate-to-source voltage (VGS) of 2.7V or less. When
selecting a PFET, key parameters include:
●Total gate charge (QG)
●Reverse transfer capacitance (CRSS)
●On-resistance (RDS(ON))
●Maximum drain-to-source voltage (VDS(MAX))
●Minimum threshold voltage (VTH(MIN))
At high-switching rates, dynamic characteristics (para-
meters 1 and 2 above) that predict switching losses
may have more impact on efficiency than RDS(ON),
which predicts DC losses. QG includes all capacitance
associated with charging the gate. In addition, this
parameter helps predict the current needed to drive the
gate at the selected operating frequency. The power
MOSFET in an inverting converter must have a high
enough voltage rating to handle the input voltage plus
the magnitude of the output voltage and any spikes
induced by leakage inductance and ringing.
An RC snubber circuit across the drain to ground might be
required to reduce the peak ringing and noise.
Choose RDS(ON)(MAX) specified at VGS < VIN(MIN) to be
one to two times RCS. Verify that VIN(MAX) < VGS(MAX)
and VDS(MAX) > VIN(MAX) - VOUT + VD. Choose the rise-
and-fall times (tR, tF) to be less than 50ns.
Output Capacitor Selection
The output capacitor (COUT) does all the filtering in an
inverting converter. The output ripple is created by the
variations in the charge stored in the output capacitor with
each pulse and the voltage drop across the capacitor’s
equivalent series resistance (ESR) caused by the current
into and out of the capacitor. There are two properties
of the output capacitor that affect ripple voltage: the
capacitance value, and the capacitor’s ESR. The output
ripple due to the output capacitor’s value is given by:
VRIPPLE-C = (ILOAD × DMAX × TOSC ) / COUT
The output ripple due to the output capacitor’s ESR is
given by:
VRIPPLE-R = ILPP × RESR
These two ripple voltages are additive and the total output
ripple is:
VRIPPLE-T = VRIPPLE-C + VRIPPLE-R
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MAX1846/MAX1847 High-Efciency, Current-Mode,
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The ESR-induced ripple usually dominates this last
equation, so typically output capacitor selection is based
mostly upon the capacitor's ESR, voltage rating, and
ripple current rating. Use the following formula to deter-
mine the maximum ESR for a desired output ripple volt-
age (VRIPPLE-D):
RESR = VRIPPLE-D / ILPP
Select a capacitor with ESR rating less than RESR. The
value of this capacitor is highly dependent on dielectric
type, package size, and voltage rating. In general, when
choosing a capacitor, it is recommended to use low-ESR
capacitor types such as ceramic, organic, or tanta-
lum capacitors. Ensure that the selected capacitor has
sufficient margin to safely handle the maximum RMS
ripple current.
For continuous inductor current the maximum RMS ripple
current in the output filter capacitor is:
2
LOAD
RMS MAX MAX
MAX
I
I xD D
ID −=−
Choosing Compensation Components
The MAX1846/MAX1847 are externally loop-compensat-
ed devices. This feature provides flexibility in designs to
accommodate a variety of applications. Proper compen-
sation of the control loop is important to prevent excessive
output ripple and poor efficiency caused by instability. The
goal of compensation is to cancel unwanted poles and
zeros in the DC-DC converter’s transfer function created
by the power-switching and filter elements. More precise-
ly, the objective of compensation is to ensure stability by
ensuring that the DC-DC converter’s phase shift is less
than 180° by a safe margin, at the frequency where the
loop gain falls below unity. One method for ensuring ade-
quate phase margin is to introduce corresponding zeros
and poles in the feedback network to approximate a sin-
gle-pole response with a -20dB/decade slope all the way
to unity-gain crossover.
Calculating Poles and Zeros
The MAX1846/MAX1847 current-mode architecture takes
the double pole caused by the inductor and output
capacitor and shifts one of these poles to a much higher
frequency to make loop compensation easier. To compen-
sate these devices, we must know the center frequencies
of the right-half plane zero (zRHP) and the higher frequen-
cy pole (pOUT2). Calculate the zRHP frequency with the
following formula:
( )
( )
( )
2
MAX IN(MIN) OUT LOAD
RHP
OUT
1 D x V V xR
Z2 xV L
−− −
=π×
The calculations for pOUT2 are very complex. For most
applications where VOUT does not exceed -48V (in a
negative sense), the pOUT2 will not be lower than 1/8th
of the oscillator frequency and is generally at a higher
frequency than zRHP. Therefore:
pOUT2 ≥ 0.125 × fOSC
A pole is created by the output capacitor and the load
resistance. This pole must also be compensated and its
center frequency is given by the formula:
pOUT1 = 1 / (2π × RLOAD × COUT)
Finally, there is a zero introduced by the ESR of the
output capacitor. This zero is determined from the follow-
ing equation:
zESR = 1 / (2π × COUT × RESR)
Calculating the Required Pole Frequency
To ensure stability of the MAX1846/MAX1847, the gain
of the error amplifier must roll-off the total loop gain to
1 before ZRHP or POUT2 occurs. First, calculate the DC
open-loop gain ADC:
xx
M O MAX LOAD
DC
CS CS
B x G R (1 D ) R
AA xR
−
=
where:
ACS is the current-sense amplifier gain = 3.3
B is the feedback-divider attenuation factor =
R2
R1 R2+
GM is the error-amplifier transconductance =
400 µA/V
RO is the error-amplifier output resistance = 3 MW
RCS is the selected current-sense resistor
Determining the Compensation Component Values
Select a unity-gain crossover frequency (fCROS), which is
lower than zRHP and pOUT2 and higher than pOUT1. Using
fCROS, calculate the compensation resistor (RCOMP).
CROS O
COMP
DC OUT1 CROS
f xR
R
A xP f
=−
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MAX1846/MAX1847 High-Efciency, Current-Mode,
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Select the next smaller standard value of resistor and
then calculate the compensation capacitor required to
cancel out the output-capacitor-induced pole (POUT1)
determined previously.
Choose the next larger standard value of capacitor.
In order for pCOMP to compensate the loop, the open-
loop gain must reach unity at a lower frequency than the
right-half-plane zero or the second output pole, whichever
is lower in frequency. If the second output pole and the
right-half-plane zero are close together in frequency, the
higher resulting phase shift at unity gain may require
a lower crossover frequency. For duty cycles greater
than 50%, slope compensation reduces ADC, reducing
the actual crossover frequency from fCROS. It is also a
good practice to reduce noise on COMP with a capacitor
(CCOMP2) to ground. To avoid adding extra phase margin
at the crossover, the capacitor (CCOMP2) should roll-off
noise at five times the crossover frequency. The value for
CCOMP2 can be found using:
It might require a couple iterations to obtain a suitable
combination of compensation components.
Finally, the zero introduced by the output capacitor's
ESR must be compensated. This compensation is
accomplished by placing a capacitor between REF
and FB creating a pole directly in the feedback loop.
Calculate the value of this capacitor using the frequency
of zESR and the selected feedback resistor values with
the formula:
12
FB ESR OUT 12
RR
C R xC x R xR
+
=
When using low-ESR, ceramic chip capacitors (MLCCs)
at the output, calculate the value of CFB as follows:
12
FB
OSC 1 2
RR
C
2 3.14 f R R
+
=× × ××
Applications Information
Maximum Output Power
The maximum output power that the MAX1846/MAX1847
can provide depends on the maximum input power
available and the circuit's efficiency:
POUT(MAX) = Efficiency × PIN(MAX)
Furthermore, the efficiency and input power are both
functions of component selection. Efficiency losses can
be divided into three categories: 1) resistive losses across
the inductor, MOSFET on-resistance, current-sense resis-
tor, rectification diode, and the ESR of the input and out-
put capacitors; 2) switching losses due to the MOSFET's
transition region, and charging the MOSFET's gate
capacitance; and 3) inductor core losses. Typically,
80% efficiency can be assumed for initial calculations.
The required input power depends on the inductor
current limit, input voltage, output voltage, output current,
inductor value, and the switching frequency. The max-
imum output power is approximated by the following
formula:
PMAX = [VIN - (VLIM + ILIM x RDS(ON))] x ILIM x
[1 - (LIR / 2)] x [(-VOUT + VD) / (VIN - VSW - VLIM
- VOUT + VD)]
where ILIM is the peak current limit and LIR is the inductor
current-ripple ratio and is calculated by:
LIR = ILPP / ILDC
Again, remember that VOUT for the MAX1846/MAX1847
is negative.
Diode Selection
The MAX1846/MAX1847's high-switching frequency
demands a high-speed rectifier. Schottky diodes are
recommended for most applications because of their
fast recovery time and low forward voltage. Ensure that
the diode's average current rating exceeds the peak
inductor current by using the diode manufacturer's data.
Additionally, the diode's reverse breakdown voltage must
exceed the potential difference between VOUT and the
input voltage plus the leakage-inductance spikes. For
high output voltages (-50V or more), Schottky diodes
may not be practical because of this voltage requirement.
In these cases, use an ultrafast recovery diode with
adequate reverse-breakdown voltage.
COMP OUT1 COMP
1
C
6.28 x P xR
=
O COMP
COMP2 CROS O COMP
R R
C
5 x 6.28 x f x R x R
+
=
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Input Filter Capacitor
The input capacitor (CIN) must provide the peak current
into the inverter. This capacitor is selected the same way
as the output capacitor (COUT). Under ideal conditions,
the RMS current for the input capacitor is the same as the
output capacitor. The capacitor value and ESR must be
selected to reduce noise to an acceptable value and also
handle the ripple current (INRMS where:
O2
NRMS MAX MAX
MAX
1.2 xI
I xD D
(I D ) −
−
=
Bypass Capacitor
In addition to CIN and COUT, other ceramic bypass
capacitors are required with the MAX1846/MAX1847.
Bypass REF to GND with a 0.1µF or larger capacitor.
Bypass VL to GND with a 0.47µF or larger capacitor. All
bypass capacitors should be located as close to their
respective pins as possible.
PC Board Layout Guidelines
Good PC board layout and routing are required in high-
frequency-switching power supplies to achieve good
regulation, high efficiency, and stability. It is strongly
recommended that the evaluation kit PC board layouts
be followed as closely as possible. Place power components
as close together as possible, keeping their traces short,
direct, and wide. Avoid interconnecting the ground pins
of the power components using vias through an internal
ground plane. Instead, keep the power components close
together and route them in a “star” ground configuration
using component-side copper, then connect the star
ground to internal ground using multiple vias.
Main Application Circuits
The MAX1846/MAX1847 are extremely versatile devices.
Figure 2 shows a generic schematic of the MAX1846.
Table 1 lists component values for several typical
applications. These component values also apply to the
MAX1847. The first two applications are featured in the
MAX1846/MAX1847 EV kit.
Figure 2. MAX1846 Main Application Circuit
COMP
CCOMP RCOMP
RFREQ
GND
FREQ
VL
VIN
VOUT
COUT
CIN
1
9
8
7
5
4
2
3
P
L1
RCS
CFB
D1
10
22k
0.47µF
0.1µF
6
EXT
PGND
REF
IN
CS
FB
MAX1846
R2
R1
APPLICATION B
ONLY
NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.
CCOMP2
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MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Table 1. Component List for Main Application Circuits
Note: Indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.
SUPPLIER COMPONENT PHONE WEBSITE
AVX Capacitors 803-946-0690 www.avxcorp.com
Central Semiconductor Diodes 516-435-1110 www.centralsemi.com
Coilcraft Inductors 847-639-6400 www.coilcraft.com
Dale Resistors 402-564-3131 www.vishay.com/company/brands/dale/
Fairchild MOSFETs 408-721-2181 www.fairchildsemi.com
International Rectifier MOSFETs 310-322-3331 www.irf.com
IRC Resistors 512-992-7900 www.irctt.com
Kemet Capacitors 864-963-6300 www.kemet.com
On Semiconductor MOSFETs, Diodes 602-303-5454 www.onsemi.com
Panasonic Capacitors, resistors 201-348-7522 www.panasonic.com
Sanyo Capacitors 619-661-6835 www.secc.co.jp
Siliconix MOSFETs 408-988-8000 www.siliconix.com
Sprague Capacitors 603-224-1961 www.vishay.com/company/brands/sprague/
Sumida Inductors 847-956-0666 www.remtechcorp.com
Vitramon Resistors 203-268-6261 www.vishay.com/company/brands/vitramon/
CIRCUIT ID A B C D
Input (V) 12 3 to 5.5 12 12
Output (V) -5 -12 -48 -72
Output (A) 2 0.4 0.1 0.1
CCOMP (µF) 0.047 0.22 0.1 0.068
CIN (µF) 3 x 10 3 x 22 10 10
COUT (µF) 2 x 100 2 x 47 39 39
CFB (pF) 390 1200 1000 1000
R1 (kW) (1%) 40.2 95.3 383 576
R2 (kW) (1%) 10 10 10 10
RCOMP (kΩ) 8.2 10 220 470
RCS (W) 0.02 0.02 0.05 0.05
RFREQ (kW) 150 150 150 150
D1 CMSH5-40 CMSH5-40 CMR1U-02 CMR1U-02
L1 (µH) 10 10 47 82
P1 FDS6685 FDS6375 IRFR5410 IRFR5410
CCOMP2 (pF) 220 220 22 12
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MAX1846/MAX1847 High-Efciency, Current-Mode,
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Component Suppliers
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
10 µMAX U10+2 21-0061 90-0330
16 QSOP E16+1 21-0055 90-0167
QSOP
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
POL SYNC
IN
EXT
CS
PGND
GND
GND
N.C.
MAX1847
VL
FREQ
FB
COMP
REF
N.C.
1
++
2
3
4
5
10
9
8
7
6
IN
EXT
CS
PGNDREF
COMP
FREQ
VL
MAX1846
µMAX
TOP VIEW
GNDFB
SHDN
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MAX1846/MAX1847 High-Efciency, Current-Mode,
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Package Information
For the latest package outline information and land patterns
(footprints), go to www.maximintegrated.com/packages. Note
that a “+”, “#”, or “-” in the package code indicates RoHS status
only. Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
Chip Information
PROCESS: BiCMOS
Pin Congurations
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
2 9/10 Added equation in the Determining the Compensation Component Values section 16
3 3/14 Removed automotive application from the Applications section 1
4 7/16 Extended maximum operating temperature from +85°C to +105°C 1, 2, 4
© 2016 Maxim Integrated Products, Inc.
│
20
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
MAX1846/MAX1847 High-Efciency, Current-Mode,
Inverting PWM Controller
Revision History
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
