REV. D
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OP295/OP495
Dual/Quad Rail-to-Rail
Operational Amplifiers
PIN CONNECTIONS
FEATURES
Rail-to-Rail Output Swing
Single-Supply Operation: 3 V to 36 V
Low Offset Voltage: 300 mV
Gain Bandwidth Product: 75 kHz
High Open-Loop Gain: 1,000 V/mV
Unity-Gain Stable
Low Supply Current/Per Amplifier: 150 ␮A max
APPLICATIONS
Battery-Operated Instrumentation
Servo Amplifiers
Actuator Drives
Sensor Conditioners
Power Supply Control
GENERAL DESCRIPTION
Rail-to-rail output swing combined with dc accuracy are the
key features of the OP495 quad and OP295 dual CBCMOS
operational amplifiers. By using a bipolar front end, lower
noise and higher accuracy than that of CMOS designs has
been achieved. Both input and output ranges include the
negative supply, providing the user zero-in/zero-out capabil-
ity. For users of 3.3 V systems such as lithium batteries, the
OP295/OP495 is specified for 3 V operation.
Maximum offset voltage is specified at 300 µV for 5 V operation,
and the open-loop gain is a minimum of 1000 V/mV. This yields
performance that can be used to implement high accuracy systems,
even in single-supply designs.
The ability to swing rail-to-rail and supply 15 mA to the load
makes the OP295/OP495 an ideal driver for power transistors
and “H” bridges. This allows designs to achieve higher efficien-
cies and to transfer more power to the load than previously
possible without the use of discrete components. For applica-
tions such as transformers that require driving inductive loads,
increases in efficiency are also possible. Stability while driving
capacitive loads is another benefit of this design over CMOS
rail-to-rail amplifiers. This is useful for driving coax cable or
large FET transistors. The OP295/OP495 is stable with loads in
excess of 300 pF.
The OP295 and OP495 are specified over the extended industrial
(–40°C to +125°C) temperature range. OP295s are available
in 8-lead plastic DIP plus SOIC-8 surface-mount packages.
OP495s are available in 14-lead plastic and SOIC-16 surface-
mount packages.
8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
V–
V+
OUT B
–IN B
+IN B
1
2
3
45
6
7
8
OP295
14-Lead PDIP
(P Suffix)
1
2
3
4
5
6
7
14
13
12
11
10
9
8
OUT A
–IN A
+IN A
V+
+IN B
–IN B
OUT B
–IN D
+IN D
V–
+IN C
–IN C
OUT C
OUT D
OP495
8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
V–
V+
OUT B
–IN B
+IN B
1
2
3
45
6
7
8
OP295
14-Lead PDIP
(P Suffix)
TOP VIEW
(Not to Scale)
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
NC = NO CONNECT
OUT A
–IN A
+IN A
V+
+IN B
–IN B
OUT B
NC
OUT D
–IN D
+IN D
V–
+IN C
–IN C
OUT C
NC
OP495
REV. D–2–
OP295/OP495–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
OS
30 300 µV
–40°C ≤ T
A
≤ +125°C800 µV
Input Bias Current I
B
820 nA
–40°C ≤ T
A
≤ +125°C30nA
Input Offset Current I
OS
±1±3nA
–40°C ≤ T
A
≤ +125°C±5nA
Input Voltage Range V
CM
04.0 V
Common-Mode Rejection Ratio CMRR 0 V ≤ V
CM
≤ 4.0 V, –40°C ≤ T
A
≤ +125°C90110 dB
Large Signal Voltage Gain A
VO
R
L
= 10 kΩ, 0.005 ≤ V
OUT
≤ 4.0 V 1,000 10,000 V/mV
R
L
= 10 kΩ, –40°C ≤ T
A
≤ +125°C500 V/mV
Offset Voltage Drift ∆V
OS
/∆T15 µV/°C
OUTPUT CHARACTERISTICS
Output Voltage Swing High V
OH
R
L
= 100 kΩ to GND 4.98 5.0 V
R
L
= 10 kΩ to GND 4.90 4.94 V
I
OUT
= 1 mA, –40°C ≤ T
A
≤ +125°C4.7 V
Output Voltage Swing Low V
OL
R
L
= 100 kΩ to GND 0.7 2 mV
R
L
= 10 kΩ to GND 0.7 2 mV
I
OUT
= 1 mA, –40°C ≤ T
A
≤ +125°C90mV
Output Current I
OUT
±11 ±18 mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR ±1.5 V ≤ V
S
≤ ±15 V 90 110 dB
±1.5 V ≤ V
S
≤ ±15 V,
–40°C ≤ T
A
≤ +125°C85dB
Supply Current Per Amplifier I
SY
V
OUT
= 2.5 V, R
L
= ∞, –40°C ≤ T
A
≤ +125°C150 µA
DYNAMIC PERFORMANCE
Skew Rate SR R
L
= 10 kΩ0.03 V/µs
Gain Bandwidth Product GBP 75 kHz
Phase Margin θ
O
86 Degrees
NOISE PERFORMANCE
Voltage Noise e
n
p-p 0.1 Hz to 10 Hz 1.5 µV p-p
Voltage Noise Density e
n
f = 1 kHz 51 nV/√Hz
Current Noise Density i
n
f = 1 kHz <0.1 pA/√Hz
Specifications subject to change without notice.
ELECTRICAL CHARACTERISTICS
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
OS
100 500 µV
Input Bias Current I
B
820 nA
Input Offset Current I
OS
±1±3nA
Input Voltage Range V
CM
02.0 V
Common-Mode Rejection Ratio CMRR 0 V ≤ V
CM
≤ 2.0 V, –40°C ≤ T
A
≤ +125°C90110 dB
Large Voltage Gain A
VO
R
L
= 10 kΩ750 V/mV
Offset Voltage Drift ∆V
OS
/∆T 1 µV/°C
OUTPUT CHARACTERISTICS
Output Voltage Swing High V
OH
R
L
= 10 kΩ to GND 2.9 V
Output Voltage Swing Low V
OL
R
L
= 10 kΩ to GND 0.7 2 mV
POWER SUPPLY
Power Supply Rejection Ratio PSRR ±1.5 V ≤ V
S
≤ ±15 V 90 110 dB
±1.5 V ≤ V
S
≤ ±15 V,
–40°C ≤ T
A
≤ +125°C85dB
Supply Current Per Amplifier I
SY
V
OUT
= 1.5 V, R
L
= ∞, –40°C ≤ T
A
≤ +125°C150 µA
DYNAMIC PERFORMANCE
Slew Rate SR R
L
= 10 kΩ0.03 V/µs
Gain Bandwidth Product GBP 75 kHz
Phase Margin θ
O
85 Degrees
NOISE PERFORMANCE
Voltage Noise e
n
p-p 0.1 Hz to 10 Hz 1.6 µV p-p
Voltage Noise Density e
n
f = 1 kHz 53 nV/√Hz
Current Noise Density i
n
f = 1 kHz <0.1 pA/√Hz
Specifications subject to change without notice.
(@ V
S
= 5.0 V, V
CM
= 2.5 V, T
A
= 25ⴗC, unless otherwise noted.)
(@ V
S
= 3.0 V, V
CM
= 1.5 V, T
A
= 25ⴗC, unless otherwise noted.)
REV. D
OP295/OP495
–3–
ELECTRICAL CHARACTERISTICS
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
OS
300 500 µV
–40°C ≤ T
A
≤ +125°C800 µV
Input Bias Current I
B
V
CM
= 0 V 7 20 nA
V
CM
= 0 V, –40°C ≤ T
A
≤ +125°C30nA
Input Offset Current I
OS
V
CM
= 0 V ±1±3nA
V
CM
= 0 V, –40°C ≤ T
A
≤ +125°C±5nA
Input Voltage Range V
CM
–15 13.5 V
Common-Mode Rejection Ratio CMRR –15.0 V ≤ V
CM
≤ +13.5 V, –40°C ≤ T
A
≤ +125°C90 110 dB
Large Signal Voltage Gain A
VO
R
L
= 10 kΩ1,000 4,000 V/mV
Offset Voltage Drift ∆V
OS
/∆T 1 µV/°C
OUTPUT CHARACTERISTICS
Output Voltage Swing High V
OH
R
L
= 100 kΩ to GND 14.95 V
R
L
= 10 kΩ to GND 14.80 V
Output Voltage Swing Low V
OL
R
L
= 100 kΩ to GND –14.95 V
R
L
= 10 kΩ to GND –14.85 V
Output Current I
OUT
±15 ±25 mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR V
S
= ±1.5 V to ±15 V 90 110 dB
V
S
= ±1.5 V to ±15 V, –40°C ≤ T
A
≤ +125°C85 dB
Supply Current I
SY
V
O
= 0 V, R
L
= ∞, V
S
= ±18 V,
–40°C ≤ T
A
≤ +125°C175 µA
Supply Voltage Range V
S
3 (±1.5) 36 (±18) V
DYNAMIC PERFORMANCE
Slew Rate SR R
L
= 10 kΩ0.03 V/µs
Gain Bandwidth Product GBP 85 kHz
Phase Margin θ
O
83 Degrees
NOISE PERFORMANCE
Voltage Noise e
n
p-p 0.1 Hz to 10 Hz 1.25 µV p-p
Voltage Noise Density e
n
f =1 kHz 45 nV/√Hz
Current Noise Density i
n
f = 1 kHz <0.1 pA/√Hz
Specifications subject to change without notice.
(@ V
S
= ±15.0 V, T
A
= 25ⴗC, unless otherwise noted.)
REV. D–4–
OP295/OP495
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; and functional operation
of the device at these or any other conditions above those indicated in the
operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
Absolute maximum ratings apply to packaged parts, unless otherwise noted.
3
For supply voltages less than ±18 V, the absolute maximum input voltage is equal
to the supply voltage.
Package Type ␪
JA
*␪
JC
Unit
8-Lead Plastic DIP (P) 103 43 °C/W
8-Lead SOIC (S) 158 43 °C/W
14-Lead Plastic DIP (P) 83 39 °C/W
16-Lead SOIC (S) 98 30 °C/W
*␪
JA
is specified for worst case mounting conditions, i.e., ␪
JA
is specified for device
in socket for CERDIP, PDIP, and LCC packages; ␪
JA
is specified for device
soldered to printed circuit board for SOIC package.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
OP295/OP495 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
ABSOLUTE MAXIMUM RATINGS
1, 2
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
Input Voltage
2
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
Differential Input Voltage
3
. . . . . . . . . . . . . . . . . . . . . . . . . 36 V
Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Operating Temperature Range
OP295G, OP495G . . . . . . . . . . . . . . . . . . . –40°C to +125°C
Junction Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300°C
ORDERING GUIDE
Temperature Package Package
Model Range Description Option
OP295GP –40°C to +125°C8-Lead Plastic DIP P-8 (N-8)
OP295GS –40°C to +125°C8-Lead SOIC S-8 (R-8)
OP295GS-REEL –40°C to +125°C8-Lead SOIC S-8 (R-8)
OP295GS-REEL7 –40°C to +125°C8-Lead SOIC S-8 (R-8)
OP495GP –40°C to +125°C14-Lead Plastic DIP P-14 (N-14)
OP495GS –40°C to +125°C16-Lead SOIC S-16 (RW-16)
OP495GS-REEL –40°C to +125°C16-Lead SOIC S-16 (RW-16)
OP495GSZ*–40°C to +125°C16-Lead SOIC S-16 (RW-16)
OP495GSZ-REEL7*–40°C to +125°C16-Lead SOIC S-16 (RW-16)
*Z = Pb-free part.
Typical Performance Characteristics
140
20
100
80
40
–25
60
–50
120
100
7550250
TEMPERATURE – ⴗC
SUPPLY CURRENT – ␮A
VS = 5V
VS = 3V
VS = 36V
TPC 1. Supply Current Per Amplifier vs. Temperature
15.2
–15.2
100
–14.6
–15.0
–25
–14.8
–50
14.2
–14.4
14.4
14.6
14.8
15.0
7550250
TEMPERATURE – C
– OUTPUT SWING – V + OUTPUT SWING – V
VS = 15V RL = 100k⍀
RL = 2k⍀
RL = 10k⍀
RL = 2k⍀
RL = 10k⍀
RL = 100k⍀
TPC 2. Output Voltage Swing vs. Temperature
REV. D
OP295/OP495
–5–
3.10
2.50
100
2.80
2.60
–25
2.70
–50
3.00
2.90
7550250
TEMPERATURE – ⴗC
OUTPUT VOLTAGE SWING – V
VS = 3V
RL = 2k⍀
RL = 10k⍀
RL = 100k⍀
TPC 3. Output Voltage Swing vs. Temperature
200
0
250
50
25
–200–250
100
75
125
150
175
200150100500–50–100–150
VS = 5V
TA = 25ⴗC
INPUT OFFSET VOLTAGE – ␮V
UNITS
BASED ON 600 OP AMPS
TPC 4. OP295 Input Offset (V
OS
) Distribution
UNITS
250
0
3.2
75
25
0.4
50
0
150
100
125
175
200
225
2.82.42.01.61.20.8
BASED ON 600 OP AMPS V
S
= 5V
–40ⴗ T
A
+85ⴗC
T
C
– V
OS
– ␮V/ⴗC
TPC 5. OP295 TC–V
OS
Distribution
5.10
4.50
100
4.80
4.60
–25
4.70
–50
5.00
4.90
7550250
TEMPERATURE – ⴗC
OUTPUT VOLTAGE SWING – V
VS = 5V
RL = 2k⍀
RL = 10k⍀
RL = 100k⍀
TPC 6. Output Voltage Swing vs. Temperature
500
0
300
150
50
–50
100
–100
300
200
250
350
400
450
250200150100500
INPUT OFFSET VOLTAGE – ␮V
UNITS
V
S
= 5V
T
A
= 25ⴗC
BASED ON 1200 OP AMPS
TPC 7. OP495 Input Offset (V
OS
) Distribution
500
0
3.2
150
50
0.4
100
0
300
200
250
350
400
450
2.82.42.01.61.20.8
TC – V OS – ␮V/ⴗC
UNITS
VS = 5V
–40ⴗ TA +85ⴗC
BASED ON 1200 OP AMPS
TPC 8. OP495 TC–V
OS
Distribution
REV. D–6–
OP295/OP495
VS = 5V
20
0
100
12
4
–25
8
–50
16
7550250
TEMPERATURE – ⴗC
INPUT BIAS CURRENT – nA
TPC 9. Input Bias Current vs. Temperature
V
S
= 5V
TEMPERATURE – ⴗC
40
0
100
10
5
–25–50
20
15
25
30
35
7550250
V
S
= 15V
SOURCE
SINK
SOURCE
SINK
OUTPUT CURRENT – mA
TPC 10. Output Current vs. Temperature
TEMPERATURE – ⴗC
OPEN-LOOP GAIN – V/␮V
R
L
= 2k⍀
100
10
1
–50 25 100
0
–25 50 75
V
S
= 15V
V
O
= 10V
R
L
= 10k⍀
R
L
= 100k⍀
TPC 11. Open-Loop Gain vs. Temperature
12
0
100
6
2
–25
4
–50
10
8
7550250
TEMPERATURE – ⴗC
OPEN-LOOP GAIN – V/␮V
V
S
= 5V
V
O
= 4V
R
L
= 100k⍀
R
L
= 10k⍀
R
L
= 2k⍀
TPC 12. Open-Loop Gain vs. Temperature
1V
100␮V
1␮A10␮A10mA1mA
100␮A
100mV
10mV
1mV
LOAD CURRENT
OUTPUT VOLTAGE TO RAIL
SOURCE
SINK
VS = 5V
TA = 25ⴗC
TPC 13. Output Voltage to Supply Rail vs.
Load Current
REV. D
OP295/OP495
–7–
APPLICATIONS
Rail-to-Rail Application Information
The OP295/OP495 has a wide common-mode input range
extending from ground to within about 800 mV of the positive
supply. There is a tendency to use the OP295/OP495 in buffer
applications where the input voltage could exceed the common-
mode input range. This may initially appear to work because of
the high input range and rail-to-rail output range. But above the
common-mode input range, the amplifier is, of course, highly
nonlinear. For this reason, it is always required that there be
some minimal amount of gain when rail-to-rail output swing is
desired. Based on the input common-mode range, this gain
should be at least 1.2.
Low Drop-Out Reference
The OP295/OP495 can be used to gain up a 2.5 V or other low
voltage reference to 4.5 V for use with high resolution ADCs
that operate from 5 V only supplies. The circuit in Figure 1 will
supply up to 10 mA. Its no-load drop-out voltage is only 20 mV.
This circuit will supply over 3.5 mA with a 5 V supply.
16k⍀
1/2
OP295/OP495
V = 4.5V
OUT
1␮F TO
10 ␮F
10⍀
0.001␮F
20k⍀
REF43
2
6
4
5V
5V
Figure 1. 4.5 V, Low Drop-Out Reference
Low Noise, Single-Supply Preamplifier
Most single-supply op amps are designed to draw low supply
current at the expense of having higher voltage noise. This
tradeoff may be necessary because the system must be powered
by a battery. However, this condition is worsened because all
circuit resistances tend to be higher; as a result, in addition to the
op amp’s voltage noise, Johnson noise (resistor thermal noise) is
also a significant contributor to the total noise of the system.
The choice of monolithic op amps that combine the character-
istics of low noise and single-supply operation is rather limited.
Most single-supply op amps have noise on the order of 30 nV/√Hz
to 60 nV/√Hz and single-supply amplifiers with noise below
5 nV/√Hz do not exist.
In order to achieve both low noise and low supply voltage opera-
tion, discrete designs may provide the best solution. The circuit
in Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a
matched PNP transistor pair—the MAT03—to achieve zero-in/
zero-out single-supply operation with an input voltage noise of
3.1 nV/√Hz at 100 Hz. R5 and R6 set the gain of 1,000, making
this circuit ideal for maximizing dynamic range when amplifying
low level signals in single-supply applications. The OP295/OP495
provide rail-to-rail output swings, allowing this circuit to operate
with 0 V to 5 V outputs. Only half of the OP295/OP495 is used,
leaving the other uncommitted op amp for use elsewhere.
10␮F
0.1␮F
R1LED
V
IN
2
35
6
3
71
MAT-03
Q1 Q2
R7
510⍀
C1
1500pF R4
R8
100⍀
R3
R2
27k⍀
R5
10k⍀
C2
10␮F
V
OUT
OP295/OP495
R6
10⍀
28
4
1
Q2
2N3906
Figure 2. Low Noise Single-Supply Preamplifier
The input noise is controlled by the MAT03 transistor pair
and the collector current level. Increasing the collector current
reduces the voltage noise. This particular circuit was tested
with 1.85 mA and 0.5 mA of current. Under these two cases,
the input voltage noise was 3.1 nV/√Hz and 10 nV/√Hz, respec-
tively. The high collector currents do lead to a tradeoff in supply
current, bias current, and current noise. All of these parameters
increase with increasing collector current. For example, typi-
cally the MAT03 has an h
FE
= 165. This leads to bias currents
of 11 µA and 3 µA, respectively. Based on the high bias cur-
rents, this circuit is best suited for applications with low source
impedance such as magnetic pickups or low impedance strain
gages. Furthermore, a high source impedance degrades the noise
performance. For example, a 1 kΩ resistor generates 4 nV/√Hz
of broadband noise, which is already greater than the noise of
the preamp.
The collector current is set by R1 in combination with the LED
and Q2. The LED is a 1.6 V Zener diode that has a temperature
coefficient close to that of Q2’s base-emitter junction, which
provides a constant 1.0 V drop across R1. With R1 equal to 270 Ω,
the tail current is 3.7 mA and the collector current is half that,
or 1.85 mA. The value of R1 can be altered to adjust the collector
current. Whenever R1 is changed, R3 and R4 should also be
adjusted. To maintain a common-mode input range that includes
ground, the collectors of the Q1 and Q2 should not go above
0.5 V—otherwise they could saturate. Thus, R3 and R4 must
be small enough to prevent this condition. Their values and the
overall performance for two different values of R1 are summa-
rized in Table I. Lastly, the potentiometer, R8, is needed to
adjust the offset voltage to null it to zero. Similar performance
can be obtained using an OP90 as the output amplifier with a
savings of about 185 µA of supply current. However, the output
swing will not include the positive rail, and the bandwidth will
reduce to approximately 250 Hz.
REV. D–8–
OP295/OP495
Table I. Single-Supply Low Noise Preamp Performance
I
C
= 1.85 mA I
C
= 0.5 mA
R1 270 Ω1.0 kΩ
R3, R4 200 Ω910 Ω
e
n
@ 100 Hz 3.15 nV/√Hz 8.6 nV/√Hz
e
n
@ 10 Hz 4.2 nV/√Hz 10.2 nV/√Hz
I
SY
4.0 mA 1.3 mA
I
B
11 µA3 µA
Bandwidth 1 kHz 1 kHz
Closed-Loop Gain 1,000 1,000
Driving Heavy Loads
The OP295/OP495 is well suited to drive loads by using a power
transistor, Darlington, or FET to increase the current to the load.
The ability to swing to either rail can assure that the device is
turned on hard. This results in more power to the load and an
increase in efficiency over using standard op amps with their
limited output swing. Driving power FETs is also possible with
the OP295/OP495 because of its ability to drive capacitive loads
of several hundred picofarads without oscillating.
Without the addition of external transistors, the OP295/OP495
can drive loads in excess of ±15 mA with ±15 V or +30 V
supplies. This drive capability is somewhat decreased at lower
supply voltages. At ±5 V supplies, the drive current is ±11 mA.
Driving motors or actuators in two directions in a single-supply
application is often accomplished using an “H” bridge. The
principle is demonstrated in Figure 3a. From a single 5 V sup-
ply, this driver is capable of driving loads from 0.8 V to 4.2 V in
both directions. Figure 3b shows the voltages at the inverting
and non- inverting outputs of the driver. There is a small cross-
over glitch that is frequency dependent and would not cause
problems unless this was a low distortion application such as
audio. If this is used to drive inductive loads, be sure to add
diode clamps to protect the bridge from inductive kickback.
10k⍀
1.67V
2N2222 2N2222
OUTPUTS
2N2907 2N2907
5V
0 V
IN
2.5V 5k⍀
10k⍀10k⍀
Figure 3a. “H” Bridge
10
90
100
0%
2V 2V 1ms
Figure 3b. “H” Bridge Outputs
Direct Access Arrangement
OP295/OP495 can be used in a single-supply direct access
arrangement (DAA) as is shown in Figure 4. This figure shows a
portion of a typical DM capable of operating from a single 5 V
supply and it may also work on 3 V supplies with minor modifica-
tions. Amplifiers A2 and A3 are configured so that the transmit
signal TxA is inverted by A2 and is not inverted by A3. This
arrangement drives the transformer differentially so that the drive
to the transformer is effectively doubled over a single amplifier
arrangement. This application takes advantage of the OP295/
OP495’s ability to drive capacitive loads, and to save power in
single-supply applications.
2.5V REF
A3
750pF
0.1␮F
A1
37.4k⍀
390pF
RXA
TXA
OP295/
OP495
OP295/
OP495
A2
OP295/
OP495
0.0047␮F
3.3k⍀20k⍀
475⍀
22.1k⍀
20k⍀
0.1␮F
20k⍀
20k⍀
20k⍀
0.033␮F
1:1
Figure 4. Direct Access Arrangement
A Single-Supply Instrumentation Amplifier
The OP295/OP495 can be configured as a single-supply instru-
mentation amplifier as in Figure 5. For our example, V
REF
is set
equal to V+/2 and V
O
is measured with respect to V
REF
. The
input common-mode voltage range includes ground and the
output swings to both rails.
V+
V
IN
V
REF
1
R1
100k⍀
R2 R3 R4
V
O
V
IN
+ V
REF
V
O
= 5 + 200k⍀
R
G
R
G
3
2
8
7
4
5
6
20k⍀20k⍀100k⍀
1/2
OP295/
OP495
1/2
OP295/
OP495
Figure 5. Single-Supply Instrumentation Amplifier
Resistor R
G
sets the gain of the instrumentation amplifier. Mini-
mum gain is 6 (with no R
G
). All resistors should be matched in
absolute value as well as temperature coefficient to maximize
common-mode rejection performance and minimize drift. This
instrumentation amplifier can operate from a supply voltage as
low as 3 V.
REV. D
OP295/OP495
–9–
A Single-Supply RTD Thermometer Amplifier
This RTD amplifier takes advantage of the rail-to-rail swing of
the OP295/OP495 to achieve a high bridge voltage in spite of a
low 5 V supply. The OP295/OP495 amplifier servos a constant
200 µA current to the bridge. The return current drops across
the parallel resistors 6.19 kΩ and the 2.55 MΩ, developing a
voltage that is servoed to 1.235 V, which is established by the
AD589 band gap reference. The 3-wire RTD provides an
equal line resistance drop in both 100 Ω legs of the bridge,
thus improving the accuracy.
The AMP04 amplifies the differential bridge signal and converts
it to a single-ended output. The gain is set by the series resis-
tance of the 332 Ω resistor plus the 50 Ω potentiometer. The
gain scales the output to produce a 4.5 V full scale. The 0.22 µF
capacitor to the output provides a 7 Hz low-pass filter to keep
noise at a minimum.
ZERO ADJ
200⍀
10-TURNS
26.7k⍀
0.5%
26.7k⍀
0.5%
100⍀
RTD
100⍀
0.5%
2.55M⍀
1%
6.19k⍀
1%
AD589
37.4k⍀
5V
1.235
1
23
1/2
OP295/
OP495
3
2
4
5
6
8
1
7
5V
AMP04
50⍀
332⍀
0.22␮F
V
O
4.5V = 450ⴗC
0V = 0ⴗC
Figure 6. Low Power RTD Amplifier
A Cold Junction Compensated, Battery-Powered
Thermocouple Amplifier
The OP295/OP495’s 150 µA quiescent current per amplifier
consumption makes it useful for battery-powered temperature
measuring instruments. The K-type thermocouple terminates
into an isothermal block where the terminated junctions’ ambi-
ent temperatures can be continuously monitored and corrected
by summing an equal but opposite thermal EMF to the amplifier,
thereby canceling the error introduced by the cold junctions.
ISOTHERMAL
BLOCK
AD589
ALUMEL
CHROMEL
AL
CR
K-TYPE
THERMOCOUPLE
40.7␮V/ ⴗC
COLD
JUNCTIONS
1N914
1.5M⍀
1%
24.9k⍀
1%
7.15k⍀
1%
24.3k⍀
1%
4.99k⍀
1%
475⍀
1%
2.1k⍀
1%
ZERO
ADJUST
500⍀
10-TURN
OP295/
OP495
24.9k⍀
1.235V
9V
1.33M⍀
20k⍀
1
2
34
8
SCALE
ADJUST
V
O
5V = 500ⴗC
0V = 0ⴗC
Figure 7. Battery-Powered, Cold-Junction Compensated
Thermocouple Amplifier
To calibrate, immerse the thermocouple measuring junction in a
0°C ice bath, adjust the 500 Ω zero-adjust potentiometer to 0 V
out. Then immerse the thermocouple in a 250°C temperature
bath or oven and adjust the scale-adjust potentiometer for an
output voltage of 2.50 V, which is equivalent to 250°C. Within
this temperature range, the K-type thermocouple is quite accu-
rate and produces a fairly linear transfer characteristic. Accuracy
of ±3°C is achievable without linearization.
Even if the battery voltage is allowed to decay to as low as 7 V,
the rail-to-rail swing allows temperature measurements to 700°C.
However, linearization may be necessary for temperatures above
250°C where the thermocouple becomes rather nonlinear. The
circuit draws just under 500 µA supply current from a 9 V battery.
A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V
Figure 8 shows a complete voltage output DAC with wide out-
put voltage swing operating off a single 5 V supply. The serial
input 12-bit DAC is configured as a voltage output device with
the 1.235 V reference feeding the current output pin (I
OUT
) of
the DAC. The V
REF
, which is normally the input now becomes
the output.
The output voltage from the DAC is the binary weighted volt-
age of the reference, which is gained up by the output amplifier
such that the DAC has a 1 mV per bit transfer function.
5V 5V
R1
17.8k⍀
1.23V
AD589
3
4765
GND CLK SRI LD
V
DD
V
REF
R
FB
2
1
3
24
1
8
DIGITAL
CONTROL
5V
OP295/
OP495
R3
5k⍀
R2
41.2k⍀R4
100k⍀
D
4096
V
O
= (4.096V)
TOTA L POWER DISSIPATION = 1.6mW
I
OUT
8
DAC8043
Figure 8. A 5 V 12-Bit DAC with 0 V to 4.095 Output Swing
4 mA to 20 mA Current Loop Transmitter
Figure 9 shows a self-powered 4 to 20 mA current loop trans-
mitter. The entire circuit floats up from the single-supply (12 V
to 36 V) return. The supply current carries the signal within the
4to 20 mA range. Thus the 4 mA establishes the baseline current
budget with which the circuit must operate. This circuit consumes
only 1.4 mA maximum quiescent current, making 2.6 mA of
current available to power additional signal conditioning circuitry
or to power a bridge circuit.
5V
62
4
REF02
GND
100⍀
220⍀
2N1711
4 TO
20mA
R
L
100⍀
100⍀
1%
1
8
3
24
1/2
OP295/
OP495
100k⍀
1%
220pF
HP
5082-2800
100k⍀
10-TURN
1.21M
1%
182k⍀
1%
10k⍀
10-TURN
V
IN
0 + 3V 12V
TO
36V
SPAN ADJ
NULL ADJ
Figure 9. 4 to 20 mA Current Loop Transmitter
REV. D–10–
OP295/OP495
A 3 V Low-Dropout Linear Voltage Regulator
Figure 10 shows a simple 3 V voltage regulator design. The
regulator can deliver 50 mA load current while allowing a 0.2 V
dropout voltage. The OP295/OP495’s rail-to-rail output swing
handily drives the MJE350 pass transistor without requiring
special drive circuitry. At no load, its output can swing less than
the pass transistor’s base-emitter voltage, turning the device
nearly off. At full load, and at low emitter-collector voltages, the
transistor beta tends to decrease. The additional base current is
easily handled by the OP295/OP495 output.
The amplifier servos the output to a constant voltage, which
feeds a portion of the signal to the error amplifier.
Higher output current, to 100 mA, is achievable at a higher
dropout voltage of 3.8 V.
V
IN
5V TO 3.2V
MJE 350
1/2
OP295/
OP495
43k⍀AD589
1.235V
30.9k⍀
1%
44.2k⍀
1%
3
2
4
1
8
1000pF
I
L
< 50mA
V
O
100␮F
Figure 10. 3 V Low Dropout Voltage Regulator
Figure 11 shows the regulator’s recovery characteristic when its
output underwent a 20 mA to 50 mA step current change.
10
100
0%
90
1ms
20m V
2V
50mA
20mA
OUTPUT
STEP
CURRENT
CONTROL
WAVEFORM
Figure 11. Output Step Load Current Recovery
Low-Dropout, 500 mA Voltage Regulator with Fold-Back
Current Limiting
Adding a second amplifier in the regulation loop, as shown in
Figure 12, provides an output current monitor as well as fold-
back current limiting protection.
Amplifier A1 provides error amplification for the normal voltage
regulation loop. As long as the output current is less than 1 A,
amplifier A2’s output swings to ground, reverse biasing the
diode and effectively taking itself out of the circuit. However, as
the output current exceeds 1 A, the voltage that develops across
the 0.1 Ω sense resistor forces the amplifier A2’s output to go
high, forward-biasing the diode, which in turn closes the current
limit loop. At this point A2’s lower output resistance dominates
the drive to the power MOSFET transistor, thereby effectively
removing the A1 voltage regulation loop from the circuit.
If the output current greater than 1 A persists, the current limit
loop forces a reduction of current to the load, which causes a
corresponding drop in output voltage. As the output voltage
drops, the current limit threshold also drops fractionally, resulting
in a decreasing output current as the output voltage decreases, to
the limit of less than 0.2 A at 1 V output. This “fold-back” effect
reduces the power dissipation considerably during a short circuit
condition, thus making the power supply far more forgiving in
terms of the thermal design requirements. Small heat sinking on
the power MOSFET can be tolerated.
The OP295’s rail-to-rail swing exacts higher gate drive to the
power MOSFET, providing a fuller enhancement to the tran-
sistor. The regulator exhibits 0.2 V dropout at 500 mA of load
current. At 1 A output, the dropout voltage is typically 5.6 V.
IRF9531
SD
G
6V
100k⍀
5%
REF43
2462.500V
124k⍀
1%
124k⍀
1%
205k⍀
1%
45.3k⍀
1%
45.3k⍀
1%
210k⍀
1%
R
SENSE
0.1⍀
1/4W
A2
A1
1
42
3
5
6
7
8
1/2
OP295/
OP495
1/2
OP295/
OP495
1N4148
0.01␮F
I
O
(NORM) = 0.5A
I
O
(MAX) = 1A
5V V
O
Figure 12. Low Dropout, 500 mA Voltage Regulator
with Fold-Back Current Limiting
Square Wave Oscillator
The circuit in Figure 13 is a square wave oscillator (note the
positive feedback). The rail-to-rail swing of the OP295/OP495
helps maintain a constant oscillation frequency even if the sup-
ply voltage varies considerably. Consider a battery-powered
system where the voltages are not regulated and drop over time.
The rail-to-rail swing ensures that the noninverting input sees
the full V+/2, rather than only a fraction of it.
The constant frequency comes from the fact that the 58.7 kΩ
feedback sets up Schmitt trigger threshold levels that are directly
proportional to the supply voltage, as are the RC charge voltage
levels. As a result, the RC charge time, and therefore, the fre-
quency, remains constant independent of supply voltage. The
slew rate of the amplifier limits oscillation frequency to a maxi-
mum of about 800 Hz at a 5 V supply.
Single-Supply Differential Speaker Driver
Connected as a differential speaker driver, the OP295/OP495 can
deliver a minimum of 10 mA to the load. With a 600 Ω load, the
OP295/OP495 can swing close to 5 V p-p across the load.
REV. D
OP295/OP495
–11–
V+
100k⍀58.7k⍀
R
1/2
OP295/
OP495
FREQ OUT
4
1
8
3
2
100k⍀F
OSC
= 1
RC
< 350Hz @ V+ = 5V
C
Figure 13. Square Wave Oscillator Has Stable Frequency
Regardless of Supply Changes
10k⍀
90.9k⍀90.9k⍀
V+
1/4
OP295/
OP495
1/4
OP295/
OP495
1/4
OP295/
OP495
100k⍀
10k⍀
20k⍀20k⍀
VIN
2.2␮F
V+
SPEAKER
Figure 14. Single-Supply Differential Speaker Driver
High Accuracy, Single-Supply, Low Power Comparator
The OP295/OP495 makes an accurate open-loop comparator.
With a single 5 V supply, the offset error is less than 300 µV.
Figure 15 shows the OP295/OP495’s response time when
operating open-loop with 4 mV overdrive. It exhibits a 4 ms
response time at the rising edge and a 1.5 ms response time at
the falling edge.
10
100
0%
90
5ms
2V
1V
OUTPUT
(5mV OVERDRIVE
@ OP-295 INPUT)
INPUT
Figure 15. Open-Loop Comparator Response Time
with 5 mV Overdrive
OP295/OP495 SPICE MODEL Macro-Model
* Node Assignments
*Noninverting Input
*Inverting Input
*Positive Supply
*Negative Supply
*Output
*
*
.SUBCKT OP295 1 2 99 50 20
*
* INPUT STAGE
*
I1 99 4 2E-6
R1 1 6 5E3
R2 2 5 5E3
CIN 1 2 2E-12
IOS 1 2 0.5E-9
D1 5 3 DZ
D2 6 3 DZ
EOS 7 6 POLY (1) (31,39) 30E-6 0.024
Q1 8 5 4 QP
Q2 9 7 4QP
R3 8 50 25.8E3
R4 9 50 25.8E3
*
* GAIN STAGE
*
R7 10 98 270E6
G1 98 10 POLY (1) (9,8) –4.26712E-9 27.8E-6
EREF 98 0 (39, 0) 1
R5 99 39 100E3
R6 39 50 100E3
*
* COMMON MODE STAGE
*
ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5
R12 30 31 1E6
R13 31 98 100
*
* OUTPUT STAGE
*
I2 18 50 1.59E-6
V2 99 12 DC 2.2763
Q4 10 14 50 QNA 1.0
R11 14 50 33
M3 15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10
M4 13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11
D8 10 22 DX
V3 22 50 DC 6
M2 20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
Q5 17 17 99 QPA 1.0
Q6 18 17 99 QPA 4.0
R8 18 99 2.2E6
Q7 18 19 99 QPA 1.0
R9 99 19 8
C2 18 99 20E-12
M6 15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12
M1 20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
D4 21 18 DX
V4 99 21 DC 6
R10 10 11 6E3
C3 11 20 50E-12
.MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3
+ ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4
+ ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4
RC=209
+ CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534
MJC=0.5
+ CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30)
.MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4
+ ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5
+ ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31
RC=354.4
+ CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762
MJC=0.4
+ CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30)
.MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3
+ TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10
VMAX=2E5
+ XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-
4
PB=0.837
+ MJ=0.407 CJSW=0.5E-9 MJSW=0.33)
.MODEL MP PMOS (LEVEL=3 VTO=–1.1 RS=0.7 RD=0.7
+ TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5
+ XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=–1 CJ=1.55E-4
PB=0.56
+ MJ=0.442 CJSW=0.4E-9 MJSW=0.33)
.MODEL DX D(IS=1E-15)
.MODEL DZ D (IS=1E-15, BV=7)
.MODEL QP PNP (BF=125)
.ENDS
REV. D–12–
OP295/OP495
OUTLINE DIMENSIONS
8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
P-Suffix
Dimensions shown in inches and (millimeters)
SEATING
PLANE
0.180
(4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79) 0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
8
14
5
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.100 (2.54)
BSC
0.375 (9.53)
0.365 (9.27)
0.355 (9.02)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-095AA
0.015
(0.38)
MIN
8-Lead Standard Small Outline Package [SOIC]
Narrow Body
(R-8)
S-Suffix
Dimensions shown in millimeters and (inches)
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
0.50 (0.0196)
0.25 (0.0099)
ⴛ 45ⴗ
8ⴗ
0ⴗ
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
85
41
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
6.20 (0.2440)
5.80 (0.2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012AA
16-Lead Standard Small Outline Package [SOIC]
Wide Body
(RW-16)
S-Suffix
Dimensions shown in millimeters and (inches)
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-013AA
SEATING
PLANE
0.30 (0.0118)
0.10 (0.0039)
2.65 (0.1043)
2.35 (0.0925)
1.27 (0.0500)
BSC
16 9
8
1
10.65 (0.4193)
10.00 (0.3937)
7.60 (0.2992)
7.40 (0.2913)
10.50 (0.4134)
10.10 (0.3976)
8ⴗ
0ⴗ
0.75 (0.0295)
0.25 (0.0098)ⴛ 45ⴗ
1.27 (0.0500)
0.40 (0.0157)
COPLANARITY
0.10
0.33 (0.0130)
0.20 (0.0079)
0.51 (0.0201)
0.31 (0.0122)
14-Lead Plastic Dual In-Line Package [PDIP]
(N-14)
P-Suffix
Dimensions shown in inches and (millimeters)
14
17
8
0.685 (17.40)
0.665 (16.89)
0.645 (16.38)
0.295 (7.49)
0.285 (7.24)
0.275 (6.99)
0.100 (2.54)
BSC
SEATING
PLANE
0.180 (4.57)
MAX
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
0.150 (3.81)
0.130 (3.30)
0.110 (2.79) 0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.015 (0.38)
MIN
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-095-AB
REV. D
OP295/OP495
–13–
Revision History
Location Page
2/04—Data Sheet changed from REV. C to REV. D.
Changes to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Changes to Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
3/02—Data Sheet changed from REV. B to REV. C.
Figure changes to PIN CONNECTIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Deletion of OP295GBC and OP495GBC from ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Deletion of WAFER TEST LIMITS table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Changes to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Deletion of DICE CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
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C00331–0–2/04(D)
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