LTC4366ITS8-1#TRMPBF - Linear Technology

LTC4366-1/LTC4366-2

436612fc

TYPICAL APPLICATION

FEATURES DESCRIPTION

High Voltage Surge Stopper

The LTC

4366 surge stopper protects loads from high

voltage transients. By controlling the gate of an external

N-channel MOSFET, the LTC4366 regulates the output

during an overvoltage transient. The load may remain

operational while the overvoltage is dropped across the

MOSFET. Placing a resistor in the return line isolates the

LTC4366 and allows it to float up with the supply; therefore,

the upper limit on the output voltage depends only on the

availability of high valued resistors and MOSFET ratings.

An adjustable overvoltage timer prevents MOSFET dam-

age during the surge while an additional 9 second timer

provides for MOSFET cool down. A shutdown pin reduces

the quiescent current to less than 14µA during shutdown.

After a fault the LTC4366-1 latches off while the LTC4366-2

will auto-retry.

Overvoltage Protected 1.5A, 28V Supply Overvoltage Protector Regulates Output at 43V During Transient

APPLICATIONS

n Rugged Floating Topology

n Wide Operating Voltage Range: 9V to >500V

n Adjustable Output Clamp Voltage

n Controls N-Channel MOSFET

n Adjustable Protection Timer

n Internal 9 Second Cool-Down Timer

n Shutdown IQ < 14µA

n Available in 8-Lead TSOT and 8-Lead 3mm × 2mm

DFN Packages

n Industrial, Automotive and Avionic Surge Protection

n High Voltage DC Distribution

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear

Technology Corporation. All other trademarks are the property of their respective owners.

Patents pending.

VDD OUT

2nF

GATE

IXTK90N25L2

SD FB

1µF

0.47µF

12.4k

VOUT

1.5A

VIN

28V

422k

436612 TA01a

46.4k

10Ω

324k

LTC4366-2

TIMER BASEVSS

28V

100ms/DIV 436612 TA01b

28V

43V CLAMP

250V INPUT SURGE

VIN

100V/DIV

VOUT

20V/DIV

LTC4366-1/LTC4366-2

436612fc

ABSOLUTE MAXIMUM RATINGS

Supply Voltage (VDD) ................................ –0.3V to 10V

Supply Voltage (OUT) ................................. –0.3V to 5V

Input Voltages

FB .............................................. –0.3V to OUT + 0.3V

TIMER ................................................... –0.3V to 3.5V

SD .......................................................... –0.3V to 10V

Output Voltages

BASE .........................................................–1.5V to 4V

OUT-BASE ............................................ –0.3V to 5.5V

GATE (Note 3) ........................................ –0.3V to 15V

GATE-OUT (Note 3) ................................ –0.3V to 10V

(Notes 1, 2) All voltages relative to VSS, unless otherwise noted.

VDD 1

SD 2

TIMER 3

VSS 4

8 GATE

7 OUT

6 FB

5 BASE

TOP VIEW

TS8 PACKAGE

8-LEAD PLASTIC TSOT-23

TJMAX = 125°C, θJA = 195°C/W

TOP VIEW

DDB PACKAGE

8-LEAD (3mm × 2mm) PLASTIC DFN

1VSS

TIMER

VDD

BASE

OUT

GATE

TJMAX = 125°C,

θJA = 75°C/W IF VSS IS SOLDERED TO PCB, θJA = 135°C/W IF VSS IS NOT SOLDERED TO PCB

EXPOSED PAD (PIN 9), PCB VSS CONNECTION OPTIONAL

PIN CONFIGURATION

Currents

IVDD ...................................................................10mA

IOUT ................................................................... 10mA

BASE ................................................. –300µA to 10µA

SD ....................................................... –10mA to 10µA

Operating Temperature Range

LTC4366C ................................................ 0°C to 70°C

LTC4366I ............................................. –40°C to 85°C

LTC4366H .......................................... –40°C to 125°C

LTC4366MP ....................................... –55°C to 125°C

Storage Temperature Range .................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)

TSOT-23 Package Only .................................... 300°C

LTC4366-1/LTC4366-2

436612fc

ORDER INFORMATION

Lead Free Finish

TAPE AND REEL (MINI) TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC4366CTS8-1#TRMPBF LTC4366CTS8-1#TRPBF LTFMC 8-Lead Plastic TSOT-23 0°C to 70°C

LTC4366ITS8-1#TRMPBF LTC4366ITS8-1#TRPBF LTFMC 8-Lead Plastic TSOT-23 –40°C to 85°C

LTC4366HTS8-1#TRMPBF LTC4366HTS8-1#TRPBF LTFMC 8-Lead Plastic TSOT-23 –40°C to 125°C

LTC4366CDDB-1#TRMPBF LTC4366CDDB-1#TRPBF LFMD 8-Lead (3mm × 2mm) Plastic DFN 0°C to 70°C

LTC4366IDDB-1#TRMPBF LTC4366IDDB-1#TRPBF LFMD 8-Lead (3mm × 2mm) Plastic DFN –40°C to 85°C

LTC4366HDDB-1#TRMPBF LTC4366HDDB-1#TRPBF LFMD 8-Lead (3mm × 2mm) Plastic DFN –40°C to 125°C

LTC4366CTS8-2#TRMPBF LTC4366CTS8-2#TRPBF LTFMF 8-Lead Plastic TSOT-23 0°C to 70°C

LTC4366ITS8-2#TRMPBF LTC4366ITS8-2#TRPBF LTFMF 8-Lead Plastic TSOT-23 –40°C to 85°C

LTC4366HTS8-2#TRMPBF LTC4366HTS8-2#TRPBF LTFMF 8-Lead Plastic TSOT-23 –40°C to 125°C

LTC4366CDDB-2#TRMPBF LTC4366CDDB-2#TRPBF LFMG 8-Lead (3mm × 2mm) Plastic DFN 0°C to 70°C

LTC4366IDDB-2#TRMPBF LTC4366IDDB-2#TRPBF LFMG 8-Lead (3mm × 2mm) Plastic DFN –40°C to 85°C

LTC4366HDDB-2#TRMPBF LTC4366HDDB-2#TRPBF LFMG 8-Lead (3mm × 2mm) Plastic DFN –40°C to 125°C

LTC4366MPTS8-1#TRMPBF LTC4366MPTS8-1#TRPBF LTFMC 8-Lead Plastic TSOT-23 –55°C to 125°C

LTC4366MPTS8-2#TRMPBF LTC4366MPTS8-2#TRPBF LTFMF 8-Lead Plastic TSOT-23 –55°C to 125°C

LTC4366MPDDB-1#TRMPBF LTC4366MPDDB-1#TRPBF LFMD 8-Lead (3mm × 2mm) Plastic DFN –55°C to 125°C

LTC4366MPDDB-2#TRMPBF LTC4366MPDDB-2#TRPBF LFMG 8-Lead (3mm × 2mm) Plastic DFN –55°C to 125°C

TRM = 500 pieces. *Temperature grades are identified by a label on the shipping container.

Consult LTC Marketing for parts specified with wider operating temperature ranges.

Consult LTC Marketing for information on lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

LTC4366-1/LTC4366-2

436612fc

ELECTRICAL CHARACTERISTICS

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. All voltages relative to VSS, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

VDD Regulator

VZ(VDD) VDD Shunt Regulator Voltage I = 1mA l11.5 12 12.5 V

∆VZ(VDD) VDD Shunt Regulator Load Regulation I = 1mA to 5mA LTC4366C/I/H

LTC4366MP

130

VDD VDD Supply Voltage (Note 3) l4.5 VZ(VDD) V

IVDD(STLO) VDD Pin Current – Start-Up, Gate Low GATE = 0V, VDD = 7V, OUT = 0V l15 23 µA

IVDD(STHI) VDD Pin Current – Start-Up, Gate High GATE Open, VDD = 7V, OUT = 0V l9 13 µA

IVDD(SD) VDD Pin Current – Shutdown VDD = 7V, OUT = 0V l5 8 µA

OUT Regulator

VZ(OUT) OUT Shunt Regulator Voltage I = 1mA, BASE = 0V l5.0 5.7 6.0 V

∆VZ(OUT) OUT Shunt Regulator Load Regulation I = 1mA to 5mA l30 70 mV

OUT OUT Supply Voltage (Note 3) l3.0 VZ(OUT) V

VUVLO1 OUT Undervoltage Lockout 1 Rising LTC4366C/I/H

LTC4366MP

2.42

2.55

2.75

2.80

∆VUVH1 OUT Undervoltage Lockout 1 Hysteresis l0.2 0.28 0.4 V

VUVLO2 OUT Undervoltage Lockout 2 Rising l4.5 4.75 4.9 V

∆VUVH2 OUT Undervoltage Lockout 2 Hysteresis l0.3 0.4 0.5 V

IOUT(AMP) OUT Pin Current – Regulation Amplifier On l37 54 µA

IOUT(CP) OUT Pin Current – Charge Pump On l150 220 µA

IOUT(SD) OUT Pin Current – Shutdown l3 6 µA

BASE, VSS

VZ(BASE) BASE Shunt Regulator Voltage (OUT-BASE) I = –10µA, OUT = 4.5V l5.5 6.2 6.6 V

∆VZ(BASE) BASE Shunt Regulator Load Regulation I = –10µA to –80µA, OUT = 4.5V l125 200 mV

IBASE BASE Pin Leakage Current OUT = 4.5V, BASE = –0.5V l–0.1 –0.8 –5.5 µA

IVSS(AMP) VSS Pin Current – Regulation Amplifier On l–30 –45 –72 µA

IVSS(CP) VSS Pin Current – Charge Pump On l–108 –160 –230 µA

IVSS(SD) VSS Pin Current – Shutdown l–7 –12 µA

GATE Drive

∆VGATE External N-Channel Gate Drive (GATE-OUT) OUT = 4.9V, I = 0, –1µA l11.2 12 12.5 V

IGATE(ST) GATE Pin Current – Start-Up GATE = OUT = 0V LTC4366C/I/H

LTC4366MP

–4.5

–3.2

–7.5

–11

µA

IGATE(CP) GATE Pin Current – Charge Pump On GATE = 5V, OUT = 4.9V l–14 –20 –28 µA

IGATE(FD) GATE Pin Current – Fast Discharge GATE = 10V, OUT = 4.9V l122 200 300 mA

IGATE(FLT) GATE Pin Current – Fault GATE = 10V, OUT = 4.9V l0.3 0.7 1.2 mA

LTC4366-1/LTC4366-2

436612fc

ELECTRICAL CHARACTERISTICS

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. All voltages relative to VSS, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

FB, SD, TIMER

VFB(REG) 3% FB pin Regulation Threshold (OUT-FB) l1.193 1.23 1.267 V

IFB FB Pin Leakage Current OUT-FB = 1.2V l0 ±1 µA

VSD(TH) SD Pin Threshold Voltage (VDD-SD) Falling l1.0 1.5 2.3 V

VSD(HYST) SD Pin Hysteresis LTC4366C/I/H

LTC4366MP

147

129

280

530

ISD SD Pin Input Pull-Up Current VDD-SD = 0.7V LTC4366C/I/H

LTC4366MP

–0.7

–0.5

–1.6

–3.5

µA

VTIMER(H) TIMER Pin Threshold TIMER Rising, VDD = 7V, OUT = VZ(OUT) l2.6 2.8 3.1 V

ITIMER(UP) TIMER Pin Pull-Up Current TIMER = 1V LTC4366C/I/H

LTC4366MP

–5.1

–4

–9

–13

µA

ITIMER(DN) TIMER Pin Pull-Down Current TIMER = 1V LTC4366C/I/H

LTC4366MP

0.9

0.7

1.8

2.8

µA

ITIMER(RATIO) TIMER Pin Current Ratio ITIMER(DN)/ITIMER(UP) l15 20 25 %

AC Characteristics

tD LY- SD SD Low to Gate Low Filter Time Step VDD-SD from 0V to 3V l420 700 1200 µs

tDLY-FAST FB Low to Gate Low Delay Time Step OUT-FB from 0V to 1.3V l60 150 300 ns

tD(COOL) Cool-Down Timer (Internal) VDD = VZ(VDD) LTC4366C/I/H

LTC4366MP

5.9

Seconds

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: All currents into pins are positive.

Note 3: Limits on the maximum rating is defined as whichever limit occurs

first. An internal clamp limits the GATE pin to a maximum of 12V above

source. Driving this pin to voltages beyond the clamp may damage the

device.

Note 4: TJ is calculated from the ambient temperature, TA, and power

dissipation, PD, according to the formula:

TJ = TA + (PD • 250°C/W)

LTC4366-1/LTC4366-2

436612fc

VSS Current (Regulation AMP On)

vs Temperature

VDD Shunt Regulator vs

VDD Current

VDD Shunt Regulator vs

Temperature

VDD Start-Up Current vs

Temperature (Gate High)

OUT Shunt Regulator vs

OUT Current

OUT Shunt Regulator vs

Temperature

IVDD (mA)

Z(VDD)

(V)

13.0

12.5

12.0

11.5 5

436612 G01

10 15 20

TEMPERATURE (°C)

–50

Z(VDD)

(V)

13.0

11.0 500 10025–25 75

436612 G02

125

12.5

12.0

11.5

TEMPERATURE (°C)

VDD(STHI)

(µA)

436612 G03

–50 500 10025–25 75 125

IOUT (mA)

Z(OUT)

(V)

5.9

5.8

5.7

5.6

5.5

436612 G04

0 5 10 15 20 –50 500 10025–25 75 125

TEMPERATURE (°C)

VZ(OUT) (V)

436612 G05

TEMPERATURE (°C)

VSS(AMP)

(µA)

436612 G10

–50 500 10025–25 75 125

TYPICAL PERFORMANCE CHARACTERISTICS

VSS Current (Charge Pump On)

vs Temperature

Gate Drive vs Gate Pull-Up

Current

Gate Current (Charge Pump On)

vs Temperature

GATE(CP)

(µA)

436612 G14

TEMPERATURE (°C)

–50 500 10025–25 75 125

TEMPERATURE (°C)

VSS(CP)

(µA)

300

200

100

436612 G11

–50 500 10025–25 75 125

IGATE (µA)

∆V

GATE

(V)

0–10 –20

436612 G12

–30

LTC4366-1/LTC4366-2

436612fc

TYPICAL PERFORMANCE CHARACTERISTICS

Gate Start-Up Current vs

Temperature

GATE(ST)

(µA)

436612 G15

TEMPERATURE (°C)

–50 500 10025–25 75 125

Base Shunt Regulator vs

Base Current

Timer Pull-Up Current vs

Temperature

TIMER(UP)

(µA)

–12

–10

–8

–6

–4

436612 G18

TEMPERATURE (°C)

–50 500 10025–25 75 125

IBASE (µA)

Z(BASE)

(V)

7.0

6.5

6.0

5.5 100

436612 G16

200 300 400 500

SD Pull-Up Current vs

Temperature

TEMPERATURE (°C)

(µA)

–3

–2

–1

436612 G21

–50 500 10025–25 75 125

FB Regulation Threshold vs

Temperature Cool-Down Time vs Temperature

FB(REG)

(V)

1.24

1.23

1.22

1.21

436612 G22

TEMPERATURE (°C)

–50 500 10025–25 75 125

TEMPERATURE (°C)

D(COOL)

(s)

436612 G23

–50 500 10025–25 75 125

LTC4366-1/LTC4366-2

436612fc

PIN FUNCTIONS

BASE: Base Driver Output for External PNP Shunt Regula-

tor. This pin is connected to the anode of an internal 6.2V

Zener with the cathode tied to OUT. In cases where lower

Q current is desired but a large VSS resistor is prohibited,

connect an external PNP base to this pin (PNP collector

is grounded, emitter is tied to VSS). Tie this pin to VSS if

unused.

Exposed Pad: The exposed pad may be left open or con-

nected to VSS.

FB: Overvoltage Regulation Amplifier Feedback Input.

Connect this pin to an external resistive divider from OUT

to ground. The overvoltage regulation amplifier controls

the gate of the external N-channel MOSFET to regulate

the FB pin voltage at 1.23V below OUT. The overvoltage

amplifier will activate a 200mA pull-down on the GATE pin

during a fast overvoltage event.

GATE: Gate Drive for External N-Channel MOSFET. Dur-

ing start-up an internal 7.5µA current source charges the

gate of the external N-channel MOSFET from the VDD pin.

Once the OUT voltage is above VSS by 4.75V, the charge

pump will finish charging the GATE to 12V above OUT.

During a fast overvoltage event, a 200mA pull-down cur-

rent source between GATE and OUT is activated, followed

by regulation of the GATE pin voltage by the overvoltage

regulation amplifier.

OUT: Charge Pump and Overvoltage Regulation Amplifier

Supply Voltage. Supply input for floating circuitry powered

from the MOSFET source. Once the OUT voltage is above

4.75V (UVLO2), the charge pump will turn on and draw

power from this pin. When OUT exceeds 2.55V (UVLO1)

it is used as a power supply and reference input for over-

voltage regulation amplifier. This pin is clamped at 5.7V

and requires a 0.22µF or greater bypass to the VSS pin.

SD: Shutdown Comparator Input. Tie to VDD if unused.

Connect pin to a limited current pull down created by adding

a resistor in series with an open-drain or open-collector

pull-down transistor. Activating the external pull down

overcomes the internal 1.6µA pull-up current source and

allows the SD pin to cross the shutdown threshold. This

threshold is defined as 1.5V below VDD with a 280mV

hysteresis. To prevent false triggers this pin must stay

below the threshold for 700µs to activate the shutdown

state. The shutdown state lowers the total quiescent cur-

rent (IVDD plus IOUT) below 20µA. This quiescent current

does not include shunt current in the VDD, OUT and BASE

regulators. After a fault on the LTC4366, putting the part

in shutdown will clear the fault and allow operation to

resume. Clearing the fault during the

second cool-down

period will shorten the time-out for the LTC4366-2 (auto-

retry) version.

TIMER: Timer Input. Leave this pin open for a 1µs overvolt-

age regulation period before fault off. Connect a capacitor

between this pin and VSS to set a 278ms/µF duration for

overvoltage regulation before the switch is turned off.

The LTC4366-2 version will restart after a nine second

cool-down period.

VDD: Start-Up Supply. Supply input for 7.5µA start-up cur-

rent source that charges the gate of the external N-channel

MOSFET. Also provides supply for timer and logic circuits

active when the external MOSFET is off. This pin is clamped

at 12V above VSS. Do not bypass this pin with a capacitor.

VSS: Device Return and Substrate. The capacitors on the

TIMER and OUT pins should be returned to this pin.

LTC4366-1/LTC4366-2

436612fc

SIMPLIFIED DIAGRAM

FUNCTIONAL DIAGRAM

–

RFB1

RFB2

436612 SD

RSS

12V

7.5µA

20µA Z3

5.7V

RIN

1.23V +

–

CHARGE

PUMP

UVLO2

4.75V

OUT

GATE OUTVDD

D17.5µA 20µA

OVERVOLTAGE

AMPLIFIER

LOGIC

SUPPLY VCC

OUT

12V

LOGIC

AND

TIMER

SHUTDOWN

COMPARATOR

–

TIMER

COMPARATOR

UVLO1

2.55V

6.2V

5.7V

12V

2.8V

VCC

VDD

9µA

TIMER

BASE VSS

1.8µA

1.6µA

VOUT

RFB1

RIN

VIN

RFB2

RSS

436612 FD

–

1.5V

–1.23V +

–

Start Run Regulate

LTC4366-1/LTC4366-2

436612fc

OPERATION

The Simplified Diagram shows three states of operation:

the start, run and regulate modes. Previous surge stop-

per parts are powered off the input supply, therefore the

surge voltage is limited to the breakdown voltage of the

input pins of the part. As demonstrated in run mode and

regulate modes, the majority of this part is powered off

the output, so the MOSFET isolates the surge from the

power pins of the part. This allows surge voltages up to

the breakdown of the external MOSFET.

In the start mode a 15µA trickle current flows through RIN,

half is used to charge the gate with the other half used as

bias current. As the GATE pin charges, the external MOSFET

brings up the OUT pin. This leads to the run mode where

the output is high enough to become a supply voltage for

the charge pump. The charge pump is then used to fully

charge the gate 12V above the source.

With the output voltage equal to the input voltage, it is

necessary to protect the load from an input supply over-

voltage. In the regulate mode, the overvoltage regulation

amplifier is referenced to the output through a 1.23V

reference. If the voltage drop across the upper feedback

resistor, RFB1, exceeds 1.23V the regulation amplifier pulls

the gate down to regulate the RFB1 voltage back to 1.23V.

Therefore, the output voltage is clamped by setting the

proper ratio between RFB1 and RFB2.

For example, if the output voltage is regulated at 100V then

the voltage drop across the RFB2 is 98.77V. If the Zener Z3

is 5.7V then the voltage drop across RSS is 94.3V. There-

fore, when the output is at a high voltage, the majority of

the voltage is dropped across the two resistors RFB2 and

RSS. This demonstrates how the LTC4366 floats up with

the supply. The adjustable 3-terminal regulators, such

as the LT

1085 and LM117, are also based on this idea.

The Functional Diagram shows the actual circuits. An

external RIN resistor on the VDD pin powers up the 12V

shunt regulator which then powers up logic supply, VCC.

After verifying that the shutdown input is not active, the

GATE pin is charged with a 7.5µA current from VDD. This

is the start mode.

Once the OUT to VSS voltage exceeds the 2.55V UVLO1

threshold, the overvoltage amplifier is enabled. Next, the

UVLO2 threshold of 4.75V is crossed and the charge pump

turns on. The charge pump charges the GATE pin with

20µA to its final value 12V above OUT (clamped by Z4).

This allows the capacitor between OUT and VSS to charge

until clamped by Z3 to 5.7V. In this run mode the MOSFET

is configured as a low resistance pass transistor with little

voltage drop and power dissipation in the MOSFET.

The powered up LTC4366 is now ready to protect the load

against an overvoltage transient. The overvoltage regula-

tion amplifier monitors the load voltage between OUT and

ground by sensing the voltage on the FB pin with respect

to the OUT pin (drop across RFB1). In an overvoltage

condition the OUT rises until the amplifier drives the M1

gate to regulate and limit the output voltage. This is the

regulate mode.

During regulation the excess voltage is dropped across the

MOSFET. To prevent overheating the MOSFET, the LTC4366

limits the overvoltage regulation time using the TIMER pin.

The TIMER pin is charged with 9µA until the pin exceeds

2.8V. At that point an overvoltage fault is set, the MOSFET

is turned off, and the part enters a cool-down period of

9 seconds. The logic and timer block are active during

cool down while the GATE pin is pulled to OUT.

The latched-off version, LTC4366-1, will remain in fault

until the SD pin is toggled low and then high. Once the fault

is cleared, the GATE is permitted to turn the MOSFET on

again. The auto-retry version, LTC4366-2, waits 9 seconds

then clears the fault and restarts.

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

The typical LTC4366 application is a protected system

that distributes power to loads safe from overvoltage

transients. External component selection is discussed in

the following sections.

Dual Shunt Regulators

The LTC4366 uses two shunt regulators coupled with

the external voltage dropping resistors, RSS and RIN, to

generate internal supply rails at the VDD and OUT pins.

These shunt-regulated rails allow overvoltage protection

from unlimited high voltage transients irrespective of the

voltage rating of the LTC4366’s internal circuitry.

At the beginning of start-up, during shutdown, or after an

overvoltage fault, the GATE pin is clamped to the OUT pin

thereby shutting off the MOSFET. This allows the VSS and

OUT pins to be pulled to ground by output load and RSS.

Under this condition the VDD pin is clamped with a 12V

shunt regulator to VSS. The full supply voltage minus 12V

is then impressed on the RIN resistor which sets the shunt

current. The shunt current can be as high as 10mA which

is several orders of magnitude higher than the typical 9µA

VDD pin quiescent current.

In normal operation the OUT voltage is equal to the input

supply. With C1 fully charged IC1 is zero at this point. Under

this condition the voltage between the OUT and VSS pins

are clamped with a 5.7V shunt regulator. The input supply

voltage minus 5.7V is impressed on RSS. The RSS current

is divided into three areas: the 5.7V shunt current, bias

current between OUT and VSS and finally the RIN current.

The 5.7V shunt current can be as high as 10mA which

greatly exceeds the typical OUT (160µA) bias current.

Turn-On Sequence

The voltage between the VDD and VSS pins is shunt regu-

lated to 12V after ramping up the input supply. Next, the

internally generated supply, VCC, produces a 30µs power-

on-reset pulse which clears the fault latch and initializes

internal latches. Next, the shutdown comparator determines

if the SD pin is externally pulled low, thereby requesting a

low bias current shutdown state. Otherwise the external

MOSFET, M1, is allowed to turn on.

Turning on the 7.5µA GATE pull-up current source from the

VDD pin begins what can be described as a “bootstrapped”

method for powering up the MOSFET gate. Once the GATE

reaches the VDD pin voltage (minus a Schottky diode), the

7.5µA source loses voltage headroom and stops charging

the GATE (middle of waveforms in Figure 2.). The bootstrap

method relies on charging C1 to a sufficient voltage after

GATE stops increasing. The voltage on C1 is then used as

a supply for a charge pump that charges the gate to its

final value 12V above OUT. C1 will discharge if the charge

pump current exceeds the C1 charging current. If the

voltage drops below 4.35V, the charge pump will pause

allowing C1 to recharge.

VDD

470k

100k

OUT

10nF

GATE

FQA62N25C

SD FB

10nF

0.47µF

RFB1

12.4k

VOUT

1.5A

(43V CLAMP)

VIN

28V

(18V DC TO 250V DC)

RFB2

422k

436612 F01

RSS

46.4k

10Ω

RIN

324k

MMBT3904

LTC4366-2

TIMER BASEVSS

Figure 1. Typical Application Figure 2. Turn-On Waveforms

VGATE

10V/DIV

VOUT

10V/DIV

VC1

5V/DIV

20ms/DIV 436612 TA01b

CHARGE

PUMP PAUSE

CHARGE

PUMP STARTS

C1 CHARGING

C1 RECHARGING

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

Starting up with a supply voltage insufficient to charge

C1 with large load current may result in overheating the

MOSFET and subsequent damage. While the gate and

output are ramping the drop across the MOSFET is the

input supply minus the output. If the supply is lower than

necessary to charge C1, then the output fails to ramp higher

than the supply minus the threshold of the MOSFET. This

3V to 5V MOSFET drop with high load current will result in

power dissipation without any protection or timeout limit.

Overvoltage Fault

The LTC4366 prevents an overvoltage on the input supply

from reaching the load. Normally, the pass transistor is

fully on, powering the load with very little voltage drop.

As the input voltage increases the OUT voltage increases

until it reaches the regulation point (VREG). From that

point any further voltage increase is dropped across the

MOSFET. Note the MOSFET is still on so the LTC4366 allows

uninterrupted operation during a short overvoltage event.

The VREG point is configured with the two FB resistors, RFB1

and RFB2. The regulation amplifier compares the FB pin to

a threshold 1.23V below the OUT pin. During regulation

the drop across RFB1 is 1.23V, while the remainder of the

VREG voltage is dropped across RFB2.

When the output is at the regulation point a timer is started

to prevent excessive power dissipation in the MOSFET.

Normally the TIMER pin is held low with a 1.8µA pull-

down current. During regulation the TIMER pin charges

with 9µA. If the regulation point is held long enough for

the TIMER pin to reach 2.8V then an overvoltage fault is

latched. The equation for setting the timer capacitor is:

CT=3.5 •t nF / ms

[ ]

Depending on which version, the part will cool down and

self start (LTC4366-2), or remain latched off until the SD

pin activates a shutdown followed by a start-up command

(LTC4366-1). The cool-down time is typically nine seconds

which provides a very low pulsed power duty cycle.

Starting up with an input supply overvoltage and full

load current does increase the power dissipation in the

MOSFET well beyond the case for an overvoltage surge.

During the gate and output ramp up, the partial supply

voltage (at full current) is dropped across the MOSFET.

After start-up the normal overvoltage surge (with timeout)

occurs before the shutting off the MOSFET. The Design

Example section only considers the normal overvoltage

surge for safe operating area (SOA) calculations for the

MOSFET. Start-up into overvoltage will require additional

SOA considerations.

Shutdown

The LTC4366 has a low current (<20µA) shutdown state

that turns off the pass FET by tying the GATE and OUT pins

together with a switched resistor. In the normal operating

condition, the SD pin is pulled up to the VDD pin voltage

with a 1.6µA current source. Tie the SD pin to VDD when

the shutdown state is not used.

Bringing the SD pin more than 1.5V below VDD pin volt-

age for greater than the 700µs filter time activates the

shutdown state. This filter time prevents unwanted activa-

tion of shutdown during transients. The SD pin is diode

clamped 0.7V below VSS which requires current limiting

(maximum 10mA) on the pull-down device. One way to

limit the current is to connect an external 470k resistor in

series with the open-collector pull-down device. Activat-

ing the external pull-down overcomes the internal 1.6µA

pull-up current source and allows the SD pin to cross the

shutdown threshold.

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

Following an overvoltage fault, putting the part in shutdown

will clear the fault, allowing operation to resume once the

LTC4366 leaves shutdown.

Output Short

A sudden short on the output can result in excessive cur-

rent into the LTC4366 GATE pin supplied from the gate

capacitor, CG. The GATE pin is internally clamped to OUT

with a 10V to 12V clamp. If the OUT pin is pulled low

while the GATE pin is held up with CG, then the clamp will

be damaged trying to discharge CG when clamp voltage

is exceeded. One solution is to add a 1k RS resistor in

series with CG with a bypass diode as shown in Figure 3.

The diode allows the capacitor to function as a bypass for

energy coming from the MOSFET drain to gate capacitor

during an supply overvoltage.

GATE

DBYPASS

436612 F03

LTC4366

OUT

10V TO

12V

Figure 3. Output Short Protection

LTC4366

BASE

436612 F04

RSS

VSS

Figure 4. External PNP Option

Resistor Power Ratings

The proper rating for the RSS resistor in Figure 1 must be

considered. During an overvoltage event the OUT pin is

at regulation voltage (VREG), so the voltage across RSS is

VREG minus 5.7V. A small minimum supply voltage reduces

the value of RSS. Therefore, large differences between

minimum supply voltage and the regulation voltage may

require a large power resistor for RSS.

The full supply voltage minus 12V can appear across

RIN during the overvoltage cool-down period. Normally

the value for RIN is several times larger than RSS which

lowers the power and size requirements for this resistor.

External PNP

In some cases the power resistor for RSS may be physically

large. A large value RSS (with lower power and size) may

be used in conjunction with a PNP as shown in Figure4.

In addition to the 0.8µA sourced from the BASE pin, the

base current from the PNP must flow through RSS which

will limit the maximum RSS value. In some cases the

minimum PNP Beta is as low as 35. The base current

becomes 10µA when the VSS current is 350µA. One can

see this allows a 35 (Beta) times larger RSS than the ap-

plication without the PNP.

Minimum Supply Start-Up

When designing for the minimum supply condition, it is

important that RSS and RIN are chosen to provide enough

current to sufficiently charge C1 to 4.75V. The parameters

that determine the minimum supply voltage include: C1

voltage, MOSFET threshold voltage, a series Schottky

diode voltage drop, resistance of RSS and RIN, current in

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

the VDD pin, and finally the current from the VSS pin (see

Figure 5.).

VIN(MIN) = (IVDD • RIN) + VD + VTH + VC1 + (IVSS • RSS)

Using the Electrical Characteristics table for above

parameters:

VC1 = VUVLO2 = 4.75V (UVLO2 threshold)

IVDD = IVDD(STHI) = 9µA (IVDD start-up, gate high)

IVSS = IVSS(AMP) = 45µA (IVSS w/regulation amp)

VD = 0.58V

VIN(MIN)

= (9µA • RIN) + 0.58V + VTH + 4.75V + (45µA•RSS)

When the MOSFET gate is fully enhanced, the OUT pin

voltage is equal to the supply voltage. This places another

constraint on the minimum supply voltage because the

charge pump increases the VSS current to 160µA. The C1

voltage is assumed to be clamped at 5.7V. These values

are specified as VZ(OUT) and IVSS(CP) (charge pump on)

in the table of Electrical Characteristics:

VIN(MIN) = VZ(OUT) + (IVSS(CP) • RSS)

VIN(MIN) = 5.7V + (160µA • RSS)

The last VIN(MIN) equation sets the maximum value for

RSS. After choosing RSS the maximum value for RIN (for

that particular RSS) is calculated from the first VIN(MIN)

equation:

RSS(MAX) =

IN(MIN)

– 5.7V

160µA

RIN(MAX) =V

IN(MIN) – 4.75V – 0.58V – VTH – 45µA •RSS

( )

9µA

These two equations maximize the values of RSS and

RIN (reducing power dissipation) while still providing

the necessary VC1 voltage to turn the charge pump on.

Increasing the supply voltage beyond the minimum sup-

ply voltage increases the current and power in RSS while

reducing the time required to charge C1. Conditions that

may require an even smaller RSS(MAX) will be discussed

in the Maximum Supply Start-Up section.

Maximum Supply Start-Up

The maximum overvoltage supply may also exist during

start-up. The overvoltage protection circuitry has to wake

up before high voltage is passed to the load. Dynamically

the GATE is ramping up while C1 is charging. Capacitor

C1 must charge to the 2.55V UVLO1 threshold to turn on

the regulation amplifier and reference before the OUT pin

voltage exceeds the overvoltage regulation point, VREG.

These conditions may reduce the value of RSS below the

maximum value dictated by the minimum supply start-up

discussed above.

When current in RSS exceeds the current sourced from

the VSS pin (essentially IRIN), the capacitor C1 begins to

charge. The voltage at the VSS pin when IRIN = IRSS is now

labeled VSS(MATCH). The VSS pin voltage is the center of a

voltage divider between RIN and RSS after the Zener clamp

voltage from VDD to VSS is subtracted from the supply.

VSS(MATCH) =RSS

RSS +RIN

•V

IN(MAX) – VZ(VDD)

( )

As VIN increases the VSS(MATCH) voltage increases. If the

match voltage exceeds the overvoltage regulation point

(VREG), then load is unprotected. This is true because

C1 will still need to charge to 2.55V while VSS already

GATE

7.5µA

VDD

RIN

IRIN

5.7V

12V

IBIAS

ISHUNT2

9µAISHUNT1 IC1

IRSS

VOUT

OUT

RLOAD

436612 F05

RSS

VSS

CIRCUITS

LOGIC

TIMER VC1

–

Figure 5. Simplified Block Diagram

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

has exceeded VREG. Since the OUT pin voltage is at least

2.55V larger than VSS it exceeds the specified maximum.

Choosing the match point (with supply at the maximum)

sufficiently below VREG (by at least 2.55V), allows C1 to

charge up in time to protect the load from overvoltage.

In reality having VSS pin voltage 7V below VREG provides

required margin for charging C1.

VSS(MATCH)(MAX) = VREG – 7V

Increasing RSS increases the match point, so determin-

ing the maximum RSS value while still protecting from

overvoltage is useful. Using IRIN = IRSS:

RSS =RIN •

RSS

RIN

Using:

VRSS = VSS(MATCH)(MAX) = VREG – 7V

VRIN = VIN – VZ(VDD) – VRSS

Substituting:

RSS(MAX) =RIN •VREG – 7V

( )

VIN(MAX) – 12V – VREG – 7V

( )

RSS(MAX) =RIN •VREG – 7V

( )

VIN(MAX) – 5V – VREG

If we guarantee that RSS < RSS(MAX) then the following

is true:

VSS(MATCH) < VSS(MATCH)(MAX)

C1 bypasses the charge pump, and requires at least a

0.22µF. The size of C1 needs limits also. The gate capaci-

tor (CG) dictates the maximum output capacitor C1(MAX)

that will charge to the 2.55V UVLO1 threshold (VUVLO1)

before the OUT voltage exceeds the overvoltage threshold.

(MAX ) =– C G•RSS +RIN

( )

VREG – VSS(MATCH )

( )

IG•RSS •RIN •In 1– 2•VUVLO1

VREG – VSS(MATCH )













In most cases:

C1(MAX) = 10 • CG to 100 • CG

GATE Capacitor, CG

The gate capacitor is used for three functions. First, CG

absorbs charge from the gate-to-drain capacitance of

the MOSFET during overvoltage transients. Second, the

capacitor also acts as a compensation element for the

overvoltage regulation amplifier. The minimum value for

CG to guarantee stability is 2nF. Finally, CG sets the slew

rate of the GATE and OUT pins. The voltage at the GATE pin

rises at a slope equal to 20µA/CG. This slope determines

the charging current into the load capacitor.

IINRUSH =CLOAD

•IG

The voltage rating for CG must be greater than the regula-

tion voltage (VREG).

MOSFET Selection

The LTC4366 drives an N-channel MOSFET to conduct

the load current. The important features of the MOSFET

are on-resistance, RDS(ON), the maximum drain-source

voltage, V(BR)DSS, the threshold voltage, and the SOA.

The maximum allowable drain-source voltage must be

higher than the supply voltage. If the output is shorted

to ground or during an overvoltage event, the full supply

voltage will appear across the MOSFET.

The threshold voltage of the MOSFET is used in the mini-

mum supply start-up calculation. For applications with

supplies less than 12V, a logic-level MOSFET is required.

Above 12V a standard threshold N-channel MOSFET is

sufficient.

The SOA of the MOSFET must encompass all fault condi-

tions. In normal operation the pass transistor is fully on,

dissipating very little power. But during overvoltage faults,

the GATE pin is servoed to regulate the output voltage

through the MOSFET. Large current and high voltage drop

across the MOSFET can coexist in these cases. The SOA

curves of the MOSFET must be considered carefully along

with the selection of the fault timer capacitor.

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

Layout Considerations

Due to the high impedances on the SD, VDD, and GATE

pins, these pins are susceptible to leakages to ground.

For example, a leakage to ground on SD will activate the

shutdown state if greater than 1.6µA. Providing adequate

spacing away from grounded traces and adding conformal

coating on exposed pins lowers the risk that leakage cur-

rent will interrupt system operation.

It is important to put the bypass capacitor, C1, as close as

possible to the OUT and VSS pins. Place the 10Ω resistor

as close as possible to the MOSFET gate pin. This will

limit the parasitic trace capacitance that leads to MOSFET

self-oscillation

The FB pin is sensitive to parasitic capacitance when the

regulation loop is closed. One result from this capacitive

loading is output oscillations during overvoltage regula-

tion. It is suggested that the resistors RFB1 and RFB2 be

placed close to the pin and that the FB trace itself be

minimized in size.

DESIGN EXAMPLE

Overview

The design process starts with minimum input voltage

start-up equations to calculate values for RSS and RIN.

These values need further refinement to meet two other

conditions: the maximum input voltage start-up condi-

tions and proper current for the charging of C1. The the

remaining element values are calculated based on the

input parameters.

Following are the input parameters for this example:

VSUPPLY(MIN) = 18V, VREG = 43V, VIN(MAX) = 250V, ILOAD =

1.5A at start-up, ILOAD = 3A after start-up, VTH = 5V

Important Electrical Characteristics table parameters used

in this example are summarized in Table 1.

Step 1: Maximum RSS

In this design example (Figure 6.) the component sizing

first considers the start-up phase after the charge pump

is active. The goal is to maximize the resistance of RSS

which still allows operation when the input voltage is at

the minimum value.

VDD

470k

100k

OUT

10nF

GATE

FQA62N25C

SD FB

10nF

0.47µF

RFB1

12.4k

VOUT

1.5A

(43V CLAMP)

VIN

28V

(18V DC TO 250V DC)

RFB2

422k

436612 F06

RSS

46.4k

10Ω

RIN

324k

MMBT3904

LTC4366-2

TIMER BASEVSS

Figure 6. Overvoltage Protected 28V, 1.5A Supply

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

After the charge pump is active the VSS current increases

to 160µA (worst-case 230µA, see Table 1) current while

the final value OUT voltage is equal to the minimum supply

voltage. The C1 voltage is clamped at 5.7V (worst-case

6.0V):

RSS(MAX) =V

IN(MIN) – VZ(OUT)

IVSS(CP)

RSS(MAX) =18V – 6V

230µA =52.3k

Step 2: Determine RIN

The value for resistor RIN is calculated using the calculated

RSS value. RIN is chosen to provide enough headroom to

sufficiently charge C1 to 4.9V the maximum undervoltage

lockout 2 threshold (VUVLO2) which starts the charge pump.

The parameters that determine RIN include: minimum

supply voltage, the final C1 voltage, MOSFET threshold

voltage, RSS, 72µA maximum VSS pin current (regulation

amplifier on, IVSS(AMP)), and finally the 13µA maximum

start-up current in the VDD pin (IVDD(STHI)):

RIN(MAX) =VIN(MIN) – VUVLO2 −VD−VTH −ISS(AMP) •RSS

( )

IVDD(STHI)

RIN(MAX) =18V −4.9V −0.58V −5V −72µA •52.3k

( )

13µA

RIN(MAX) =287k

Table 1. Electrical Parameters Used in Design Example

SYMBOL PARAMETER CONDITIONS TYP MAX

VZ(OUT) OUT Shunt Reg. Voltage I = 1mA, BASE = 0V 5.7V 6.0V

VUVLO2 OUT Undervoltage Lockout 2 Rising 4.75V 4.9V

IVSS(CP) VSS Pin Current – Charge Pump On –160µA –230µA

IVSS(AMP) VSS Pin Current – Regulation Amplifier On –45µA –72µA

IVDD(STHI) VDD Pin Current – Start-Up, Gate High GATE Open, VDD = 7V, OUT = 0V 9µA 13µA

IGATE(ST) GATE Pin Current – Start-Up GATE = OUT = 0V –7.5µA –11µA

VUVLO1 OUT Undervoltage Lockout 1 Rising 2.55V 2.75V

Step 3: Find RSS(MAX)

In some cases this value for RSS is too large to charge C1

and power the overvoltage amplifier before the maximum

input voltage passes to the output. The voltage at the VSS

pin when IRIN = IRSS is called the match point (VSS(MATCH)).

Choosing the match point (with supply at the maximum)

sufficiently below VREG (by at least 7V), allows C1 to charge

up in time to protect the load from overvoltage:

RSS(MAX) =RIN •V

REG – 7V

( )

IN(MAX) −5V −V

REG

RSS(MAX) =287k •43V – 7V

( )

250V – 5V – 43V =51.1k

In this case the RSS value of 52.3k calculated in Step 1

is too large.

Step 4: Iterate Smaller RSS

Using 51.1k (RSS(MAX)) as the next guess for RSS, we can

now calculate RIN and RSS(MAX):

RIN =18V – 4.9V −0.58V −5V −72µA •51.1k

( )

13µA

RIN =294k

RSS(MAX)

294k •43V – 7V

( )

250V – 5V −43V =52.3k

In this case the RSS value of 51.1k is less than RSS(MAX)

and the solution is acceptable.

LTC4366-1/LTC4366-2

436612fc

Step 5: Determine CG, C1(MAX), Check RSS

The gate capacitor (CG) determines the gate slew rate and

therefore the slew rate of the OUT pin since the output

voltage follows the GATE pin. The voltage at the GATE pin

rises with a slope equal to 7.5µA/CG at startup and 20µA/CG

when the charge pump is on. Limiting this slope will limit

the inrush current charging the load capacitance where:

IINRUSH =CLOAD

•IG

In this example we choose CG to be 10nF which limits the

inrush current to be 660mA for a 330µF CLOAD.

C1 is used as a bypass capacitor for the circuitry between

the OUT and VSS pins. C1 also stabilizes the shunt regula-

tor that clamps the voltage between these pins where the

minimum value for regulator stability is 0.22µF. An even

greater 0.47µF value is desired for C1 to protect the OUT

to VSS circuitry from transients on the OUT pin.

The startup into an overvoltage creates an upper bound-

ary on the value of C1. The value of CG, RSS and RVIN

determines a maximum C1 that will reach UVLO1 and

power the regulation amplifier before the OUT pin voltage

exceeds the overvoltage threshold. If our desired value

for C1 (0.47µF) exceeds the maximum allowed C1 then

a smaller RSS must be used to iterate a new solution for

C1(MAX). We start with calculating VSS(MATCH):

VSS(MATCH) =RSS

RSS +RVIN

•V

IN – VZ(VDD)

( )

If we use the worst-case 1% maximum value for RSS

(51.6k) and minimum value for RVIN (291k):

VSS(MATCH) = 35.8V

(MAX) =–CG•RSS +RIN

( )

VREG – VSS(MATCH)

( )

IG•RSS •RIN •In 1– 2•VUVLO1

VREG – VSS(MATCH)













Use the worst-case maximum gate current of 11µA instead

of the typical 7.5µA and the worst-case minimum UVLO1

APPLICATIONS INFORMATION

threshold, 2.75V:

(MAX ) =–10nF •51.6k

291k

( )

43V – 35.8V

( )

11µA •51.6k •291k •In 1– 2•2.75V

43V −35.8V











C1(MAX) = 0.1µF

This limit on C1 does not meet the shunt regulator stability

requirements (C1 > 0.22µF).

If we desire a larger value of C1 then a lower size of RSS

is required. A lower value for RSS is 48.7k, which calls

out an RIN value of 309k and a max C1 value of 0.27µF.

The next lower value of 46.4k with RVIN of 324k, results

in the worst-case maximum C1 value of 0.49µF. A larger

C1 increases circuit immunity to transients in exchange

for slightly higher current. Therefore, a selection of com-

ponents that allow a 0.47µF C1 is recommended.

The lowered RSS value of 46.4k now considers the toler-

ances of all the components that set the C1 ramp rate to

guarantee it charges to the 2.55V UVLO1 threshold before

the OUT voltage exceeds the overvoltage threshold.

Step 6: Determine RFB1, RFB2

The feedback resistors, RFB1 and RFB2, are chosen to

regulate the overvoltage at 43V. One way to quickly choose

these resistors is to assign 100µA or 1.2V across a 12.4k

RFB1. RFB2 would need to drop the remainder of the regu-

lated voltage. Dividing this remainder by 100µA yields the

value for RFB2. In this example RFB2 drops 41.8V. When

divided by 100µA it results in a 422k value.

Step 7: Determine CT, R1

During an overvoltage the power dissipated in the MOS-

FET is dependent on the load current and the difference

between the supply and regulated voltages. It is necessary

to keep the device power in a safe range. In the power

MOSFET data sheets there is a maximum safe operating

curve displaying current versus drain to source voltage

for a fixed pulsed time. Other pulsed time data from DC

to 10µs are plotted on the one graph. The different lines

of operation generally follow a constant power squared

LTC4366-1/LTC4366-2

436612fc

APPLICATIONS INFORMATION

VDD

470k

100k

OUT

2nF

GATE

IXTH12N100L

SD FB

3.3nF

0.47µF

RFB1

12.4k

VOUT

0.5A

(200V CLAMP)

VIN

160V (RECTIFIED 110V AC)

100V TO 800V

RFB2

436612 F07

RSS

412k

10Ω

RIN

4.64M

BF722

LTC4366-2

TIMER BASEVSS

DANGER! Lethal Voltages Present

Figure 7. Rectified 110V AC Supply Protected from 220V AC

times time or P2t. Knowing the power we then adjust

the time using the timer capacitor to limit the P2t during

overvoltage. In this example the MOSFET data sheet has

a 6400W2s P2t for a 10ms single pulse.

In this application 250V minus 43V is applied across the

MOSFET at 3A. If the power is applied for less than 16.5ms

then MOSFET P2t limit is not exceeded:

P = (250V – 43V) • 3A = 621W

P2t = (621W)2 • 16.5ms = 6363W2s

Prior to the moment when the output is regulated at 43V,

the output is ramping from 28V to 43V. This ramp time is

based on the 20µA gate current charging the 10nF capaci-

tor. Using the equation for ramp time:

∆t=CG•∆V

=10nF •15V

20µA =7.5ms

To be safe we set the overvoltage time to 10ms. We set the

regulation time to be 2.5ms (the remainder of the 10ms

overvoltage time minus the ramp time). In this example

it is assumed the 250V overvoltage is a constant DC volt-

age for 10ms. This duration exceeds Mil-Std-1275 which

specifies a 70µs surge to 250V that decays in 1.6ms. Us-

ing the following equation (based on charging with 9µA)

to set the CT:

CT=IT•

∆

V=9µA •2.5ms

2.5V ≈10nF

In order to limit the SD pin current (10mA max) a collector

resistor, R1, in series with Q2 is required. The maximum

value for this resistor is around 5M. This requirement oc-

curs when the pull-down is required to sink 1.6µA from

SD and VDD is clamped at 12V. High valued resistors are

susceptible to leakage currents so we chose a 470k resistor

for R1. Resistor R2 provides ESD protection for Q2’s base.

The gate resistor RG limits the parasitic trace capacitance

on Q1’s gate node that could lead to parasitic MOSFET

self-oscillation. The recommended value for RG is 10Ω.

High Voltage Application

In Figure 7 the circuit accepts 110V AC (rectified to 160V)

and protects the load from accidental connection to 220V

AC by limiting the output to less than 200V. The circuit has

a 100V to 800V VIN operating range where the FET break-

down voltage limits the maximum input voltage. The C1

is set to 0.47µF to provide a bypass for the charge pump

that is large enough to provide good noise immunity from

outside voltage transients. The timer capacitor is sized to

give a 1ms overvoltage regulation time that keeps the P2t

below the 640W2s specified for this MOSFET.

LTC4366-1/LTC4366-2

436612fc

28V Vehicle Application

The circuit in Figure 8 adds reverse voltage protection to

the standard 28V application shown in Figure 6. There are

three modes to this circuit: pass FET On when the input

is 18V to 41V, clamping the output to 43V when more

than 43V appears at the input and finally reverse voltage

protection when up to –250V DC is present at the input.

The reverse voltage protection consists of the circuitry

inside the dotted box in Figure 8. When a positive voltage

is first applied to the input, D3 and the forward biased

base-collector junction of Q2 allow the gate of M2 to

follow the input voltage minus a two diode drop. During

this condition the body diode of M2 is used to transmit

power to the LTC4366. Once the LTC4366 is powered up

it fully enhances the gate of M1 and M2 (via D1). The M1

and M2 pass FETs then provide a low impedance path to

the load. In an overvoltage condition, D1 blocks excessive

positive voltage from the input supply passing to the GATE

pin of the LTC4366. D4 eliminates current flow through

R6 when the input is positive while D3 prevents emitter

base breakdown of Q2 when the input is powering up.

During negative input voltages Q2 turns on when current

from R6 (via D4) develops a forward diode drop on R5.

Q2 then holds the gate of M2 at the input voltage which

turns M2 off. This blocks negative input voltages from

reaching M1 and the load. D2 prevents damage to the

LTC4366’s GATE pin by clamping it at ground when the

M2’s gate is negative.

Low Voltage Application

The circuit on the last page (Surge Protected Automotive

Supply) starts up with minimum input voltage of 9V. In

order to successfully start up at 9V and clamp the output

voltage at 18V for input voltages up to 100V the value of

RSS has to be small (1.91k). The FET used in this case has

a 3V threshold to ease the start-up requirements. The timer

capacitor is sized to give a 2.5ms overvoltage regulation

time that keeps the P2t below the 420W2s specified for

this MOSFET.

VDD

470k

270k

100k

OUT

10nF

GATE

FQA62N25C

FDB33N25

REVERSE VOLTAGE

PROTECTION

SD FB

10nF

0.47µF

RFB1

12.4k

VOUT

1.5A

(43V CLAMP)

VIN

18V TO 41V

(±250V DC)

RFB2

422k

436612 F08

RSS

46.4k

10Ω

RIN

324k

MMBT3904

BAV3004W

LTC4366-2

TIMER BASEVSS

470k

MMBT3904

270k

BAV3004W

Figure 8. 28V Vehicle Application with Reverse Voltage Protection

APPLICATIONS INFORMATION

LTC4366-1/LTC4366-2

436612fc

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

1.50 – 1.75

(NOTE 4)

2.80 BSC

0.22 – 0.36

8 PLCS (NOTE 3)

DATUM ‘A’

0.09 – 0.20

(NOTE 3)

TS8 TSOT-23 0710 REV A

2.90 BSC

(NOTE 4)

0.65 BSC

1.95 BSC

0.80 – 0.90

1.00 MAX 0.01 – 0.10

0.20 BSC

0.30 – 0.50 REF

PIN ONE ID

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DRAWING NOT TO SCALE

3. DIMENSIONS ARE INCLUSIVE OF PLATING

4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

5. MOLD FLASH SHALL NOT EXCEED 0.254mm

6. JEDEC PACKAGE REFERENCE IS MO-193

3.85 MAX

0.40

MAX

0.65

REF

RECOMMENDED SOLDER PAD LAYOUT

PER IPC CALCULATOR

1.4 MIN

2.62 REF

1.22 REF

TS8 Package

8-Lead Plastic TSOT-23

(Reference LTC DWG # 05-08-1637 Rev A)

LTC4366-1/LTC4366-2

436612fc

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

2.00 ±0.10

(2 SIDES)

NOTE:

1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE

0.40 ±0.10

BOTTOM VIEW—EXPOSED PAD

0.56 ±0.05

(2 SIDES)

0.75 ±0.05

R = 0.115

TYP

R = 0.05

TYP

2.15 ±0.05

(2 SIDES)

3.00 ±0.10

(2 SIDES)

PIN 1 BAR

TOP MARK

(SEE NOTE 6)

0.200 REF

0 – 0.05

(DDB8) DFN 0905 REV B

0.25 ±0.05

0.50 BSC

PIN 1

R = 0.20 OR

0.25 × 45°

CHAMFER

0.25 ±0.05

2.20 ±0.05

(2 SIDES)

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

0.61 ±0.05

(2 SIDES)

1.15 ±0.05

0.70 ±0.05

2.55 ±0.05

PACKAGE

OUTLINE

0.50 BSC

DDB Package

8-Lead Plastic DFN (3mm × 2mm)

(Reference LTC DWG # 05-08-1702 Rev B)

LTC4366-1/LTC4366-2

436612fc

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER

A 1/12 Added Patents Pending statement

Revised Figure 4 in Applications Information section

B 2/12 Removed reference to overcurrent faults under MOSFET Selection

Fixed orientation of M2 in Figure 8

C 8/12 Updated Shutdown current from <20µA to <14µA

Changed MOSFET part number and Gate Capacitor value used in the Typical Application

Added MP-grade order information and specifications

Added negative sign to graphs G12 x-axis and G18, G21 y-axis

Changed MOSFET part number in Figure 1 and Figure 6

Added section GATE Capacitor, CG

Changed ILOAD current from 5A to 3A in Design Example

Updated C1(MAX) values in Step 5 calculations to 0.27µF and worst case 0.49µF

Updated calculated values in Step 7, added supporting text

Changed MOSFET part number and GATE capacitor used in Figure 7

2, 3, 4, 5

6, 7

11, 16

LTC4366-1/LTC4366-2

436612fc

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

 LINEAR TECHNOLOGY CORPORATION 2011

LT 0812 REV C • PRINTED IN USA

TYPICAL APPLICATION

Surge Protected Automotive 12V Supply

VDD

470k

100k

OUT

2nF

GATE

HUF76639S3S

SD FB

3.3nF

0.47µF

RFB1

12.4k

VOUT

(18V CLAMP)

VIN

12V

(9V TO 100V)

RFB2

169k

436612 TA02

RSS

1.91k

10Ω

RIN

29.4k

MMBT3904

LTC4366-2

TIMER BASEVSS

RELATED PARTS

PART NUMBER DESCRIPTION COMMENTS

LTC1696 Overvoltage Protection Controller ThinSOT™ Package, 2.7V to 28V

LTC2909 Triple/Dual Inputs UV/OV Negative Monitor Pin Selectable Input Polarity Allows Negative and OV Monitoring

LTC2912/LTC2913 Single/Dual UV/OV Voltage Monitor Ads UV and OV Trip Values, ±1.5% Threshold Accuracy

LTC2914 Quad UV/OV Monitor For Positive and Negative Supplies

LTC3827/LTC3827-1 Low IQ, Dual, Synchronous Controller 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, 80µA Quiescent Current

LTC3835/LTC3835-1 Low IQ, Synchronous Step-Down Controller Single Channel LTC3827/LTC3827-1

LT3845 Low IQ, Synchronous Step-Down Controller 4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, 120µA Quiescent Current

LT3850 Dual, 550kHz, 2-Phase Sychronous Step-Down

Controller

Dual 180° Phased Controllers, VIN 4V to 24V, 97% Duty Cycle, 4mm × 4mm

QFN-28, SSOP-28 Packages

LTC3890 Low IQ, Dual 2-Phase, Synchronous Step-Down

Controller

4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V, 50µA Quiescent Current

LT4256 Positive 48V Hot Swap Controller with

Open-Circuit Detect

Foldback Current Limiting, Open-Circuit and Overcurrent Fault Output, Up to 80V

Supply

LTC4260 Positive High Voltage Hot Swap Controller with

8-Bit ADC and I2C

Wide Operating Range 8.5V to 80V

LT4352 Ideal MOSFET ORing Diode External N-Channel MOSFETs Replace ORing Diodes, 0V to 18V

LTC4354 Negative Voltage Diode-OR Controller Controls Two N-Channel MOSFETs, 1µs Turn-Off, 80V Operation

LTC4355 Positive Voltage Diode-OR Controller Controls Two N-Channel MOSFETs, 0.5µs Turn-Off, 80V Operation

LT4363 High Voltage Surge Stopper 100V Overvoltage and Overcurrent Protection, Latch-Off and Auto-Retry Options

LTC4365 Window Passer - OV, UV and Reverse Supply

Protection Controller

2.5V to 34V Operation, Protects 60V to –40V