LTC3851A
1
3851afa
Typical applicaTion
FeaTures
applicaTions
DescripTion
Synchronous
Step-Down Switching
Regulator Controller
The LTC
®
3851A is a high performance synchronous
step-down switching regulator controller that drives
an all N-channel synchronous power MOSFET stage. A
con stant frequency current mode architecture allows a
phase-lockable frequency of up to 750kHz.
OPTI-LOOP compensation allows the transient response
to be optimized over a wide range of output capacitance
and ESR values. The LTC3851A features a precision 0.8V
reference that is compatible with a wide 4V to 38V input
supply range.
The TK/SS pin ramps the output voltage during start-up.
Current foldback limits MOSFET heat dissipation during
short-circuit conditions. The MODE/PLLIN pin selects
among Burst Mode operation, pulse-skipping mode or
continuous inductor current mode at light loads and al-
lows the IC to be synchronized to an external clock. The
LTC3851A contains an improved PLL compared to the
LTC3851.
The LTC3851A-1 is a version with a power good output
signal instead of adjustable current limit.
High Efficiency Synchronous Step-Down Converter
n Wide VIN Range: 4V to 38V Operation
n RSENSE or DCR Current Sensing
n ±1% Output Voltage Accuracy
n Phase-Lockable Fixed Frequency: 250kHz to 750kHz
n Dual N-Channel MOSFET Synchronous Drive
n Very Low Dropout Operation: 99% Duty Cycle
n Adjustable Output Voltage Soft-Start or Tracking
n Output Current Foldback Limiting
n Output Overvoltage Protection
n 5V Internal Regulator
n OPTI-LOOP
®
Compensation Minimizes COUT
n Selectable Continuous, Pulse-Skipping or
Burst Mode
®
Operation at Light Loads
n Low Shutdown IQ: 20µA
n VOUT Range: 0.8V to 5.5V
n Thermally Enhanced 16-Lead MSOP, 16-Lead Narrow
SSOP or 3mm × 3mm QFN Package
n Automotive Systems
n Telecom Systems
n Industrial Equipment
n Distributed DC Power Systems
Efficiency and Power Loss
vs Load Current
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Linear Technology and the Linear logo are registered
trademarks and No RSENSE, UltraFast are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners. Protected by U.S. Patents including
5408150, 5481178, 5705919, 5929620, 6304066, 6498466, 6580258, 6611131.
ILIM
RUN
TK/SS
ITH
FREQ/PLLFLTR
VIN
TG
SW
INTVCC
MODE/PLLIN
BG
SENSE+
SENSE
VFB
GND
0.1µF
0.047µF
3.01k
LTC3851A
22µF
VIN
4.5V TO 36V
VOUT
3.3V
15A
3851A TA01a
0.68µH
30.1k
330pF
2200pF
0.1µF 330µF
×2
BOOST
15k
154k
48.7k
82.5k 0.1µF
4.7µF
LOAD CURRENT (mA)
10
70
EFFICIENCY (%)
POWER LOSS (mW)
75
80
85
90
100 1000 10000 100000
3851A TA01b
65
60
55
50
95
100
1000
100
10
10000
VIN = 12V
VOUT = 3.3V
EFFICIENCY
POWER LOSS
LTC3851A
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absoluTe MaxiMuM raTings
Input Supply Voltage (VIN) ......................... 40V to –0.3V
Topside Driver Voltage (BOOST) ................ 46V to –0.3V
Switch Voltage (SW) ..................................... 40V to –5V
INTVCC, (BOOST – SW), RUN ...................... 6V to –0.3V
TK/SS, ILIM ...........................................INTVCC to –0.3V
SENSE+, SENSE .......................................... 6V to –0.3V
MODE/PLLIN, FREQ/PLLFLTR ..............INTVCC to –0.3V
ITH, VFB Voltages .......................................... 3V to –0.3V
(Note 1)
GN PACKAGE
16-LEAD PLASTIC SSOP NARROW
1
2
3
4
5
6
7
8
TOP VIEW
16
15
14
13
12
11
10
9
MODE/PLLIN
FREQ/PLLFLTR
RUN
TK/SS
ITH
FB
SENSE
SENSE+
SW
TG
BOOST
VIN
INTVCC
BG
GND
ILIM
TJMAX = 125°C, θJA = 110°C/W
1
2
3
4
5
6
7
8
MODE/PLLIN
FREQ/PLLFLTR
RUN
TK/SS
ITH
FB
SENSE
SENSE+
16
15
14
13
12
11
10
9
SW
TG
BOOST
VIN
INTVCC
BG
GND
ILIM
TOP VIEW
MSE PACKAGE
16-LEAD PLASTIC MSOP
17
GND
TJMAX = 125°C, θJA = 35°C/W TO 40°C/W
EXPOSED PAD (PIN 17) IS GND,
MUST BE SOLDERED TO PCB
16 15 14 13
5678
TOP VIEW
17
GND
UD PACKAGE
16-LEAD (3mm × 3mm) PLASTIC QFN
9
10
11
12
4
3
2
1RUN
TK/SS
ITH
FB
BOOST
VIN
INTVCC
BG
FREQ/PLLFLTR
MODE/PLLIN
SW
TG
SENSE
SENSE+
ILIM
GND
TJMAX = 125°C, θJA = 68°C/W, θJC = 4.2°C/W
EXPOSED PAD (PIN 17) IS GND,
MUST BE SOLDERED TO PCB
pin conFiguraTion
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING*PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3851AEGN#PBF LTC3851AEGN#TRPBF 3851A 16-Lead Plastic SSOP 40°C to 125°C
LTC3851AIGN#PBF LTC3851AIGN#TRPBF 3851A 16-Lead Plastic SSOP 40°C to 125°C
LTC3851AEMSE#PBF LTC3851AEMSE#TRPBF 3851A 16-Lead Plastic MSOP 40°C to 125°C
LTC3851AIMSE#PBF LTC3851AIMSE#TRPBF 3851A 16-Lead Plastic MSOP 40°C to 125°C
LTC3851AHMSE#PBF LTC3851AHMSE#TRPBF 3851A 16-Lead Plastic MSOP 40°C to 150°C
LTC3851AMPMSE#PBF LTC3851AMPMSE#TRPBF 3851A 16-Lead Plastic MSOP 55°C to 150°C
LTC3851AEUD#PBF LTC3851AEUD#TRPBF LFPZ 16-Lead (3mm × 3mm) Plastic QFN 40°C to 125°C
LTC3851AIUD#PBF LTC3851AIUD#TRPBF LFPZ 16-Lead (3mm × 3mm) Plastic QFN 40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
INTVCC Peak Output Current ..................................50mA
Operating Junction Temperature Range (Notes 2, 3)
E-Grade, I-Grade ................................ 40°C to 125°C
H-Grade ............................................. 40°C to 150°C
MP-Grade .......................................... –55°C to 150°C
Storage Temperature Range .................. 65°C to 150°C
Lead Temperature (Soldering, 10 sec)
GN/MSE ............................................................300°C
LTC3851A
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elecTrical characTerisTics
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Main Control Loops
VIN Operating Input Voltage Range l4 38 V
VFB Regulated Feedback Voltage ITH = 1.2V (Note 4) 0°C to 85°C
ITH = 1.2V (Note 4) –40°C to 125°C
ITH = 1.2V (Note 4) –40°C to 150°C
ITH = 1.2V (Note 4) –55°C to 150°C
l
l
l
l
0.792
0.788
0.788
0.788
0.800 0.808
0.812
0.812
0.812
V
V
V
V
IFB Feedback Current (Note 4) –10 –50 nA
VREFLNREG Reference Voltage Line Regulation VIN = 6V to 38V (Note 4) 0.002 0.02 %/V
VLOADREG Output Voltage Load Regulation (Note 4) Measured in Servo Loop,
ITH = 1.2V to 0.7V
(Note 4) Measured in Servo Loop,
ITH = 1.2V to 0.7V (H-Grade, MP-Grade)
l
l
0.01
0.1
0.2
%
%
(Note 4) Measured in Servo Loop,
ITH = 1.2V to 1.6V
(Note 4) Measured in Servo Loop,
ITH = 1.2V to 1.6V (H-Grade, MP-Grade)
l
l
0.01
0.1
–0.2
%
%
gmTransconductance Amplifier gmITH = 1.2V, Sink/Source = 5µA (Note 4) 2 mmho
gm GBW Transconductance Amp Gain Bandwidth ITH = 1.2V (Note 8) 3 MHz
IQInput DC Supply Current
Normal Mode
Shutdown
(Note 5)
VRUN = 5V
VRUN = 0V
1.2
25
50
mA
µA
UVLO Undervoltage Lockout on INTVCC VINTVCC Ramping Down 3.25 V
UVLO Hys UVLO Hysteresis 0.4 V
VOVL Feedback Overvoltage Lockout VFB with Respect to Set Regulated Voltage VFB Ramping
Positive (OV)
7.5 10 12.5 %
ISENSE SENSE Pins Current ±1 ±2 µA
ITK/SS Soft-Start Charge Current VTK/SS = 0V 0.6 1 2 µA
VRUN RUN Pin On—Threshold VRUN Rising l1.10 1.22 1.35 V
VRUNHYS RUN Pin On—Hysteresis 120 mV
VSENSE(MAX) Maximum Current Sense Threshold VFB = 0.7V, VSENSE = 3.3V, ILIM = 0V
VFB = 0.7V, VSENSE = 3.3V, ILIM = 0V (H-/MP-Grade)
VFB = 0.7V, VSENSE = 3.3V, ILIM = Float
VFB = 0.7V, VSENSE = 3.3V, ILIM = Float (H-/MP-Grade)
VFB = 0.7V, VSENSE = 3.3V, ILIM = INTVCC
VFB = 0.7V, VSENSE = 3.3V, ILIM = INTVCC (H-/MP-Grade)
l
l
l
l
l
l
20
15
40
35
65
60
30
53
80
40
45
65
70
95
100
mV
mV
mV
mV
mV
mV
TG RUP TG Driver Pull-Up On-Resistance TG High 2.2 Ω
TG RDOWN TG Driver Pull-Down On-Resistance TG Low 1.2 Ω
BG RUP BG Driver Pull-Up On-Resistance BG High 2.1 Ω
BG RDOWN BG Driver Pull-Down On-Resistance BG Low 1.1 Ω
TG tr
TG tf
TG Transition Time
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG tr
BG tf
BG T
ransition Time
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
TG/BG t1D Top Gate Off to Bottom Gate On Delay
Bottom Switch-On Delay Time
CLOAD = 3300pF Each Driver
(Note 6)
30 ns
BG/TG t2D Bottom Gate Off to Top Gate On Delay Top
Switch-On Delay Time
CLOAD = 3300pF Each Driver
(Note 6)
30 ns
tON(MIN) Minimum On-Time (Note 7) 90 ns
The l denotes the specifications which apply over the specified operating
temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 15V, VRUN = 5V unless otherwise noted.
LTC3851A
4
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elecTrical characTerisTics
The l denotes the specifications which apply over the specified operating
temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 15V, VRUN = 5V unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
INTVCC Linear Regulator
VINTVCC Internal VCC Voltage 6V < VIN < 38V 4.8 5 5.2 V
VLDO INT INTVCC Load Regulation ICC = 0mA to 50mA 0.5 2 %
Oscillator and Phase-Locked Loop
fNOM Nominal Frequency RFREQ = 60k 460 500 540 kHz
fLOW Lowest Frequency RFREQ = 160k 205 235 265 kHz
fHIGH Highest Frequency RFREQ = 36k 690 750 810 kHz
RMODE/PLLIN MODE/PLLIN Input Resistance 100
fMODE MODE/PLLIN Minimum Input Frequency
MODE/PLLIN Maximum Input Frequency
VMODE = External Clock
VMODE = External Clock
250
750
kHz
kHz
IFREQ Phase Detector Output Current
Sinking Capability
Sourcing Capability
fMODE > fOSC
fMODE < fOSC
–90
75
µA
µA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may
cause permanent damage to the device. Exposure to any Absolute Maximum
Rating condition for extended periods may affect device reliability and lifetime.
Note 2: The LTC3851A is tested under pulsed load conditions such that
TA ≈ TJ. The LTC3851AE is guaranteed to meet performance specifications
from 0°C to 85°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3851AI is guaranteed to meet specifications over the –40°C to 125°C
operating junction temperature range, the LTC3851AH is guaranteed
over the –40°C to 150°C operating junction temperature range and the
LTC3851AMP is tested and guaranteed over the –55°C to 150°C operating
junction temperature range. High junction temperatures degrade operating
lifetimes; operating lifetime is derated for junction temperatures greater
than 125°C. Note that the maximum ambient temperature consistent with
these specifications is determined by specific operating conditions in
conjunction with board layout, the rated package thermal impedance and
other environmental factors.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC3851AGN: TJ = TA + (PD • 110°C/W)
LTC3851AUD: TJ = TA + (PD • 68°C/W)
LTC3851AMSE: TJ = TA + (PD • 40°C/W)
Note 4: The LTC3851A is tested in a feedback loop that servos VITH to a
specified voltage and measures the resultant VFB.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels. Rise and fall times are assured by
design, characterization and correlation with statistical process controls.
Note 7: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ~40% of IMAX (see Minimum On-Time
Considerations in the Applications Information section).
Note 8: Guaranteed by design; not tested in production.
Typical perForMance characTerisTics
Efficiency vs Output Current
and Mode
Efficiency vs Output Current
and Mode
Efficiency vs Output Current
and Mode
LOAD CURRENT (mA)
10
40
EFFICIENCY(%)
50
60
70
80
100 1000 10000 100000
3851A G01
30
20
10
0
90
100 VIN = 12V
VOUT = 1.5V
BURST
CCM
PULSE
SKIP
LOAD CURRENT (mA)
10
40
EFFICIENCY(%)
50
60
70
80
100 1000 10000 100000
3851A G02
30
20
10
0
90
100
VIN = 12V
VOUT = 3.3V
FIGURE 11 CIRCUIT
BURST
CCM
PULSE
SKIP
LOAD CURRENT (mA)
10
40
EFFICIENCY (%)
50
60
70
80
100 1000 10000 100000
3851A G03
30
20
10
0
90
100
VIN = 12V
VOUT = 5V
BURST
CCM
PULSE
SKIP
LTC3851A
5
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Typical perForMance characTerisTics
Load Step
(Burst Mode Operation)
Load Step
(Forced Continuous Mode)
Load Step
(Pulse-Skipping Mode)
Inductor Current at Light Load
Start-Up with Prebiased Output
at 2V
Coincident Tracking with Master
Supply
Efficiency and Power Loss
vs Input Voltage
ILOAD
5A/DIV
0.2A TO 7.5A
VOUT
100mV/DIV
AC-COUPLED
IL
5A/DIV
100µs/DIVVOUT = 1.5V
VIN = 12V
FIGURE 11 CIRCUIT
3851A G05
ILOAD
5A/DIV
0.2A TO 7.5A
VOUT
100mV/DIV
AC-COUPLED
IL
5A/DIV
100µs/DIVVOUT = 1.5V
VIN = 12V
FIGURE 11 CIRCUIT
3851A G06
ILOAD
5A/DIV
0.2A TO 7.5A
VOUT
100mV/DIV
AC-COUPLED
IL
5A/DIV
100µs/DIVVOUT = 1.5V
VIN = 12V
FIGURE 11 CIRCUIT
3851A G07
FORCED
CONTINOUS
MODE
5A/DIV
PULSE-SKIPPING
MODE
5A/DIV
Burst Mode
OPERATION
5A/DIV
1µs/DIVVOUT = 1.5V
VIN = 12V
ILOAD = 1mA
FIGURE 11 CIRCUIT
3851A G08
VOUT
2V/DIV
VFB
0.5V/DIV
20ms/DIV 3851A G09
TK/SS
0.5V/DIV
10ms/DIV 3851A G10
VOUT
2A LOAD
0.5V/DIV
VMASTER
0.5V/DIV
INPUT VOLTAGE (V)
4
70
EFFICIENCY (%)
POWER LOSS (mW)
75
85
90
95
12 20 24 32
3851A G04
80
8 16 28
100
100
1000
10000
VIN = 12V
VOUT = 3.3V
FIGURE 11 CIRCUIT
POWER LOSS,
IOUT = 5A
POWER LOSS,
IOUT = 0.5A
EFFICIENCY,
IOUT = 0.5A
EFFICIENCY,
IOUT = 5A
LTC3851A
6
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Typical perForMance characTerisTics
Maximum Current Sense Threshold
vs Common Mode Voltage
Maximum Peak Current Sense
Threshold vs ITH Voltage
Burst Mode Peak Current Sense
Threshold vs ITH Voltage
Maximum Current Sense
Threshold vs Duty Cycle
INTVCC Line Regulation
Maximum Current Sense
Threshold vs Feedback Voltage
(Current Foldback)
TK/SS Pull-Up Current
vs Temperature
INPUT VOLTAGE (V)
4 8 12
3.5
INTVCC VOLTAGE (V)
3.7
4.1
4.3
4.5
28 32 36
5.3
3851A G13
3.9
16 20 24 40
4.7
4.9
5.1 ILOAD = 0mA
ILOAD = 25mA
VSENSE COMMON MODE VOLTAGE (V)
0
0
VSENSE THRESHOLD (mV)
3 3.5 4 4.5
90
80
70
60
50
40
30
20
10
3851A G14
0.5 1 1.5 2 2.5 5
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
VITH (V)
0 0.2 0.4 0.6
VSENSE (mV)
0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2
3851A G15
2.4
90
80
70
60
50
40
30
20
10
0
–10
–20
DUTY CYCLE RANGE: 0% TO 100%
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
VITH (V)
VSENSE (mV)
60
50
40
30
20
10
0
0.8 1.2 1.6 2.0
3851A G16
2.40.60.4 1.0 1.4 1.8 2.2
ILIM = FLOAT
BURST COMPARATOR FALLING THESHOLD:
VITH = 0.4V
MAXIMUIM
MINIMUIM
FEEDBACK VOLTAGE (V)
0
MAXIMUM VSENSE (mV)
0.8
3851A G18
0
0.2 0.4 0.6
0.1 0.3 0.5 0.7
90
80
70
60
50
40
30
20
10
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
TEMPERATURE (°C)
–75
0.5
TK/SS CURRENT (µA)
0.6
0.8
0.9
1.0
1.5
1.2
050 75
3851A G19
0.7
1.3
1.4
1.1
–25–50 25 100 125 150
DUTY CYCLE (%)
0
0
CURRENT SENSE THRESHOLD (mV)
10
30
40
50
40 80 100
90
3851A G17
20
20 60
60
70
80 ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
Ratiometric Tracking with Master
Supply
Input DC Supply Current
vs Input Voltage
VMASTER
0.5V/DIV
VOUT
2A LOAD
0.5V/DIV
10ms/DIV 3851A G11
INPUT VOLTAGE (V)
4
SUPPLY CURRENT (mA)
12 20 24 40
3851A G12
8 16 28 32 36
3.0
2.5
2.0
1.5
1.0
0.5
0
LTC3851A
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Regulated Feedback Voltage
vs Temperature
Oscillator Frequency
vs Temperature
Undervoltage Lockout Threshold
(INTVCC) vs Temperature
Oscillator Frequency
vs Input Voltage
Typical perForMance characTerisTics
Shutdown Input DC Supply
Current vs Input Voltage
Shutdown (RUN) Threshold
vs Temperature
Shutdown Input DC Supply
Current vs Temperature
Input DC Supply Current
vs Temperature
INPUT VOLTAGE (V)
5
FREQUENCY (kHz)
415
20
3851A G23
400
390
10 15 25
385
380
420
410
405
395
30 35 40
RFREQ = 80k
INPUT VOLTAGE (V)
0
SHUTDOWN SUPPLY CURRENT (µA)
20
30
40
3851A G25
10
010 20 30
515 25 35
40
15
25
5
35
Maximum Current Sense
Threshold vs INTVCC Voltage
INTVCC VOLTAGE(V)
3.2 3.4 3.6
0
MAXIMUM VSENSE (mV)
10
30
40
50
4.4 4.6 4.8
90
3851A G28
20
3.8 4.0 4.2 5.0
60
70
80 ISET = INTVCC
ISET = FLOAT
ISET = GND
TEMPERATURE (°C)
–75 –50 –25
0.9
RUN PIN VOLTAGE (V)
1.1
1.4
050 75
3851A G20
1.0
1.3
1.2
25 100 150125
RUN FALLING THRESHOLD (OFF)
RUN RISING THRESHOLD (ON)
TEMPERATURE (°C)
–75 –50
REGULATED FEEDBACK VOLTAGE (mV)
802
804
806
25 75
3851A G21
800
798
–25 0 50 100 150125
796
794
TEMPERATURE (°C)
–75 –50
600
700
900
25 75
3851A G22
500
400
–25 0 50 100 150125
300
200
800
FREQUENCY (kHz)
RPLLLPF = 36k
RPLLLPF = 60k
RPLLLPF = 160k
TEMPERATURE (°C)
–75 –50 –25
0
INTVCC VOLTAGE AT UVLO THRESHOLD (V)
2
5
0 50 75
3851A G24
1
4
3
25 100 150125
INTVCC RAMPING UP
INTVCC RAMPING DOWN
TEMPERATURE (°C)
–75 –50
SHUTDOWN INPUT DC SUPPLY CURRENT (µA)
35
25
3851A G26
20
10
–25 0 50
5
0
40
30
25
15
75 100 150125
TEMPERATURE (°C)
–75 –50
INPUT DC SUPPLY CURRENT (mA)
2.0
2.5
3.0
25 75
3851A G27
1.5
1.0
–25 0 50 100 150125
0.5
0
LTC3851A
8
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pin FuncTions
(GN and MSE/UD)
MODE/PLLIN (Pin 1/Pin 15): Forced Continuous Mode,
Burst Mode or Pulse-Skipping Mode Selection Pin and Ex-
ternal Synchronization Input to Phase Detector Pin. Connect
this pin to INTVCC to force continuous conduction mode
of operation. Connect to GND to enable pulse-skipping
mode of operation. To select Burst Mode operation, tie
this pin to INTVCC through a resistor no less than 50k,
but no greater than 250k. A clock on the pin will cause
the controller to operate in forced continuous mode of
operation and synchronize the internal oscillator.
FREQ/PLLFLTR (Pin 2/Pin 16): The phase-locked loop’s
lowpass filter is tied to this pin. Alternatively, a resistor
can be connected between this pin and GND to vary the
frequency of the internal oscillator.
RUN (Pin 3/Pin 1): Run Control Input. A voltage above
1.22V on this pin turns on the IC. However, forcing this
pin below 1.1V causes the IC to shut down the IC. There
is a 2μA pull-up current on this pin.
TK/SS (Pin 4/Pin 2): Output Voltage Tracking and Soft-Start
Input. A capacitor to ground at this pin sets the ramp rate
for the output voltage. An internal soft-start current of of
1μA charges this capacitor.
ITH (Pin 5/Pin 3): Current Control Threshold and Error
Amplifier Compensation Point. The current comparator
tripping threshold increases with its ITH control voltage.
FB (Pin 6/Pin 4): Error Amplifier Feedback Input. This pin
receives the remotely sensed feedback voltage from an
external resistive divider across the output.
SENSE (Pin 7/Pin 5): Current Sense Comparator Inverting
Input. The (–) input to the current comparator is connected
to the output.
SENSE+ (Pin 8/Pin 6): Current Sense Comparator Non-
inverting Input. The (+) input to the current comparator
is normally connected to the DCR sensing network or
current sensing resistor.
ILIM (Pin 9/Pin 7): Current Comparator Sense Voltage
Range Input. Tying this pin to GND, FLOAT or INTVCC
selects the maximum current sense threshold from three
dif ferent levels.
GND (Pin 10/Pin 8, Exposed Pad Pin 17): Ground. All
small-signal components and compensation components
should be Kelvin connected to this ground. The (–) terminal
of CVCC and the (–) terminal of CIN should be closely con-
nected to this pin. The exposed pad should be soldered
to ground for good thermal conductivity.
BG (Pin 11/Pin 9): Bottom Gate Driver Output. This pin
drives the gate of the bottom N-channel MOSFET between
GND and INTVCC.
INTVCC (Pin 12/Pin 10): Internal 5V Regulator Output. The
control circuit is powered from this voltage. Decouple this
pin to GND with a minimum 2.2μF low ESR tantalum or
ceramic capacitor.
VIN (Pin 13/Pin 11): Main Input Supply. Decouple this pin
to GND with a capacitor.
BOOST (Pin 14/Pin 12): Boosted Floating Driver Supply.
The (+) terminal of the boost-strap capacitor is connected
to this pin. This pin swings from a diode voltage drop
below INTVCC up to VIN + INTVCC.
TG (Pin 15/Pin 13): Top Gate Driver Output. This is the
output of a floating driver with a voltage swing equal to
INTVCC superimposed on the switch node voltage.
SW (Pin 16/Pin 14): Switch Node Connection to the In-
ductor. Voltage swing at this pin is from a Schottky diode
(external) voltage drop below ground to VIN.
LTC3851A
9
3851afa
FuncTional DiagraM
+
+
+
+
VIN
2µA
SLOPE COMPENSATION
UVLO
OSC S
RQ
5k
RUN
SWITCH
LOGIC
AND
ANTI-
SHOOT
THROUGH
BG
ON
PULSE SKIP
0.8V
OV
1
100k
1.22V0.64V
ITH
RC
INTVCC
ILIM
ITHB
ICMP
CC1
EA
SS
R1
0.88V
R2
RUN
GND
INTVCC
IREV
SW
TG CB
VIN
CIN
VIN SLEEP
BOOST
BURSTEN
+
OV
CVCC
VOUT
COUT
M2
M1
L1
DB
MODE/PLLIN
100k
SENSE+
SENSE
+
0.8V
REF
TK/SSRUN
0.4V
+
VFB
FREQ/PLLFLTR
PLL-SYNC
5V REG
MODE/SYNC
DETECT
+
+
1µA
CSS
+
3851A FD
+
LTC3851A
10
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operaTion
Main Control Loop
The LTC3851A is a constant frequency, current mode
step-down controller. During normal operation, the top
MOSFET is turned on when the clock sets the RS latch,
and is turned off when the main current comparator, ICMP
,
resets the RS latch. The peak inductor current at which
ICMP resets the RS latch is controlled by the voltage on
the ITH pin, which is the output of the error amplifier, EA.
The VFB pin receives the voltage feedback signal, which is
compared to the internal reference voltage by the EA. When
the load current increases, it causes a slight decrease in
VFB relative to the 0.8V reference, which in turn causes the
ITH voltage to increase until the average inductor current
matches the new load current. After the top MOSFET has
turned off, the bottom MOSFET is turned on until either
the inductor current starts to reverse, as indicated by the
reverse current comparator, IREV, or the beginning of the
next cycle.
INTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. An
internal 5V low dropout linear regulator supplies INTVCC
power from VIN.
The top MOSFET driver is biased from the floating boot-
strap capacitor, CB, which normally recharges during each
off cycle through an external diode when the top MOSFET
turns off. If the input voltage, VIN, decreases to a voltage
close to VOUT, the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detec tor detects this and forces the top MOSFET off for
about 1/10 of the clock period every tenth cycle to allow
CB to recharge. However, it is recommended that there is
always a load present during the drop-out transition to
ensure CB is recharged.
Shutdown and Start-Up (RUN and TK/SS)
The LTC3851A can be shut down using the RUN pin. Pull-
ing this pin below 1.1V disables the controller and most
of the internal circuitry, including the INTVCC regulator.
Releasing the RUN pin allows an internal 2µA current to
pull up the pin and enable that control ler. Alternatively,
the RUN pin may be externally pulled up or driven directly
by logic. Be careful not to exceed the absolute maximum
rating of 6V on this pin.
The start-up of the controllers output voltage, VOUT
, is
controlled by the voltage on the TK/SS pin. When the
voltage on the TK/SS pin is less than the 0.8V internal
reference, the LTC3851A regulates the VFB voltage to the
TK/SS pin voltage instead of the 0.8V reference. This al-
lows the TK/SS pin to be used to program a soft-start by
connecting an external capacitor from the TK/SS pin to
GND. An internal 1µA pull-up current charges this capacitor
creating a voltage ramp on the TK/SS pin. As the TK/SS
voltage rises linearly from 0V to 0.8V (and beyond), the
output voltage VOUT rises smoothly from zero to its final
value. Alternatively, the TK/SS pin can be used to cause
the start-up of VOUT to track another supply. Typically,
this requires connecting to the TK/SS pin an external
resistor divider from the other supply to ground (see the
Applica tions Information section). When the RUN pin
is pulled low to disable the controller, or when INTVCC
drops below its undervoltage lockout threshold of 3.2V,
the TK/SS pin is pulled low by an internal MOSFET. When
in undervoltage lockout, the controller is disabled and the
external MOSFETs are held off.
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping or Continuous Conduction)
The LTC3851A can be enabled to enter high efficiency
Burst Mode operation, constant frequency pulse-skipping
mode or forced continuous conduction mode. To select
forced continuous operation, tie the MODE/PLLIN pin to
INTVCC. To select pulse-skipping mode of operation, float
the MODE/PLLIN pin or tie it to GND. To select Burst Mode
operation, tie MODE/PLLIN to INTVCC through a resistor
no less than 50k, but no greater than 250k.
When the controller is enabled for Burst Mode operation,
the peak current in the inductor is set to approximately
one-forth of the maximum sense voltage even though
the voltage on the ITH pin indicates a lower value. If the
average inductor current is higher than the load current,
LTC3851A
11
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operaTion
the error amplifier, EA, will decrease the voltage on the ITH
pin. When the ITH voltage drops below 0.4V, the internal
sleep signal goes high (enabling sleep mode) and both
external MOSFETs are turned off.
In sleep mode, the load current is supplied by the output
capacitor. As the output voltage decreases, the EAs output
begins to rise. When the output voltage drops enough, the
sleep signal goes low, and the controller resumes normal
operation by turning on the top external MOSFET on the
next cycle of the internal oscillator. When a controller is
enabled for Burst Mode operation, the inductor current is
not allowed to reverse. The reverse current comparator,
IREV
, turns off the bottom external MOSFET just before the
inductor current reaches zero, preventing it from revers-
ing and going negative. Thus, the controller operates in
discontinuous operation. In forced continuous operation,
the inductor current is allowed to reverse at light loads or
under large transient conditions. The peak inductor cur-
rent is determined by the voltage on the ITH pin, just as in
normal operation. In this mode the efficiency at light loads
is lower than in Burst Mode operation. However, continu-
ous mode has the advantages of lower output ripple and
less interference to audio circuitry.
When the MODE/PLLIN pin is connected to GND, the
LTC3851A operates in PWM pulse-skipping mode at light
loads. At very light loads the current comparator, ICMP
, may
remain tripped for several cycles and force the external top
MOSFET to stay off for the same number of cycles (i.e.,
skipping pulses). The inductor current is not allowed to
reverse (discontinuous operation). This mode, like forced
continuous operation, exhibits low output ripple as well as
low audio noise and reduced RF interference as compared
to Burst Mode operation. It provides higher low current
efficiency than forced continuous mode, but not nearly as
high as Burst Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ/PLLFLTR and MODE/PLLIN Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency opera-
tion increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to main tain low output ripple voltage. The switching fre-
quency of the LTC3851A can be selected using the FREQ/
PLLFLTR pin. If the MODE/PLLIN pin is not being driven
by an external clock source, the FREQ/PLLFLTR pin can
be used to program the controllers operating frequency
from 250kHz to 750kHz.
A phase-locked loop (PLL) is available on the LTC3851A
to synchronize the internal oscillator to an external clock
source that is connected to the MODE/PLLIN pin. The
controller operates in forced continuous mode of operation
when it is synchronized. A series RC should be connected
between the FREQ/PLLFLTR pin and GND to serve as the
PLLs loop filter. It is suggested that the external clock be
applied before enabling the controller unless a second
resistor is connected in parallel with the series RC network.
The second resistor prevents very low switching frequency
operation if the controller is enabled before the clock.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (>10%) as well as other more serious con-
ditions that may overvoltage the output. In such cases,
the top MOSFET is turned off and the bottom MOSFET is
turned on until the overvoltage condition is cleared.
LTC3851A
12
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applicaTions inForMaTion
The Typical Application on the first page of this data sheet
is a basic LTC3851A application circuit. The LTC3851A
can be configured to use either DCR (inductor resistance)
sensing or low value resistor sensing. The choice of the
two current sensing schemes is largely a design trade-off
between cost, power consumption and accuracy. DCR
sensing is becoming popular because it saves expensive
current sensing resis tors and is more power efficient,
especially in high current applications. However, current
sensing resistors provide the most accurate current limits
for the controller. Other external component selection
is driven by the load require ment, and begins with the
selection of RSENSE (if RSENSE is used) and the inductor
value. Next, the power MOSFETs and Schottky diodes are
selected. Finally, input and output capacitors are selected.
The circuit shown on the first page can be configured for
operation up to 38V at VIN.
Current Limit Programming
The ILIM pin is a tri-level logic input to set the maximum
current limit of the controller. When ILIM is grounded, the
maximum current limit threshold of the current compara-
tor is programmed to be 30mV. When ILIM is floated, the
maximum current limit threshold is 50mV. When ILIM is
tied to INTVCC, the maximum current limit threshold is
set to 75mV.
SENSE+ and SENSE Pins
The SENSE+ and SENSE pins are the inputs to the current
comparators. The common mode input voltage range of
the current comparators is 0V to 5.5V. Both SENSE pins
are high impedance inputs with small base currents of
less than 1μA. When the SENSE pins ramp up from 0V
to 1.4V, the small base currents flow out of the SENSE
pins. When the SENSE pins ramp down from 5V to 1.1V,
the small base currents flow into the SENSE pins. The
high impedance inputs to the current comparators allow
accurate DCR sensing. However, care must be taken not
to float these pins during normal operation.
Low Value Resistors Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 1. RSENSE is chosen based on the required output
current.
The current comparator has a maximum threshold, VMAX,
determined by the ILIM setting. The current comparator
threshold sets the maximum peak of the inductor current,
yielding a maximum average output current, IMAX, equal to
the maximum peak value less half the peak-to-peak ripple
current, IL. Allowing a margin of 20% for variations in
the IC and external component values yields:
RSENSE =0.8
V
MAX
IMAX + IL/2
Inductor DCR Sensing
For applications requiring the highest possible efficiency,
the LTC3851A is capable of sensing the voltage drop
across the inductor DCR, as shown in Figure 2. The
DCR of the inductor represents the small amount of
DC winding resis tance of the copper, which can be less
than 1mΩ for todays low value, high current inductors.
If the external R1||R2 C1 time constant is chosen to
be exactly equal to the L/DCR time constant, the voltage
drop across the external capacitor is equal to the voltage
drop across the inductor DCR multiplied by R2/(R1 + R2).
Therefore, R2 may be used to scale the voltage across the
sense terminals when the DCR is greater than the target
sense resistance. Check the manufacturers data sheet
for specifications regarding the inductor DCR, in order
to properly dimension the external filter components.
The DCR of the inductor can also be measured using a
good RLC meter.
VIN VIN
INTVCC
BOOST
TG
SW
BG
GND
FILTER COMPONENTS
PLACED NEAR SENSE PINS
SENSE+
SENSE
LTC3851A VOUT
RSENSE
3851A F01
Figure 1. Using a Resistor to Sense Current with the LTC3851A
LTC3851A
13
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applicaTions inForMaTion
VIN VIN
INTVCC
BOOST
TG
SW
BG
GND
INDUCTOR
DCRL
SENSE+
SENSE
LTC3851A
VOUT
3851A F02
R1
R2
*PLACE C1 NEAR SENSE+, SENSE PINS
C1*
R1||R2 • C1 =
RSENSE(EQ) = DCR
L
DCR
R2
R1 + R2
Figure 2. Current Mode Control Using the Inductor DCR
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant fre-
quency architectures by preventing sub-harmonic oscil-
lations at high duty cycles. It is accomplished inter nally
by adding a compensating ramp to the inductor current
signal. Normally, this results in a reduction of maximum
inductor peak cur rent for duty cycles > 40%. However, the
LTC3851A uses a novel scheme that allows the maximum
inductor peak current to remain unaffected throughout all
duty cycles.
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use of
smaller inductor and capacitor values. A higher frequency
generally results in lower efficiency because of MOSFET
gate charge losses. In addition to this basic trade-off, the
effect of inductor value on ripple current and low current
operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current IL decreases with higher
inductance or frequency and increases with higher VIN:
ΔIL=1
fLVOUT 1 VOUT
VIN
Accepting larger values of IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is IL = 0.3(IMAX). The maximum
IL occurs at the maximum input voltage.
The inductor value also has secondary effects. The tran-
sition to Burst Mode operation begins when the average
inductor current required results in a peak current below
≈10% of the current limit determined by RSENSE. Lower
inductor values (higher IL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to increase.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
selected. As inductance increases, core losses go down.
Unfortunately, increased inductance requires more turns
of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates hard, which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode (Optional)
Selection
Two external power MOSFETs must be selected for the
LTC3851A controller: one N-channel MOSFET for the top
(main) switch, and one N-channel MOSFET for the bottom
(synchronous) switch.
LTC3851A
14
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The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is typically 5V during start-up. Consequently,
logic-level threshold MOSFETs must be used in most
applications. The only exception is if low input voltage
is expected (VIN < 5V); then, sub-logic level threshold
MOSFETs (VGS(TH) < 3V) should be used. Pay close atten-
tion to the BVDSS specification for the MOSFETs as well;
most of the logic-level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the on-
resistance, RDS(ON), Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the gate charge curve specified VDS. When the IC is
operating in continuous mode, the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle = VOUT
VIN
Synchronous Switch Duty Cycle = V
IN VOUT
V
IN
The MOSFET power dissipations at maximum output
current are given by:
P
MAIN =VOUT
V
IN
IMAX
( )
21+ δ
( )
RDS(ON) +
V
IN
( )
2IMAX
2
RDR
( )
CMILLER
( )
1
V
INTVCC VTH(MIN)
+1
VTH(MIN)
(f)
P
SYNC =V
IN VOUT
V
IN
IMAX
( )
21+ δ
( )
RDS(ON)
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFETs Miller threshold voltage. VTH(MIN) is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V,
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V, the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
short-circuit when the synchronous switch is on close to
100% of the period.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diode conducts during the dead time
between the conduction of the two power MOSFETs. This
prevents the body diode of the bottom MOSFET from turn-
ing on, storing charge during the dead time and requiring
a reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good size due to the relatively small average current.
Larger diodes result in additional transition losses due to
their larger junction capacitance.
Soft-Start and Tracking
The LTC3851A has the ability to either soft-start by itself
with a capacitor or track the output of another channel
or external supply. When the LTC3851A is configured to
soft-start by itself, a capacitor should be connected to
the TK/SS pin. The LTC3851A is in the shutdown state if
the RUN pin voltage is below 1.10V. TK/SS pin is actively
pulled to ground in this shutdown state.
Once the RUN pin voltage is above 1.22V, the LTC3851A
powers up. A soft-start current of 1μA then starts to charge
its soft-start capacitor. Note that soft-start or tracking is
achieved not by limiting the maximum output current of
the controller but by controlling the output ramp voltage
according to the ramp rate on the TK/SS pin. Current
foldback is disabled during this phase to ensure smooth
soft-start or tracking. The soft-start or tracking range is
applicaTions inForMaTion
LTC3851A
15
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0V to 0.8V on the TK/SS pin. The total soft-start time can
be calculated as:
tSOFT-START =0.8 CSS
1.0µA
Regardless of the mode selected by the MODE/PLLIN pin,
the regulator will always start in pulse-skipping mode up
to TK/SS = 0.64V. Between TK/SS = 0.64V and 0.72V, it
will operate in forced continuous mode and revert to the
selected mode once TK/SS > 0.72V. The output ripple
is minimized during the 80mV forced continuous mode
window.
When the regulator is configured to track another supply,
the feedback voltage of the other supply is duplicated by a
resistor divider and applied to the TK/SS pin. Therefore, the
voltage ramp rate on this pin is determined by the ramp rate
of the other supplys voltage. Note that the small soft-start
capacitor charging current is always flowing, producing
a small offset error. To minimize this error, one can select
the tracking resistive divider value to be small enough to
make this error negligible.
In order to track down another supply after the soft-start
phase expires, the LTC3851A must be configured for
forced continuous operation by connecting MODE/PLLIN
to INTVCC.
Output Voltage Tracking
The LTC3851A allows the user to program how its output
ramps up and down by means of the TK/SS pins. Through
this pin, the output can be set up to either coincidentally or
ratiometrically track with another supplys output, as shown
in Figure 3. In the following discussions, VMASTER refers to
a master supply and VOUT refers to the LTC3851As output
as a slave supply. To implement the coincident tracking
in Figure 3a, connect a resistor divider to VMASTER and
connect its midpoint to the TK/SS pin of the LTC3851A.
The ratio of this divider should be selected the same as
that of the LTC3851As feedback divider as shown in Figure
4a. In this tracking mode, VMASTER must be higher than
VOUT. To implement ratiometric tracking, the ratio of the
resistor divider connected to VMASTER is determined by:
VOUT
V
MASTER
=R2
R4
R3 +R4
R1+R2
So which mode should be programmed? While either
mode in Figure 4 satisfies most practical applications,
the coincident mode offers better output regulation.
This concept can be better understood with the help of
Figure 5. At the input stage of the error amplifier, two
common anode diodes are used to clamp the equivalent
reference voltage and an additional diode is used to match
the shifted common mode voltage. The top two current
sources are of the same amplitude. In the coincident
applicaTions inForMaTion
TIME
(3a) Coincident Tracking
VMASTER
VOUT
OUTPUT VOLTAGE
VMASTER
VOUT
TIME 3851A F03
(3b) Ratiometric Tracking
OUTPUT VOLTAGE
Figure 3. Two Different Modes of Output Voltage Tracking
LTC3851A
16
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applicaTions inForMaTion
mode, the TK/SS voltage is substantially higher than
0.8V at steady state and effectively turns off D1. D2 and
D3 will therefore conduct the same current and offer
tight matching between VFB and the internal precision
0.8V reference. In the ratiometric mode, however, TK/SS
equals 0.8V at steady state. D1 will divert part of the bias
current to make VFB slightly lower than 0.8V.
Although this error is minimized by the exponential I-V
characteristic of the diode, it does impose a finite amount
of output voltage deviation. Furthermore, when the master
supplys output experiences dynamic excursion (under
load transient, for example), the slave channel output will
be affected as well. For better output regulation, use the
coincident tracking mode instead of ratiometric.
INTVCC Regulator
The LTC3851A features a PMOS low dropout linear regula tor
(LDO) that supplies power to INTVCC from the VIN supply.
INTVCC powers the gate drivers and much of the LTC3851A ’s
internal circuitry. The LDO regulates the voltage at the
INTVCC pin to 5V.
The LDO can supply a peak current of 50mA and must
be bypassed to ground with a minimum of 2.2μF ceramic
capacitor or low ESR electrolytic capacitor. No matter
what type of bulk capaci tor is used, an additional 0.1μF
ceramic capacitor placed directly adjacent to the INTVCC
and GND pins is highly recommended. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC3851A to be
exceeded. The INTVCC current, which is dominated by the
gate charge current, is supplied by the 5V LDO.
Power dissipation for the IC in this case is highest and
is approximately equal to VIN IINTVCC. The gate charge
current is dependent on operating frequency as discussed
in the Efficiency Considerations section. The junction tem-
perature can be estimated by using the equa tions given in
Note 3 of the Electrical Characteristics. For example, the
LTC3851A INTVCC current is limited to less than 14mA
from a 36V supply in the GN package:
TJ = 70°C + (14mA)(36V)(110°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode (MODE/PLLIN
= INTVCC) at maximum VIN.
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor, CB, connected to the
BOOST pin supplies the gate drive voltage for the topside
MOSFET. Capacitor CB in the Functional Diagram is charged
though external diode DB from INTVCC when the SW pin
is low. When the topside MOSFET is to be turned on, the
driver places the CB voltage across the gate source of the
MOSFET. This enhances the MOSFET and turns on the
topside switch. The switch node voltage, SW, rises to VIN
R3
VOUT
R4
(4a) Coincident Tracking Setup
TO
VFB
PIN
R3
VMASTER
R4
TO
TK/SS
PIN
R1 R3
VOUT
R4R2
3851A F04
(4b) Ratiometric Tracking Setup
TO
VFB
PIN
TO
TK/SS
PIN
VMASTER
+
I I
D1
TK/SS
0.8V
VFB
D2
D3
3851A F05
EA
Figure 4. Setup for Coincident and Ratiometric Tracking
Figure 5. Equivalent Input Circuit of Error Amplifier
LTC3851A
17
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applicaTions inForMaTion
and the BOOST pin follows. With the topside MOSFET on,
the boost voltage is above the input supply:
VBOOST = VIN + VINTVCC
The value of the boost capacitor CB needs to be 100 times
that of the total input capa citance of the topside MOSFET.
The reverse break down of the external Schottky diode
must be greater than VIN(MAX).
Undervoltage Lockout
The LTC3851A has two functions that help protect the
controller in case of undervoltage conditions. A precision
UVLO comparator constantly monitors the INTVCC voltage
to ensure that an adequate gate-drive voltage is present.
It locks out the switching action when INTVCC is below
3.2V. To prevent oscillation when there is a disturbance
on the INTVCC, the UVLO comparator has 400mV of preci-
sion hysteresis.
Another way to detect an undervoltage condition is to
monitor the VIN supply. Because the RUN pin has a preci-
sion turn-on reference of 1.22V, one can use a resistor
divider to VIN to turn on the IC when VIN is high enough.
CIN Selection
In continuous mode, the source current of the top N-
channel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
IRMS IO(MAX)
VOUT
VIN
VIN
VOUT
1
1/2
This formula has a maximum at VIN = 2VOUT, where
IRMS = IO(MAX)/2. This simple worst-case condition is
com monly used for design because even significant
deviations do not offer much relief. Note that capacitor
manufacturers’ ripple current ratings are often based on
only 2000 hours of life. This makes it advisable to further
derate the capacitor or to choose a capacitor rated at a
higher temperature than required. Several capacitors may
also be paralleled to meet size or height requirements in
the design. Always consult the manufacturer if there is
any question.
COUT Selection
The selection of COUT is primarily determined by the effec-
tive series resistance, ESR, to minimize voltage ripple. The
output ripple, ΔVOUT, in continuous mode is determined by:
ΔVOUT ΔILESR +1
8fCOUT
where f = operating frequency, COUT = output capaci tance
and ΔIL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ΔIL increases
with input voltage. Typically, once the ESR require-
ment for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. With
ΔIL = 0.3IOUT(MAX) and allowing 2/3 of the ripple to be
due to ESR, the output ripple will be less than 50mV at
maximum VIN if the ILIM pin is configured to float and:
COUT Required ESR < 2.2RSENSE
COUT >1
8fRSENSE
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guaran tees
that the output capacitance does not significantly discharge
during the operating frequency period due to ripple current.
The choice of using smaller output capaci tance increases
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage at or below 50mV. The ITH pin
OPTI-LOOP compensation compo nents can be optimized
to provide stable, high perfor mance transient response
regardless of the output capaci tors selected.
The selection of output capacitors for applications with
large load current transients is primarily determined by the
voltage tolerance specifications of the load. The resistive
component of the capacitor, ESR, multiplied by the load
current change, plus any output voltage ripple must be
within the voltage tolerance of the load.
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The required ESR due to a load current step is:
RESR ΔV
ΔI
where I is the change in current from full load to zero load
(or minimum load) and V is the allowed voltage devia-
tion (not including any droop due to finite capacitance).
The amount of capacitance needed is determined by the
maximum energy stored in the inductor. The capacitance
must be sufficient to absorb the change in inductor
current when a high current to low current transition
occurs. The opposite load current transition is generally
determined by the control loop OPTI-LOOP components,
so make sure not to over compensate and slow down
the response. The minimum capacitance to assure the
inductors’ energy is adequately absorbed is:
COUT >L ΔI
( )
2
2 ΔV
( )
VOUT
where
I is the change in load current.
Manufacturers such as Nichicon, United Chemi-Con and
Sanyo can be considered for high performance through-
hole capacitors. The OS-CON semiconductor electrolyte
capacitor available from Sanyo has the lowest (ESR)
(size) product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications, ESR, RMS current han dling
and load step specifications may require multiple capaci-
tors in parallel. Aluminum electrolytic, dry tantalum and
special polymer capacitors are available in surface mount
packages. Special polymer surface mount capaci tors offer
very low ESR but have much lower capacitive density per
unit volume than other capacitor types. These capacitors
offer a very cost-effective output capacitor solution and are
an ideal choice when combined with a controller having
high loop bandwidth. Tantalum capaci tors offer the highest
capacitance density and are often used as output capaci-
tors for switching regulators having controlled soft-start.
Several excellent surge-tested choices are the AVX TPS,
AVX TPSV or the KEMET T510 series of surface mount
tantalums, available in case heights rang ing from 1.5mm
to 4.1mm. Aluminum electrolytic capaci tors can be used
in cost-driven applications, provided that consideration is
given to ripple current ratings, tempera ture and long-term
reliability. A typical application will require several to many
aluminum electrolytic capacitors in parallel. A combina-
tion of the above mentioned capaci tors will often result
in maximizing performance and minimizing overall cost.
Other capacitor types include Nichicon PL series, NEC
Neocap, Panasonic SP and Sprague 595D series. Consult
manufacturers for other specific recommendations.
Like all components, capacitors are not ideal. Each ca-
pacitor has its own benefits and limitations. Combina tions
of different capacitor types have proven to be a very cost
effective solution. Remember also to include high frequency
decoupling capacitors. They should be placed as close as
possible to the power pins of the load. Any inductance
present in the circuit board traces negates their usefulness.
Setting Output Voltage
The LTC3851A output voltage is set by an external feed-
back resistive divider carefully placed across the output,
as shown in Figure 6. The regulated output volt age is
determined by:
ΔIL(SC) =tON(MIN)
V
IN
L
To improve the transient response, a feed-forward ca-
pacitor, CFF
, may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
LTC3851A
VFB
VOUT
RBCFF
RA
3851A F06
Figure 6. Settling Output Voltage
LTC3851A
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Fault Conditions: Current Limit and Current Foldback
The LTC3851A includes current foldback to help limit
load current when the output is shorted to ground. If the
output falls below 40% of its nominal output level, the
maximum sense voltage is progressively lowered from
its maximum programmed value to about 25% of the that
value. Foldback current limiting is disabled during soft-
start or tracking. Under short-circuit conditions with very
low duty cycles, the LTC3851A will begin cycle skipping
in order to limit the short-circuit current. In this situation
the bottom MOSFET will be dissipating most of the power
but less than in normal operation. The short-circuit ripple
current is determined by the minimum on-time tON(MIN)
of the LTC3851A (≈90ns), the input voltage and inductor
value:
VOUT =0.8V 1+RB
RA
The resulting short-circuit current is:
ISC =
1/4MaxV
SENSE
RSENSE
1
2ΔIL(SC)
Programming Switching Frequency
To set the switching frequency of the LTC3851A, connect
a resistor, RFREQ, between FREQ/PLLFLTR and GND. The
relationship between the oscillator frequency and RFREQ
is shown in Figure 7. A 0.1µF bypass capacitor should be
connected in parallel with RFREQ.
Phase-Locked Loop and Frequency Synchronization
The LTC3851A has a phase-locked loop (PLL) comprised
of an internal voltage-controlled oscillator (VCO) and a
phase detector. This allows the turn-on of the top MOSFET
to be locked to the rising edge of an external clock signal
applied to the MODE/PLLIN pin. This phase detector is
an edge sensitive digital type that provides zero degrees
phase shift between the external and internal oscillators.
This type of phase detector does not exhibit false lock to
harmonics of the external clock.
The output of the phase detector is a pair of complemen-
tary current sources that charge or discharge the external
filter network connected to the FREQ/PLLFLTR pin. Note
that the LTC3851A can only be synchronized to an external
clock whose frequency is within range of the LTC3851As
internal VCO.This is guaranteed to be between 250kHz and
750kHz. A simplified block diagram is shown in Figure 8.
If the external clock frequency is greater than the internal
oscillators frequency, fOSC , then current is sunk con-
tinuously from the phase detector output, pulling down
the FREQ/PLLFLTR pin. When the external clock frequency
is less than fOSC, current is sourced continuously, pull-
ing up the FREQ/PLLFLTR pin. If the external and internal
frequencies are the same but exhibit a phase difference,
the current sources turn on for an amount of time corre-
sponding to the phase difference. The voltage on the FREQ/
PLLFLTR pin is adjusted until the phase and frequency of
the internal and external oscillators are identical. At the
stable operating point, the phase detector output is high
impedance and the filter capacitor CLP holds the voltage.
Figure 7. Relationship Between Oscillator Frequency
and Resistor Connected Between FREQ/PLLFLTR and GND
DIGITAL
PHASE/
FREQUENCY
DETECTOR VCO
2.7V RLP
CLP
3851A F08
FREQ/PLLFLTR
EXTERNAL
OSCILLATOR
MODE/
PLLIN
Figure 8. Phase-Locked Loop Block Diagram
RFREQ (k)
20
250
OSCILLATOR FREQUENCY (kHz)
500
550
600
650
300
350
400
450
700
750
40 60 80 100 120 140 160
3851 F07
LTC3851A
20
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applicaTions inForMaTion
The loop filter components, CLP and RLP, smooth out
the current pulses from the phase detector and provide
a stable input to the voltage-controlled oscillator. The
filter components CLP and RLP determine how fast the
loop acquires lock. Typically RLP is 1k to 10k and CLP is
2200pF to 0.01μF.
When the external oscillator is active before the LTC3851A
is enabled, the internal oscillator frequency will track the
external oscillator frequency as described in the preceding
paragraphs. In situations where the LTC3851A is enabled
before the external oscillator is active, a low free-running
oscillator frequency of approximately 50kHz will result. It is
possible to increase the free-running, pre-synchronization
frequency by adding a second resistor, RFREQ, in parallel
with RLP and CLP. RFREQ will also cause a phase difference
between the internal and external oscillator signals. The
magnitude of the phase difference is inversely proportional
to the value of RFREQ. The free-running frequency may be
programmed by using Figure 7 to determine the appropri-
ate value of RFREQ. In order to maintain adequate phase
margin for the PLL, the typical value for CLP is 0.01µF
and for RLP is 1k.
The external clock (on MODE/PLLIN pin) input high
threshold is nominally 1.6V, while the input low thres hold
is nominally 1.2V.
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time dura-
tion that the LTC3851A is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
tON(MIN) <VOUT
V
IN(f)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3851A is approximately
90ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases. This is of particu-
lar concern in forced continuous applications with low
ripple current at light loads. If the duty cycle drops below
the minimum on-time limit in this situation, a significant
amount of cycle skipping can occur with correspondingly
larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percent-
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3851A circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) topside MOSFET
transition losses.
1. The VIN current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET
driver current. VIN current typi cally results in a small
(<0.1%) loss.
2.
INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a cur rent
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
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21
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3. I2R losses are predicted from the DC resistances of
the fuse (if used), MOSFET, inductor and current sense
resistor. In continuous mode, the average output current
flows through L and RSENSE, but is chopped between
the topside MOSFET and the synchronous MOSFET. If
the two MOSFETs have approximately the same RDS(ON),
then the resistance of one MOSFET can simply be
summed with the resistances of L and RSENSE to obtain
I2R losses. For example, if each RDS(ON) = 10mΩ, DCR
= 10mΩ and RSENSE = 5mΩ, then the total resistance
is 25mΩ. This results in losses ranging from 2% to
8% as the output current increases from 3A to 15A for
a 5V output, or a 3% to 12% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7)VIN2 • IO(MAX) • CRSS • f
Other hidden losses such as copper trace and the battery
internal resistance can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these system level losses during the
design phase. The internal battery and fuse resistance
losses can be minimized by making sure that CIN has ad-
equate charge storage and very low ESR at the switch ing
frequency. A 25W supply will typically require a minimum of
20μF to 40μF of capacitance having a maximum of 20mΩ
to 50mΩ of ESR. Other losses including Schottky con-
duction losses during dead time and inductor core losses
generally account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ILOAD (ESR), where ESR is the effective
series resistance of COUT
. ILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed-loop response test point. The DC step,
rise time and settling at this test point truly reflects the
closed-loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining the
rise time at the pin. The ITH external components shown
in the Typical Application circuit will provide an adequate
starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without break-
ing the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This is
why it is better to look at the ITH pin signal which is in the
feedback loop and is the filtered and compensated control
loop response. The midband gain of the loop will be in-
creased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
LTC3851A
22
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stability of the closed-loop system and will demonstrate
the actual overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3851A. These items are also illustrated graphically
in the layout diagram of Figure 9. Check the following in
your layout:
1. Are the board signal and power grounds segregated?
The LTC3851A GND pin should tie to the ground plane
close to the input capacitor(s). The low current or signal
ground lines should make a single point tie directly to
the GND pin. The synchronous MOSFET source pins
should connect to the input capacitor(s) ground.
applicaTions inForMaTion
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
MODE/PLLIN
FREQ/PLLFLTR
RUN
TK/SS
ITH
VFB
SENSE
SENSE+
SW
TG
BOOST
VIN
INTVCC
BG
GND
ILIM
LTC3851A
47pF
CC
CSS
0.1µF
CC2
RFREQ
RC
1000pF
+
COUT
R1
R2
CB
DB
RSENSE
M2
4.7µF
M1
+
CIN
+
L1
VIN
+
VOUT
3851A F09
10Ω
10Ω
+
Figure 9. LTC3851A Layout Diagram
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2. Does the VFB pin connect directly to the feedback resis-
tors? The resistive divider R1, R2 must be connected
between the (+) plate of COUT and signal ground. The
47pF to 100pF capacitor should be as close as pos-
sible to the LTC3851A. Be careful locating the feedback
resistors too far away from the LTC3851A. The VFB line
should not be routed close to any other nodes with high
slew rates.
3. Are the SENSE and SENSE+ leads routed together
with minimum PC trace spacing? The filter capacitor
between SENSE+ and SENSE should be as close as
possible to the LTC3851A. Ensure accurate current
sensing with Kelvin connections as shown in Figure 10.
Series resistance can be added to the SENSE lines to
increase noise rejection and to compensate for the ESL
of RSENSE.
4. Does the (+) terminal of CIN connect to the drain of
the topside MOSFET(s) as closely as possible? This
capacitor provides the AC current to the MOSFET(s).
5. Is the INTVCC decoupling capacitor connected closely
between INTVCC and GND? This capacitor carries the
MOSFET driver peak currents. An addi tional 1μF ceramic
capacitor placed immediately next to the INTVCC and
GND pins can help improve noise performance.
6. Keep the switching node (SW), top gate node (TG) and
boost node (BOOST) away from sensitive small-signal
nodes, especially from the voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and therefore should be kept on
the output side (Pin 9 to Pin 16) of the LTC3851AEGN
and occupy minimum PC trace area.
PC Board Layout Debugging
It is helpful to use a DC-50MHz current probe to monitor
the current in the inductor while testing the circuit. Monitor
the output switching node (SW pin) to synchronize the
oscilloscope to the internal oscillator and probe the actual
output voltage as well. Check for proper performance over
the operating voltage and current range expected in the
application. The frequency of operation should be main-
tained over the input voltage range down to dropout and
until the output load drops below the low current opera-
tion threshold—typically 10% of the maximum designed
cur rent level in Burst Mode operation.
The duty cycle percentage should be maintained from
cycle to cycle in a well designed, low noise PCB imple-
mentation. Variation in the duty cycle at a subharmonic
rate can suggest noise pick-up at the current or voltage
sensing inputs or inadequate loop compensation. Over-
compensation of the loop can be used to tame a poor PC
layout if regulator bandwidth optimization is not required.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher out-
put currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, the Schottky and the
top MOSFET to the sensitive current and voltage sens-
ing traces. In addition, investigate common ground path
voltage pickup between these components and the GND
pin of the IC.
SENSE+SENSE
HIGH CURRENT PATH
3851A F10
CURRENT SENSE
RESISTOR
(RSENSE)
Figure 10. Kelvin Sensing RSENSE
LTC3851A
24
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Design Example
As a design example, assume VIN = 12V (nominal), VIN =
22V (maximum), VOUT = 1.8V, IMAX = 5A, and f = 250kHz.
Refer to Figure 13.
The inductance value is chosen first based on a 30%
ripple current assumption. The highest value of ripple
current occurs at the maximum input voltage. Connect a
160k resistor between the FREQ/PLLFLTR and GND pins,
generating 250kHz op eration. The minimum inductance
for 30% ripple current is:
ΔIL=1
f
( )
L
( )
VOUT 1VOUT
VIN
A 4.7µH inductor will produce 28% ripple current and a
3.3µH will result in 40%. The peak inductor current will be
the maximum DC value plus one-half the ripple current, or
6A, for the 3.3µH value. Increasing the ripple current will
also help ensure that the minimum on-time of 90ns is not
violated. The minimum on-time occurs at maximum VIN:
tON(MIN) =
V
OUT
VIN(MAX) f
( )
=
1.8V
22V 250kHz
( )
=327ns
The RSENSE resistor value can be calculated by connect-
ing ILIM to INTVCC and using the maximum current sense
voltage specification with some accommodation for toler-
ances. Tie ILIM to INTVCC.
RSENSE 75mV
6A
=0.0125Ω, so 0.01Ω is selected
Choosing 1% resistors: R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
applicaTions inForMaTion
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON)