MIC28500 75V/4A Hyper Speed ControlTM Synchronous DC-DC Buck Regulator SuperSwitcher IITM General Description Features The Micrel MIC28500 is an adjustable frequency, synchronous buck regulator featuring a unique adaptive ontime control architecture. The MIC28500 operates over an input supply range of 30V to 75V and provides a regulated output of up to 4A of output current. The output voltage is adjustable down to 0.8V with a guaranteed accuracy of 1%. * Micrel's Hyper Speed ControlTM architecture allows for ultrafast transient response while reducing the output capacitance and also makes (High VIN)/(Low VOUT) operation possible. This adaptive tON ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. The MIC28500 offers a full suite of protection features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, foldback current limit, "hiccup" mode shortcircuit protection and thermal shutdown. All support documentation can be found on Micrel's web site at: www.micrel.com. * * * * * * * * * * * * Hyper Speed ControlTM architecture enables -High Delta V operation (VIN = 75V and VOUT = 0.8V) -Small output capacitance 30V to 75V voltage input Adjustable output down to 0.8V 1% FB accuracy Any CapacitorTM Stable - Zero-ESR to high-ESR output capacitors 4A output current capability, up to 90% efficiency 100kHz to 500kHz switching frequency Internal compensation Foldback current-limit and "hiccup" mode short-circuit protection Thermal shutdown Supports safe startup into a pre-biased load -40C to +125C junction temperature range 28-pin 5mm x 6mm MLF(R) package Applications * Distributed power systems * Communications/networking infrastructure * Set-top box, gateways and routers * Printers, scanners, graphic cards and video cards ___________________________________________________________________________________________________________ Typical Application Efficiency (VIN = 48V) vs. Output Current 100 90 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.8V EFFICIENCY (%) 80 70 60 fSW = 250kHz 50 40 30 20 10 0 1 2 3 4 5 6 OUTPUT CURRENT (A) Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc. MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com June 2011 M9999-060311-B Micrel, Inc. MIC28500 Ordering Information Part Number Voltage Switching Frequency Junction Temperature Range Package Lead Finish MIC28500YJL Adjustable Adjustable -40C to +125C 28-pin 5mm x 6mm MLF(R) Pb-Free Pin Configuration 28-Pin 5mm x 6mm MLF(R) (YJL) Pin Description Pin Number Pin Name Pin Function PVIN High-Side internal N-channel MOSFET Drain Connection (Input): The PVIN operating voltage range is from 30V to 75V. Input capacitors between the PVIN pins and the power ground (PGND) are required and keep the connection short. Enabling the device below 30V VIN and under maximum loading could heat up the device beyond safe operating conditions. 24 EN Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is reduced (typically 0.7mA). Do not pull the EN pin above the VDD supply. Enabling the device below 30V VIN and under maximum loading could heat up the device beyond safe operating conditions. 25 FB Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 26 SGND Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer, see PCB layout guidelines for details. VDD VDD Bias (Input): Power to the internal reference and control sections of the MIC28500. The VDD operating voltage range is from 4.5V to 5.5V. A 2.2F ceramic capacitor from the VDD pin to the PGND pin must be placed next to the IC. VDD must be powered up at the same time or after PVIN to make the soft-start function correctly. PGND Power Ground. PGND is the ground path for the MIC28500 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the signal ground (SGND) loop. 13, 14, 15, 16, 17, 18, 19 27 2, 5, 6, 7, 8, 21 22 June 2011 CS Current Sense (Input): High current output driver return. The CS pin connects directly to the switch node. Due to the high-speed switching on this pin, the CS pin should be routed away from sensitive nodes. CS pin also senses the current by monitoring the voltage across the low-side internal MOSFET during OFF-time. 2 M9999-060311-B Micrel, Inc. MIC28500 Pin Description (Continued) Pin Number Pin Name 20 BST Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1F is connected between the BST pin and the SW pin. 4, 9, 10, 11, 12 SW Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET drain. 23 FS Frequency Setting Pin. 28 PVDD 1, 3 NC June 2011 Pin Function Power Supply for gate driver of bottom MOSFET. No Connect. 3 M9999-060311-B Micrel, Inc. MIC28500 Absolute Maximum Ratings(1, 2) Operating Ratings(3) PVIN to PGND................................................ -0.3V to +76V FS to PGND ....................................................-0.3V to PVIN PVDD, VDD to PGND ......................................... -0.3V to +6V VSW, VCS to PGND .............................. -0.3V to (PVIN +0.3V) VBST to VSW ........................................................ -0.3V to 6V VBST to PGND .................................................. -0.3V to 82V VEN to PGND ...................................... -0.3V to (VDD + 0.3V) VFB to PGND....................................... -0.3V to (VDD + 0.3V) PGND to SGND ........................................... -0.3V to +0.3V Junction Temperature .............................................. +150C Storage Temperature (TS).........................-65C to +150C Lead Temperature (soldering, 10sec)........................ 260C Supply Voltage (PVIN) ........................................ 30V to 75V Bias Voltage (PVDD, VDD).................................. 4.5V to 5.5V Enable Input (VEN) ................................................. 0V to VDD Junction Temperature (TJ) ........................ -40C to +125C Maximum Power Dissipation......................................Note 4 Package Thermal Resistance(4) 5mm x 6mm MLF(R) (JA) ....................................36C/W Electrical Characteristics(5) PVIN = VFS = 48V, VDD = 5V; VBST - VSW = 5V; TA = 25C, unless noted. Bold values indicate -40C TJ +125C. Parameter Condition Min. Typ. Max. Units Power Supply Input Input Voltage Range ( PVIN) 30 75 V FS Voltage Range 2 75 V VDD Bias Voltage Operating Bias Voltage (VDD) Under-Voltage Lockout Trip Level VDD Rising 4.5 5 5.5 V 3.2 3.85 4.45 V UVLO Hysteresis Quiescent Supply Current (IVDD) Shutdown Supply Current (IVDD) 380 mV VFB = 1.5V 1.4 3 mA VDD = VBST = 5.5V, VIN = 48V 0.7 2 mA SW = unconnected, VEN = 0V Reference Feedback Reference Voltage 0C TJ 85C (1.0%) 0.792 0.8 0.808 -40C TJ 125C (1.5%) 0.788 0.8 0.812 V Load Regulation IOUT = 0A to 4A 0.04 % Line Regulation PVIN = 30 to 75V 0.1 % FB Bias Current VFB = 0.8V -0.5 0.005 EN Logic Level High 4.5V < VDD < 5.5V 1.2 0.85 EN Logic Level Low 4.5V < VDD < 5.5V EN Bias Current VEN = 0V 0.5 A Enable Control V 0.78 0.4 V 50 100 A Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 3. The device is not guaranteed to function outside operating range. 4. PD(MAX) = (TJ(MAX) - TA)/ JA, where JA depends upon the printed circuit layout. See "Applications Information." 5. Specification for packaged product only. June 2011 4 M9999-060311-B Micrel, Inc. MIC28500 Electrical Characteristics(5) (Continued) PVIN = VFS = 48V, VDD = 5V; VBST - VSW = 5V; TA = 25C, unless noted. Bold values indicate -40C TJ +125C. Parameter Condition Min. Typ. Max. Units VFS=PVIN 375 500 625 kHz Oscillator Switching Frequency (6) Maximum Duty Cycle (7) Minimum Duty Cycle VFB = 0V, VFS=PVIN 82 % VFB > 0.8V 0 % 360 ns 6 ms Minimum Off-time Soft-Start Soft-Start time Short Circuit Protection Current-Limit Threshold Short-Circuit Current VFB = 0.8V, TJ = 25C 5.5 7 9 A VFB = 0.8V, TJ = 125C 4.2 5.5 8.5 A 2 3.6 5.2 A VFB = 0V Internal FETs Top-MOSFET RDS (ON) ISW = 1A 175 m Bottom-MOSFET RDS (ON) ISW = 1A 31 m SW Leakage Current PVIN = 36V, VSW = 36V, VEN = 0V, VBST = 41V 55 A PVIN Leakage Current PVIN = 36V, VSW = 0V, VEN = 0V, VBST = 41V 55 A Thermal Protection Over-Temperature Shutdown TJ Rising Over-Temperature Shutdown Hysteresis 160 C 25 C Notes: 6. Measured in test mode. 7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns. June 2011 5 M9999-060311-B Micrel, Inc. MIC28500 Typical Characteristics IOUT = 0A 16 VDD = 5V fSW = 250kHz 12 8 4 0 4 VDD = 5V VEN = 0V 3 2 1 35 40 45 50 55 60 65 70 75 30 35 40 INPUT VOLTAGE (V) 8 fSW = 250kHz 6 4 2 45 50 55 60 65 70 30 75 35 40 IOUT = 0A fSW = 250kHz 0.800 0.796 60 65 70 75 VOUT = 3.3V VOUT = 3.3V 0.6% VDD = 5V 0.4% IOUT = 0A to 4A 0.2% fSW = 250kHz CURRENT LIMIT (A) TOTAL REGULATION (%) VDD = 5V 0.8% 55 15 1.0% VOUT = 3.3V 50 Current Limit vs. Input Voltage Total Regulation vs. Input Voltage 0.808 45 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Feedback Voltage vs. Input Voltage 0.804 VOUT = 3.3V VDD = 5V 0 0 30 FEEDBACK VOLTAGE (V) VDD SUPPLY CURRENT (mA) VOUT = 3.3V VDD Operating Supply Current vs. Input Voltage 10 5 SHUTDOWN CURRENT (mA) 20 SUPPLY CURRENT (mA) VIN Shutdown Current vs. Input Voltage VIN Operating Supply Current vs. Input Voltage 0.0% -0.2% -0.4% VDD = 5V 12 fSW = 250kHz 9 6 3 -0.6% -0.8% 0.792 0 -1.0% 30 35 40 45 50 55 60 65 70 75 30 35 40 45 50 INPUT VOLTAGE (V) 30 60 65 70 75 VDD = 5V IOUT = 1A R18=100k R19=100k 100 SHUTDOWN CURRENT (mA) SUPPLY CURRENT (mA) VOUT = 3.3V VOUT = 3.3V 10 VDD = 5V IOUT = 0A 8 fSW = 250kHz 6 4 2 35 40 45 50 55 60 65 70 75 55 60 65 70 75 VIN = 48V IOUT = 0A 0.8 VDD = 5V VEN = 0V 0.6 0.4 0.2 0.0 0 30 50 1.0 VIN = 48V 250 45 VDD Shutdown Current vs. Temperature 12 150 40 INPUT VOLTAGE (V) VDD Operating Supply Current vs. Temperature 300 200 35 INPUT VOLTAGE (V) Switching Frequency vs. Input Voltage SWITCHING FREQUENCY (kHz) 55 -50 -25 0 25 50 75 TEMPERATURE (C) 100 125 -50 -25 0 25 50 75 100 125 TEMPERATURE (C) INPUT VOLTAGE (V) June 2011 6 M9999-060311-B Micrel, Inc. MIC28500 Typical Characteristics (Continued) 4.2 VIN Shutdown Current vs. Temperature 20 3.8 3.7 3.6 3.5 Falling 16 -25 0 25 50 75 100 VDD = 5V IOUT = 0A fSW = 250kHz 12 8 4 3.3 -50 SHUTDOWN CURRENT (mA) Rising 3.4 VOUT = 3.3V SUPPLY CURRENT (mA) 4.0 3.9 5 VIN = 48V VIN = 48V IOUT = 0A 4.1 VDD THRESHOLD (V) VIN Operating Supply Current vs. Temperature VDD UVLO Threshold vs. Temperature V IN = 48V V EN = 0V 2 1 0 -50 TEMPERATURE (C) IOUT = 0A 3 0 125 V DD = 5V 4 -25 0 25 50 75 100 -50 125 -25 0 25 50 75 100 125 TEMPERATURE (C) TEMPERATURE (C) Current Limit vs. Temperature Feedback Voltage vs. Temperature 15 0.808 V OUT = 3.3V 12 V DD = 5V fSW = 250kHz 9 6 3 VIN = 48V VIN = 48V LOAD REGULATION (%) FEEBACK VOLTAGE (V) V IN = 48V CURRENT LIMIT (A) Load Regulation vs. Temperature 0.4% VOUT = 3.3V VDD = 5V 0.804 IOUT = 0A 0.800 0.796 0.3% VOUT = 3.3V 0.2% IOUT = 0A to 4A VDD = 5V fSW = 250kHz 0.1% 0.0% -0.1% -0.2% -0.3% 0 -25 0 25 50 75 100 125 -50 50 75 100 Line Regulation vs. Temperature Switching Frequency vs. Output Current 300 SWITCHING FREQUENCY (kHz) 0.3% VDD = 5V 0.2% IOUT = 0A 0.1% 0.0% -0.1% -0.2% -0.3% 0 25 50 75 TEMPERATURE (C) June 2011 25 TEMPERATURE (C) VIN = 30V to 75V VOUT = 3.3V -25 0 TEMPERATURE (C) 0.4% -50 -25 100 125 -25 125 0 25 50 75 100 125 TEMPERATURE (C) EN Bias Current vs. Temperature 100 VIN = 48V -40C EN BIAS CURRENT (A) -50 LINE REGULATION (%) -50 0.792 250 25C 125C 200 VIN = 48V VOUT = 3.3V VDD = 5V 150 VDD = 5V 80 VEN = 0V 60 40 20 0 -50 100 0 1 2 3 OUTPUT CURRENT (A) 7 4 -25 0 25 50 75 100 125 TEMPERATURE (C) M9999-060311-B Micrel, Inc. MIC28500 Typical Characteristics (Continued) Efficiency vs. Output Current Enable Threshold vs. Temperature 90 VDD = 5V 85 EFFICIENCY (%) 0.9 Rising 0.8 30VIN FEEDBACK VOLTAGE (V) VIN = 48V ENABLE THRESHOLD (V) 0.808 95 1.0 0.7 Falling 80 48VIN 75 75VIN VOUT = 3.3V 70 VDD = 5V 65 fSW = 250kHz 60 0.6 Feedback Voltage vs. Output Current VIN = 48V VOUT = 3.3V VDD = 5V 0.804 fSW = 250kHz 0.800 0.796 55 0.792 50 0.5 0 -50 -25 0 25 50 75 100 125 1 2 3 OUTPUT CURRENT (A) TEMPERATURE (C) 4 0 1 2 3 4 OUTPUT CURRENT (A) Line Regulation vs. Output Current 0.3% LINE REGULATION (%) VIN = 30V to 75V 0.2% VOUT = 3.3V 0.1% fSW = 250kHz VDD = 5V 0.0% -0.1% -0.2% -0.3% 0 1 2 3 4 OUTPUT CURRENT (A) June 2011 8 M9999-060311-B Micrel, Inc. MIC28500 Typical Characteristics (Continued) 80 DIE TEMPERATURE (C) DIE TEMPERATURE (C) 80 60 40 VIN = 30V VOUT = 3.3V 20 Die Temperature* (VIN = 48V) vs. Output Current VDD = 5V fSW = 250kHz 60 40 VIN = 48V VOUT = 3.3V 20 Die Temperature* (VIN = 75V) vs. Output Current 100 DIE TEMPERATURE (C) Die Temperature* (VIN = 30V) vs. Output Current 80 60 40 VIN = 75V VOUT = 3.3V 20 VDD = 5V VDD = 5V fSW = 250kHz fSW = 250kHz 0 0 1 2 3 4 0 0 0 OUTPUT CURRENT (A) 1 2 3 0 4 1 OUTPUT CURRENT (A) 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 70 60 VIN = 30V fSW = 250kHz 50 70 60 50 VIN = 48V 40 40 90 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 80 EFFICIENCY (%) 80 fSW = 250kHz 0 1 2 3 4 5 50 20 0 1 2 3 4 5 10 6 0 VOUT = 2.5V 1 Maximum Ambient Temperature (C) 100 5 6 VIN = 48V fSW =250kHz L=10uH Tj_MAX =125C 3 VOUT = 1.2V 2 VOUT = 0.8V 1 0 0 85 4 4 Load Current (A) Load Current (A) VOUT = 5V 2 70 3 VIN = 48V fSW =250kHz L=10uH Tj_MAX =125C 55 2 Thermal Derating 4 40 1 OUTPUT CURRENT (A) Thermal Derating VIN = 48V VOUT = 3.3V fSW = 250kHz OUTPUT CURRENT (A) 4 3 VIN = 75V 40 20 Thermal Derating Load Current (A) 60 30 OUTPUT CURRENT (A) 25 70 30 6 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 80 10 30 4 100 100 90 90 3 Efficiency (VIN = 75V) vs. Output Current EFFICIENCY (%) 100 EFFICIENCY (%) Efficiency (VIN = 48V) vs. Output Current Efficiency (VIN = 30V) vs. Output Current 2 OUTPUT CURRENT (A) 25 40 55 70 85 Maximum Ambient Temperature (C) 100 VOUT = 12V 3 fSW=250kHz L=33H Tj_MAX =125C 2 1 0 25 40 55 70 85 100 Maximum Ambient Temperature (C) Die Temperature* : The temperature measurement was taken at the hottest point on the MIC28500 case mounted on a 5 square inch 4 layer, 0.62", FR-4 PCB with 2oz. finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. June 2011 9 M9999-060311-B Micrel, Inc. MIC28500 Functional Characteristics June 2011 10 M9999-060311-B Micrel, Inc. MIC28500 Functional Characteristics (Continued) June 2011 11 M9999-060311-B Micrel, Inc. MIC28500 Functional Characteristics (Continued) June 2011 12 M9999-060311-B Micrel, Inc. MIC28500 Functional Diagram Figure 1. MIC28500 Block Diagram June 2011 13 M9999-060311-B Micrel, Inc. MIC28500 where tS = 1/fSW. It is not recommended to use MIC28500 with a OFF-time close to tOFF(min) during steady-state operation.. The actual ON-time and resulting switching frequency will vary with the part-to-part variation in the rise and fall times of the internal MOSFETs, the output load current, and variations in the VDD voltage. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications, such as 75V to 1.0V. The minimum tON measured on the MIC28500 evaluation board is about 184ns. During load transients, the switching frequency is changed due to the varying OFF-time. Figure 2 shows the MIC28500 control loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. Functional Description The MIC28500 is an adaptive ON-time synchronous step-down DC-DC regulator. It is designed to operate over a wide input voltage range from, 30V to 75V, and provides a regulated output voltage at up to 4A of output current. A digitally modified adaptive ON-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. Over current protection is implemented without the use of an external sense resistor. The device includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Theory of Operation Figure 1 illustrates the block diagram for the control loop of the MIC28500. The output voltage is sensed by the MIC28500 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low gain transconductance (gm) amplifier. If the feedback voltage decreases and the output of the gm amplifier is below 0.8V, then the error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the "FIXED tON ESTIMATION" circuitry: t ON(estimated) = VOUT VIN x fSW Eq. 1 where VOUT is the output voltage and VIN is the power stage input voltage and fSW is the switching frequency. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 360ns, then the MIC28500 control logic will apply the tOFF(min) instead. The minimum tOFF(min) period is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. The maximum duty cycle is obtained from the 360ns tOFF(min): Dmax = June 2011 t S - t OFF(min) tS = 1- 360ns tS Figure 2. MIC28500 Control Loop Timing Eq. 2 14 M9999-060311-B Micrel, Inc. MIC28500 Figure 3 shows the operation of the MIC28500 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This will cause the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time tOFF(min) is generated to charge CBST since the feedback voltage is still below VREF. Then, the next ON-time period is triggered due to the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC28500 converter. Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC28500 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with 9.7mV steps. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function correctly. Current Limit The MIC28500 uses the RDS(ON) of the internal low-side power MOSFET to sense over-current conditions. This method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. In each switching cycle of the MIC28500 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. If the peak inductor current is greater than 7A, then the MIC28500 turns off the highside MOSFET and a soft-start sequence is triggered. This mode of operation is called "hiccup mode" and its purpose is to protect the downstream load in case of a hard short. The current-limit threshold has a foldback characteristic related to the feedback voltage, as shown in Figure 4. Figure 3. MIC28500 Load Transient Response Unlike true current-mode control, the MIC28500 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. In order to meet the stability requirements, the MIC28500 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV. If a low-ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to "Ripple Injection" subsection in Application Information for more details about the ripple injection technique. June 2011 Current Limit Threshold vs. Feedback Voltage CURRENT LIMIT THRESHOLD (A) 12 VIN = 48V 10 8 6 4 2 0 0.0 0.2 0.4 0.6 0.8 1.0 FEEDBACK VOLTAGE (V) Figure 4. MIC28500 Current Limit Foldback Characteristic 15 M9999-060311-B Micrel, Inc. MIC28500 Internal MOSFET Gate Drive Figure 1 (Block Diagram) shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reverse biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1F to 1F is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. BST = 10mA x 3.33s/0.1F = 333mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the PVDD supply voltage. The nominal low-side gate drive voltage is PVDD and the nominal high-side gate drive voltage is approximately PVDD - VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. June 2011 16 M9999-060311-B Micrel, Inc. MIC28500 programmed to either lower end or higher end, the design needs optimization. Application Information Setting the Switching Frequency The MIC28500 is an adjustable-frequency, synchronous buck regulator featuring a unique digitally modified adaptive on-time control architecture. The switching frequency can be adjusted between 100kHz and 500kHz by changing the resistor divider connected network consisting of R18 and R19. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by Equation 3: Figure 5. Switching Frequency Adjustment L= The following formula gives the estimated switching frequency: R19 Eq. 2 fSW _ ADJ = fO x R18 + R19 Eq. 3 VIN(max) x fsw x 20% x IOUT(max) where: fSW = switching frequency, 300kHz 20% = ratio of AC ripple current to DC output current VIN(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is: Where fO = Switching Frequency when R18 is 100k and R19 being open, fO should be typically 500kHz. For more precise setting, it is recommended to use the following graph. Switching Frequency IL(pp) = 500 VOUT x (VIN(max) - VOUT ) VIN(max) x fsw x L Eq. 4 R18 = 100k, IOUT =1A 450 The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. VIN = 48V 400 SW FREQ (kHz) VOUT x (VIN(max) - VOUT ) 350 300 VIN = 75V 250 IL(pk) =IOUT(max) + 0.5 x IL(pp) 200 Eq. 5 150 100 The RMS inductor current is used to calculate the I2R losses in the inductor. 50 0 10.00 100.00 1000.00 10000.00 R19 (k Ohm) 2 IL(RMS) = IOUT(max) + IL(PP) 12 2 Eq. 6 Figure 6. Switching Frequency vs. R19 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The The evaluation board design is optimized for a switching frequency of 250kHz. If the switching frequency is June 2011 17 M9999-060311-B Micrel, Inc. MIC28500 The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated in Equation 10: high frequency operation of the MIC28500 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by Equation 7: PINDUCTOR(Cu) = IL(RMS)2 x RWINDING 2 IL(PP) 2 + IL(PP) x ESR C VOUT(pp) = OUT x x C f 8 OUT SW Eq. 10 As described in the "Theory of Operation" subsection in Functional Description, the MIC28500 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the "Ripple Injection" subsection for more details. The voltage rating of the capacitor should be 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 11: Eq. 7 PWINDING(Ht) = RWINDING(20C) x (1 + 0.0042 x (TH - T20C)) Eq. 8 where: TH = temperature of wire under full load T20C = ambient temperature RWINDING(20C) = room temperature winding resistance (usually specified by the manufacturer) ICOUT (RMS) = Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are ceramic, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor's ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated: VOUT(pp) IL(PP) IL(PP) Eq. 11 12 The power dissipated in the output capacitor is: 2 PDISS(COUT ) = ICOUT (RMS) x ESR COUT Eq. 12 Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor's voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor's ESR. The peak input current is equal to the peak inductor current, so: Eq. 9 where: VOUT(pp) = peak-to-peak output voltage ripple IL(PP) = peak-to-peak inductor current ripple June 2011 ) where: COUT = output capacitance value fSW = switching frequency The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature: ESR COUT ( 18 M9999-060311-B Micrel, Inc. MIC28500 VIN = IL(pk) x CESR VFB(pp) ESR x IL (pp) Eq. 13 3) Virtually no ripple at the FB pin voltage due to the very low ESR of the output capacitors. The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN(RMS) IOUT(max) x D x (1 - D) Eq. 17 Eq. 14 The power dissipated in the input capacitor is: PDISS(CIN) = ICIN(RMS)2 x CESR Eq. 15 Figure 7a. Enough Ripple at FB Ripple Injection The VFB ripple required for proper operation of the MIC28500 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as a 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator can't sense it, then the MIC28500 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1) Enough ripple at the feedback voltage due to the large ESR of the output capacitors. As shown in Figure 7a, the converter is stable without any ripple injection. The feedback voltage ripple is: VFB(pp) = R2 x ESR COUT x IL (pp) R1 + R2 Figure 7b. Inadequate Ripple at FB Eq. 16 where IL(pp) is the peak-to-peak value of the inductor current ripple. 2) Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 7b. The typical Cff value is between 1nF and 22nF. With the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: June 2011 Figure 7c. Invisible Ripple at FB In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 7c. 19 M9999-060311-B Micrel, Inc. MIC28500 The injected ripple is: VFB(pp) = VIN x K div x D x (1 - D) x K div = R1//R2 R inj + R1//R2 1 fSW x Setting Output Voltage The MIC28500 requires two resistors to set the output voltage as shown in Figure 8. Eq. 18 Eq. 19 where VIN = Power stage input voltage D = duty cycle fSW = switching frequency = (R1//R2//Rinj) x Cff Figure 8. Voltage-Divider Configuration The output voltage is determined by Equation 23: In Equations 18 and 19, it is assumed that the time constant associated with Cff must be much greater than the switching period: 1 fSW x = T << 1 R1 VO = VFB x 1 + R2 where, VFB = 0.8V. A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: Eq. 20 If the voltage divider resistors R1 and R2 are in the k range, a Cff of 1nF to 22nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 22nF if R1 and R2 are in k range. Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 21: K div = VFB(pp) VIN x fSW x D x (1 - D) R2 = 1 - 1) K div Eq. 24 Thermal Measurements Measuring the IC's case temperature is recommended to ensure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher then (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal Eq. 21 Eq. 22 Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. June 2011 VFB x R1 VOUT - VFB The inverting input voltage VINJ is clamped to 1.2V. As the injected ripple increases, the swing of VINJ will be clamped. The clamped VINJ reduces the line regulation because it is reflected back as a DC error on the FB terminal. Then the value of Rinj is obtained as: R inj = (R1//R2) x ( Eq. 23 20 M9999-060311-B Micrel, Inc. MIC28500 grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. External VIN Limiter Circuit The external VIN limiter circuit can be implemented either on EN pin or VDD pin. Only one of these VIN limiter circuits is required. The external VIN limiter circuit limits the minimum input to 30V. If the minimum input in certain applications is more than 30V then neither of these limiter circuits is needed. Enabling the device below 30V VIN and under maximum loading could heat up the device beyond safe operating conditions. The following figures show the external VIN limiter circuit on EN and VDD pins Figure 9A and Figure 9B respectively. The VIN limiter on EN consists of D5, R22, D6, R23 and VIN limiter on the VDD pin along with VDD supply regulator consists of D4, R14, D2, and Q1. Figure 9B: VIN Limiter On VDD pin Figure 9A: VIN Limiter On EN pin June 2011 21 M9999-060311-B Micrel, Inc. MIC28500 placed in parallel with the input capacitor. PCB Layout Guidelines * Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. Thickness of the copper planes is also important in terms of dissipating heat. The 2oz copper thickness is adequate from thermal point of view and also thick copper plain helps in terms of noise immunity. Keep in mind thinner planes can be easily penetrated by noise * In "Hot-Plug" applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. Inductor The following guidelines should be followed to insure proper operation of the MIC28500 converter. IC * The 2.2F ceramic capacitor, which is connected to the VDD pin, must be located right at the IC. The VDD pin is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the VDD and PGND pins. * The signal ground pin (SGND) must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. * Place the IC close to the point of load (POL). * Use fat traces to route the input and output power lines. * Signal and power grounds should be kept separate and connected at only one location. * Keep the inductor connection to the switch node (SW) short. * Do not route any digital lines underneath or close to the inductor. * Keep the switch node (SW) away from the feedback (FB) pin. * The CS pin should be connected directly to the SW pin to accurate sense the voltage across the lowside MOSFET. * To minimize noise, place a ground plane underneath the inductor. * The inductor can be placed on the opposite side of the PCB with respect to the IC. It does not matter whether the IC or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. The input and output capacitors must be placed on the same side of the board as the IC. Output Capacitor * Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. * Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. Input Capacitor * Place the input capacitor next to the power pins. * Place the input capacitors on the same side of the board and as close to the IC as possible. * Keep both the PVIN pin and PGND connections short. * Place several vias to the ground plane close to the input capacitor ground terminal. * Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. * Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be June 2011 If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. * The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. RC Snubber * Place the RC snubber on either side of the board and as close to the SW pin as possible. 22 M9999-060311-B Micrel, Inc. MIC28500 Evaluation Board Schematic Figure 10. Schematic of MIC28500 Evaluation Board (J9, J10, J11, R13, R15 are for testing purposes) June 2011 23 M9999-060311-B Micrel, Inc. MIC28500 Bill of Materials Item C1 C2, C3 C13 C6 Part Number EEU-FC2A101B GRM32ER72A225KA35L C8, C9 GRM32ER60J107ME20L Murata 12106D107MAT2A AVX(4) 06035C104KAT2A AVX GRM188R71H104KA93D GRM188R72A104KA35D AVX AVX TDK 06035C472KAT2A AVX Open Open C14, C15 Open Murata TDK BAT46W-TP MCC(5) BAT46W-7-F Diodes Inc. (6) MMXZ5232B-TP CMDZ5L6 1 2.2F Ceramic Capacitor, X7R, Size 1210, 100V 2 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 1 0.1F Ceramic Capacitor, X7R, Size 0603, 100V 1 2.2F Ceramic Capacitor, X7R, Size 0805, 10V 2 1nF Ceramic Capacitor, X7R, Size 0603, 100V 1 4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1 1F Ceramic Capacitor, X7R, Size 0805, 10V 1 Small Signal Schottky Diode 1 5.6V Zener Diode 1 Murata C1608X7R2A472K C2012X7R1A105K 100F Aluminum Capacitor, SMD, 100V Murata 06031C102KAT2A GRM21BR71A105KA01L Qty. TDK TDK C7 D2 Murata C1608X7R2A102K C4, C5 D1 Murata 0805ZC225MAT2A GRM21BR71A225KA01L Description TDK TDK GRM188R71H472KA01D C16 Murata C1608X7S2A104K GRM188R72A102KA01D C12 Murata (1) (2) TDK(3) C2012X7R1A225K C11 Panasonic C3225X7R2A225KT5 C1608X7R1H104K C10 Manufacturer MCC (7) Central Semi 1 D5 CMDZ24L-MIC Central Semi 24V Zener 1 D6 CMDZ3L6-MIC Central Semi 3.6V Zener 1 D4 Open L1 DR125-100-R Cooper Bussmann(8) 10H Inductor, 5.35A RMS, 7A Saturation Current Diodes Inc/ZETEX 1 100V NPN Transistor 1 Q1 FCX493 R1 CRCW06034R75FKEA Vishay Dale 4.75 Resistor, Size 0603, 1% 1 R2, R16 CRCW08051R21FKEA Vishay Dale 1.21 Resistor, Size 0805, 1% 2 R3 CRCW060319K6FKEA Vishay Dale 19.6k Resistor, Size 0603, 1% 1 R4 CRCW060310K0FKEA Vishay Dale 10k Resistor, Size 0603, 1% 1 R5 CRCW060380K6FKEA Vishay Dale 80.6k Resistor, Size 0603, 1% 1 R6 CRCW060340K2FKEA Vishay Dale 40.2k Resistor, Size 0603, 1% 1 June 2011 24 M9999-060311-B Micrel, Inc. MIC28500 Bill of Materials (Continued) Item Part Number Manufacturer Description Qty. R7 CRCW060320K0FKEA Vishay Dale 20k Resistor, Size 0603, 1% R8 CRCW060311K5FKEA Vishay Dale 11.5k Resistor, Size 0603, 1% 1 R9 CRCW06038K06FKEA Vishay Dale 8.06k Resistor, Size 0603, 1% 1 R10, R23 CRCW06034K75FKEA Vishay Dale 4.75k Resistor, Size 0603, 1% 1 R11 CRCW06033K24FKEA Vishay Dale 3.24k Resistor, Size 0603, 1% 1 R12 CRCW06031K91FKEA Vishay Dale 1.91k Resistor, Size 0603, 1% 1 R13, R24 CRCW06030000Z0EAHP Vishay Dale 0 Resistor, Size 0603 2 R14 CRCW080510K0JNEA Vishay Dale 10k Resistor, Size 0805, 1% 1 R15 CRCW060349R9FKEA Vishay Dale 49.9 Resistor, Size 0603, 1% 1 R17 (OPEN) CRCW0603715RFKEA Vishay Dale 715 Resistor, Size 0603, 1% R18, R19 CRCW0603100KFKEAHP Vishay Dale 100k Resistor, Size 0603, 1% 2 R20 1 CRCW06032R00FKEA Vishay Dale 2 Resistor, Size 0603, 1% 1 R21 (OPEN) CRCW060333K2FKEA Vishay Dale 33.2k Resistor, Size 0603, 1% 1 R22 Vishay Dale 36.5k Resistor, Size 0603, 1% 1 75V/4A Synchronous Buck DC-DC Regulator 1 U1 CRCW060336K5FKEA MIC28500YJL Micrel. Inc. (9) Notes: 1. Panasonic: www.panasonic.com. 2. Murata: www.murata.com. 3. TDK: www.tdk.com. 4. AVX: www.avx.com. 5. MCC: www.mccsemi.com. 6. Diode Inc.: www.diodes.com. 7. Central Semi: www.centralsemi.com. 8. Cooper: www.cooperbussman.com. 9. Micrel, Inc.: www.micrel.com. June 2011 25 M9999-060311-B Micrel, Inc. MIC28500 PCB Layout Figure 11. MIC28500 Evaluation Board Top Layer Figure 12. MIC28500 Evaluation Board Mid-Layer 1 (Ground Plane) June 2011 26 M9999-060311-B Micrel, Inc. MIC28500 PCB Layout (Continued) Figure 13. MIC28500 Evaluation Board Mid-Layer 2 Figure 14. MIC28500 Evaluation Board Bottom Layer June 2011 27 M9999-060311-B Micrel, Inc. MIC28500 Recommended Land Pattern June 2011 28 M9999-060311-B Micrel, Inc. MIC28500 Package Information 28-Lead 5mm x 6mm MLF(R) (YJL) MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel's terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2010 Micrel, Incorporated. June 2011 29 M9999-060311-B Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Micrel: MIC28500YJL TR MIC28500YJL-TR