MIC28500
75V/4A Hyper Speed Control™
Synchronous DC-DC Buck Regulator
SuperSwitcher II™
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
General Description
The Micrel MIC28500 is an adjustable frequency,
synchronous buck regulator featuring a unique adaptive on-
time control architecture. The MIC28500 operates over an
input supply range of 30V to 75V and provides a regulated
output of up to 4A of output current. The output voltage is
adjustable down to 0.8V with a guaranteed accuracy of ±1%.
Micrel’s Hyper Speed Control™ architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This adaptive tON ripple control architecture combines the
advantages of fixed-frequency operation and fast transient
response in a single device.
The MIC28500 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Features
• Hyper Speed Control™ architecture enables
-High Delta V operation (VIN = 75V and VOUT = 0.8V)
-Small output capacitance
• 30V to 75V voltage input
• Adjustable output down to 0.8V
• ±1% FB accuracy
• Any Capacitor™ Stable
- Zero-ESR to high-ESR output capacitors
• 4A output current capability, up to 90% efficiency
• 100kHz to 500kHz switching frequency
• Internal compensation
• Foldback current-limit and “hiccup” mode short-circuit
protection
• Thermal shutdown
• Supports safe startup into a pre-biased load
• –40°C to +125°C junction temperature range
• 28-pin 5mm × 6mm MLF® package
Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
___________________________________________________________________________________________________________
Typical Application
Ef ficiency (V
IN
= 48V)
vs. Output Current
10
20
30
40
50
60
70
80
90
100
0123456
O UTP UT CURR ENT (A )
EFFI CIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.8V
f
SW
= 250kHz
June 2011 M9999-060311-B
Micrel, Inc. MIC28500
June 2011 2 M9999-060311-B
Ordering Information
Part Number Voltage Switching Frequency Junction Temperature Range Package Lead Finish
MIC28500YJL Adjustable Adjustable −40°C to +125°C 28-pin 5mm × 6mm MLF®Pb-Free
Pin Configur ation
28-Pin 5mm × 6mm MLF® (YJL)
Pin Description
Pin
Number Pin Name Pin Function
13, 14, 15,
16, 17, 18,
19
PVIN
High-Side internal N-channel MOSFET Drain Connection (Input): The PVIN operating voltage range
is from 30V to 75V. Input capacitors between the PVIN pins and the power ground (PGND) are
required and keep the connection short. Enabling the device below 30V VIN and under maximum
loading could heat up the device beyond safe operating conditions.
24 EN
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA). Do not pull the EN pin above the VDD supply. Enabling the device below
30V VIN and under maximum loading could heat up the device beyond safe operating conditions.
25 FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
26 SGND
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to
the PGND Pad on the top layer, see PCB layout guidelines for details.
27 VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC28500. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin to the
PGND pin must be placed next to the IC. VDD must be powered up at the same time or after PVIN to
make the soft-start function correctly.
2, 5, 6, 7,
8, 21 PGND
Power Ground. PGND is the ground path for the MIC28500 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The loop for the power ground should be
as small as possible and separate from the signal ground (SGND) loop.
22 CS
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high-speed switching on this pin, the CS pin should be routed away from sensitive
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.
Micrel, Inc. MIC28500
June 2011 3 M9999-060311-B
Pin Description (Continued)
Pin
Number Pin Name Pin Function
20 BST
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1F is connected
between the BST pin and the SW pin.
4, 9, 10,
11, 12 SW Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain.
23 FS Frequency Setting Pin.
28 PVDD Power Supply for gate driver of bottom MOSFET.
1, 3 NC No Connect.
Micrel, Inc. MIC28500
June 2011 4 M9999-060311-B
Absolute Maximum Ratings(1, 2)
PVIN to PGND................................................ −0.3V to +76V
FS to PGND ....................................................−0.3V to PVIN
PVDD, VDD to PGND......................................... −0.3V to +6V
VSW, VCS to PGND ..............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 82V
VEN to PGND ......................................−0.3V to (VDD + 0.3V)
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Operating Ratings(3)
Supply Voltage (PVIN) ........................................ 30V to 75V
Bias Voltage (PVDD, VDD).................................. 4.5V to 5.5V
Enable Input (VEN)................................................. 0V to VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF® (θJA) ....................................36°C/W
Electrical Characteristics(5)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Power Supply Input
Input Voltage Range ( PVIN) 30 75 V
FS Voltage Range 2 75 V
VDD Bias Voltage
Operating Bias Voltage (VDD) 4.5 5 5.5 V
Under-Voltage Lockout Trip Level VDD Rising 3.2 3.85 4.45 V
UVLO Hysteresis 380 mV
Quiescent Supply Current (IVDD) VFB = 1.5V 1.4 3 mA
Shutdown Supply Current (IVDD) VDD = VBST = 5.5V, VIN = 48V
SW = unconnected, VEN = 0V
0.7 2 mA
Reference
0°C TJ 85°C (±1.0%) 0.792 0.8 0.808
Feedback Reference Voltage −40°C TJ 125°C (±1.5%) 0.788 0.8 0.812 V
Load Regulation IOUT = 0A to 4A 0.04 %
Line Regulation PVIN = 30 to 75V 0.1 %
FB Bias Current VFB = 0.8V -0.5 0.005 0.5 µA
Enable Control
EN Logic Level High 4.5V < VDD < 5.5V 1.2 0.85 V
EN Logic Level Low 4.5V < VDD < 5.5V 0.78 0.4 V
EN Bias Current VEN = 0V 50 100 µA
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.
Micrel, Inc. MIC28500
June 2011 5 M9999-060311-B
Electrical Characteristics(5) (Continued)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Oscillator
Switching Frequency (6) V
FS=PVIN 375 500 625 kHz
Maximum Duty Cycle (7) V
FB = 0V, VFS=PVIN 82 %
Minimum Duty Cycle VFB > 0.8V 0 %
Minimum Off-time 360 ns
Soft-Start
Soft-Start time 6 ms
Short Circuit Protection
VFB = 0.8V, TJ = 25°C 5.5 7 9 A Current-Limit Threshold
VFB = 0.8V, TJ = 125°C 4.2 5.5 8.5 A
Short-Circuit Current VFB = 0V 2 3.6 5.2 A
Internal FETs
Top-MOSFET RDS (ON) I
SW = 1A 175 m
Bottom-MOSFET RDS (ON) I
SW = 1A 31 m
SW Leakage Current PVIN = 36V, VSW = 36V, VEN = 0V, VBST = 41V 55 µA
PVIN Leakage Current PVIN = 36V, VSW = 0V, VEN = 0V, VBST = 41V 55 µA
Thermal Protection
Over-Temperature Shutdown TJ Rising 160
°C
Over-Temperature Shutdown
Hysteresis 25
°C
Notes:
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
Micrel, Inc. MIC28500
June 2011 6 M9999-060311-B
Typical Characteristics
V
IN
Operat ing Supply Current
vs. Input Voltage
0
4
8
12
16
20
30 35 40 45 50 55 60 65 70 75
I NPUT VO LTAGE ( V)
SUPPL Y CURRENT ( m A )
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
f
SW
= 250kHz
V
IN
Shut down Current
vs. Input Voltage
0
1
2
3
4
5
30 35 40 45 50 55 60 65 70 75
I NPUT VOLTA GE (V)
SHUTDO W N CURRENT ( m A)
V
DD
= 5V
V
EN
= 0V
V
DD
Operating Supply Current
vs. Input V oltage
0
2
4
6
8
10
30 35 40 45 50 55 60 65 70 75
INPUT VOLTA GE (V)
V
DD
SUPPLY CURRENT (mA)
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Feedback Vol tage
vs. Input Voltage
0.792
0.796
0.800
0.804
0.808
30 35 40 45 50 55 60 65 70 75
I NPUT VO LTAG E (V)
FEEDBACK VOLTAGE (V)
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
f
SW
= 250kHz
Total Regulat ion
vs. Input Voltage
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
30 35 40 45 50 55 60 65 70 75
I NPUT VOLTAGE ( V)
TOTAL REGULATI ON (%)
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A to 4A
f
SW
= 250kHz
Current Lim it
vs. Input Vo ltage
0
3
6
9
12
15
30 35 40 45 50 55 60 65 70 75
INPUT VOLTA GE (V)
CURRENT L IMIT (A)
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Switching Frequency
vs. Input Volt age
100
150
200
250
300
30 35 40 45 50 55 60 65 70 75
I NPUT VO LTAGE ( V)
SWITCHING FREQUENCY ( k Hz)
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 1A
R18=100k
Ω
R19=100k
Ω
V
DD
Operating Supply Current
vs. Temperature
0
2
4
6
8
10
12
-50 -25 0 25 50 75 100 125
TEMPERA T URE ( °C)
SUPPLY CURRENT (m A)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
f
SW
= 250kHz
V
DD
Shutdown Current
vs. Temperat ure
0.0
0.2
0.4
0.6
0.8
1.0
-50-25 0 255075100125
TEMPERA TURE (° C)
SHUTDOWN CURRENT (m A)
V
IN
= 48V
I
OUT
= 0A
V
DD
= 5V
V
EN
= 0V
Micrel, Inc. MIC28500
June 2011 7 M9999-060311-B
Typical Characteristics (Continued)
V
DD
UVLO Threshold
vs. Temperat ure
3.3
3.4
3.5
3.6
3.7
3.8
3.9
4.0
4.1
4.2
-50 -25 0 25 50 75 100 125
TEMPERA T URE (° C)
V
DD
THRESHOLD ( V)
Rising
Falling
V
IN
= 48V
I
OUT
= 0A
V
IN
Operating Supply Current
vs. Temperat u re
0
4
8
12
16
20
-50-25 0 255075100125
TEMPERA TURE (° C)
SUPPLY CURRENT ( mA)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
f
SW
= 250kHz
V
IN
Shut dow n Current
vs. Temperat ure
0
1
2
3
4
5
-50 -25 0 25 50 75 100 125
TEMPERATURE (° C)
SHUTDOWN CURRENT (mA)
V
IN
= 48V
V
DD
= 5V
V
EN
= 0V
I
OUT
= 0A
Current Limi t
vs. Temperat ure
0
3
6
9
12
15
-50 -25 0 25 50 75 100 125
TEM PERATURE (° C)
CURRENT LIM IT (A)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Feedback Vol tage
vs. Temperature
0.792
0.796
0.800
0.804
0.808
-50 -25 0 25 50 75 100 125
TEMPERATURE (° C)
FEEBACK VOLTAG E ( V)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
Load Regulation
vs. Temperat ure
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0.4%
-50 -25 0 25 50 75 100 125
TEM PERATURE ( °C)
LO AD REGULATIO N ( % )
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A to 4A
f
SW
= 250kHz
Li ne Regul ati on
vs. Temperat ure
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0.4%
-50 -25 0 25 50 75 100 125
TEMPERA T URE (° C)
LI NE REGULATIO N (%)
V
IN
= 30V to
75V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
Switching Frequency
vs. Output Current
100
150
200
250
300
01234
O UTPUT CURRENT (A)
SW ITCHI NG FREQUENCY (kHz)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
-40C
25C
125C
EN Bias Current
vs. Temperature
0
20
40
60
80
100
-50 -25 0 25 50 75 100 125
TEM PERATURE ( °C)
EN BI AS CURRENT (µA)
V
IN
= 48V
V
DD
= 5V
V
EN
= 0V
Micrel, Inc. MIC28500
June 2011 8 M9999-060311-B
Typical Characteristics (Continued)
Enable Threshold
vs. Temperat ure
0.5
0.6
0.7
0.8
0.9
1.0
-50 -25 0 25 50 75 100 125
TEM PERATURE (°C)
ENABLE THRESHO LD ( V)
Falling
Rising
V
IN
= 48V
V
DD
= 5V
Efficiency
vs. O utput Current
50
55
60
65
70
75
80
85
90
95
01234
O UTPUT CURRENT ( A)
EFFI CIENCY (% )
30V
IN
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
48V
IN
75V
IN
Feedback V o ltag e
vs. Ou tput Cu rrent
0.792
0.796
0.800
0.804
0.808
01234
O UTPUT CURRENT (A )
FEEDBACK VOLTAGE (V)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Li ne Regulat ion
vs. Output Current
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
01234
O UTPUT CURRENT ( A)
LI NE RE GULATI ON ( % )
V
IN
= 30V to 75V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Micrel, Inc. MIC28500
June 2011 9 M9999-060311-B
Typical Characteristics (Continued)
Die Temperat ure* ( V
IN
= 30V)
vs. O u tput Cu rrent
0
20
40
60
80
01234
O UT PUT CURRENT (A )
DI E T E M P ERATURE (°C)
V
IN
= 30V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Die Temperature* (V
IN
= 48V)
vs. Output Current
0
20
40
60
80
01234
O UT PUT CURRENT (A )
DI E T EMPERATURE (° C)
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Die Tem p erature* (V
IN
= 75V)
vs. O utput Current
0
20
40
60
80
100
01234
O UTPUT CURRENT (A)
DI E TEM PERATURE (° C)
V
IN
= 75V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 250kHz
Efficiency (V
IN
= 30V)
vs. O utput Current
30
40
50
60
70
80
90
100
0123456
O U TPUT CU RRENT (A )
EFFICI ENCY (%)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
V
IN
= 30V
f
SW
= 250kHz
Ef ficiency ( V
IN
= 48V)
vs. O utput Current
10
20
30
40
50
60
70
80
90
100
0123456
O UT PUT CURREN T ( A)
EFFI CI ENCY (% )
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
V
IN
= 48V
f
SW
= 250kHz
Ef ficiency (V
IN
= 75V)
vs. O utput Current
10
20
30
40
50
60
70
80
90
100
0123456
OUTPUT CURRE NT (A)
EFFI CI ENCY ( % )
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
V
IN
= 75V
f
SW
= 250kHz
Th er mal De ra ting
0
1
2
3
4
25 40 55 70 85 100
Maxi m um Ambi ent Tem perat ur e (° C)
Load Current (A)
V
IN
= 48V
f
SW
=250kHz
L=10uH
T
j_MAX
=125°C
V
OUT
= 5V
V
OUT
= 3.3V V
OUT
= 2.5V
Therma l Dera ting
0
1
2
3
4
25 40 55 70 85 100
Maxi m um A m bi ent Temperat ure ( ° C)
Load Curre nt (A)
V
IN
= 48V
f
SW
=250kHz
L=10uH
T
j_MAX
=125°C
V
OUT
= 1.2V
V
OUT
= 0.8V
Therma l Derating
0
1
2
3
4
25 40 55 70 85 100
Maxi m um Ambi ent Tem perat ur e (° C)
Load Current (A)
VIN = 48V
VOUT = 12V
fSW=250kHz
L=33μH
Tj_MAX =125°C
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC28500 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz. finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
Micrel, Inc. MIC28500
June 2011 10 M9999-060311-B
Functional Characteristics
Micrel, Inc. MIC28500
June 2011 11 M9999-060311-B
Functional Characteristics (Continued)
Micrel, Inc. MIC28500
June 2011 12 M9999-060311-B
Functional Characteristics (Continued)
Micrel, Inc. MIC28500
June 2011 13 M9999-060311-B
Functional Diagram
Figure 1. MIC28500 Block Diagram
Micrel, Inc. MIC28500
June 2011 14 M9999-060311-B
Functional Description
The MIC28500 is an adaptive ON-time synchronous
step-down DC-DC regulator. It is designed to operate
over a wide input voltage range from, 30V to 75V, and
provides a regulated output voltage at up to 4A of output
current. A digitally modified adaptive ON-time control
scheme is employed in to obtain a constant switching
frequency and to simplify the control compensation.
Over current protection is implemented without the use
of an external sense resistor. The device includes an
internal soft-start function which reduces the power
supply input surge current at start-up by controlling the
output voltage rise time.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC28500. The output voltage is sensed by the
MIC28500 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
SW
f×
=
IN
OUT
ed)ON(estimat V
V
t Eq. 1
where VOUT is the output voltage and VIN is the power
stage input voltage and fSW is the switching frequency.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, then the MIC28500 control logic will apply the
tOFF(min) instead. The minimum tOFF(min) period is required
to maintain enough energy in the boost capacitor (CBST)
to drive the high-side MOSFET. The maximum duty
cycle is obtained from the 360ns tOFF(min):
SS
OFF(min)S
max t
360ns
1
t
tt
D−=
−
= Eq. 2
where tS = 1/fSW. It is not recommended to use
MIC28500 with a OFF-time close to tOFF(min) during
steady-state operation..
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 75V to 1.0V. The minimum tON
measured on the MIC28500 evaluation board is about
184ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
Figure 2 shows the MIC28500 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Figure 2. MIC28500 Control Loop Timing
Micrel, Inc. MIC28500
June 2011 15 M9999-060311-B
Figure 3 shows the operation of the MIC28500 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC28500 converter.
Figure 3. MIC28500 Load Transien t Response
Unlike true current-mode control, the MIC28500 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC28500 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC28500 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC28500 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC28500 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the peak inductor current
is greater than 7A, then the MIC28500 turns off the high-
side MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The current-limit threshold has a foldback
characteristic related to the feedback voltage, as shown
in Figure 4.
Current Limit Threshold
vs. Feedback Voltage
0
2
4
6
8
10
12
0.00.20.40.60.81.0
FEEDBACK VOLTAG E (V)
CURRENT LIM IT THRE SHOLD
(A)
V
IN
= 48V
Figure 4. MIC28500 Current Li mit Fold b ack Characteristic
Micrel, Inc. MIC28500
June 2011 16 M9999-060311-B
Internal MOSFET Gate Drive
Figure 1 (Block Diagram) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1F to 1F is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. BST =
10mA x 3.33s/0.1F = 333mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
The drive voltage is derived from the PVDD supply
voltage. The nominal low-side gate drive voltage is PVDD
and the nominal high-side gate drive voltage is
approximately PVDD – VDIODE, where VDIODE is the voltage
drop across D1. An approximate 30ns delay between the
high-side and low-side driver transitions is used to
prevent current from simultaneously flowing unimpeded
through both MOSFETs.
Micrel, Inc. MIC28500
June 2011 17 M9999-060311-B
Application Information
Setting the Switching Frequency
The MIC28500 is an adjustable-frequency, synchronous
buck regulator featuring a unique digitally modified
adaptive on-time control architecture. The switching
frequency can be adjusted between 100kHz and 500kHz
by changing the resistor divider connected network
consisting of R18 and R19.
Figure 5. Switching Frequency Adjustment
The following formula gives the estimated switching
frequency:
1918
19
OADJ_SW RR
R
ff +
×= Eq. 2
Where fO = Switching Frequency when R18 is 100k and
R19 being open, fO should be typically 500kHz. For more
precise setting, it is recommended to use the following
graph.
Switching Frequency
0
50
100
150
200
250
300
350
400
450
500
10.00 100.00 1000.00 10000.00
R19 (k Ohm)
SW FREQ (kHz
)
R18 = 100k, I
OUT
=1A
VIN = 48V
VIN = 75V
Figure 6. Switching Frequency vs. R19
The evaluation board design is optimized for a switching
frequency of 250kHz. If the switching frequency is
programmed to either lower end or higher end, the
design needs optimization.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by Equation 3:
OUT(max)swIN(max)
OUTIN(max)OUT
I20% f V
)V(VV
L×××
−×
= Eq. 3
where:
fSW = switching frequency, 300kHz
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
L f V
)V(VV
I
swIN(max)
OUTIN(max)OUT
L(pp) ××
−×
=Δ Eq. 4
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × IL(pp) Eq. 5
The RMS inductor current is used to calculate the I2R
losses in the inductor.
12
I
II
2
L(PP)
2
OUT(max)L(RMS) += Eq. 6
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
Micrel, Inc. MIC28500
June 2011 18 M9999-060311-B
high frequency operation of the MIC28500 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 7:
PINDUCTOR(Cu) = IL(RMS)
2 × RWINDING Eq. 7
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature:
PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 8
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are ceramic, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
L(PP)
OUT(pp)
CI
V
ESR OUT ≤ Eq. 9
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
IL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
()
2
CL(PP)
2
SWOUT
L(PP)
OUT(pp) OUT
ESRI
8fC
I
V×+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
××
=
Eq. 10
where:
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
Functional Description, the MIC28500 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be 20%
greater for aluminum electrolytic or OS-CON. The output
capacitor RMS current is calculated in Equation 11:
12
I
IL(PP)
(RMS)COUT = Eq. 11
The power dissipated in the output capacitor is:
OUTOUTOUT C
2
(RMS)C)DISS(C ESRIP ×= Eq. 12
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
Micrel, Inc. MIC28500
June 2011 19 M9999-060311-B
VIN = IL(pk) × CESR Eq. 13
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
D)(1DII OUT(max)CIN(RMS) −××≈ Eq. 14
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)
2 × CESR Eq. 15
Ripple Injection
The VFB ripple required for proper operation of the
MIC28500 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC28500 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
As shown in Figure 7a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
(pp)
LC
21
2
FB(pp) IESR
RR
R
VOUT ××
+
= Eq. 16
where IL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 7b. The typical Cff value is between 1nF and
22nF.
With the feedforward capacitor, the feedback voltage
ripple is very close to the output voltage ripple:
(pp)
LFB(pp) IESRV×
≈
Eq. 17
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
Figure 7a. Enough Ripple at FB
Figure 7b. Inadequate Ripple at FB
Figure 7c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 7c.
Micrel, Inc. MIC28500
June 2011 20 M9999-060311-B
The injected ripple is:
τ
×
××××=
SW
divINFB(pp) f
1
D)-(1DKVV Eq. 18
R1//R2R
//RR
K
inj
21
div +
= Eq. 19
where
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
In Equations 18 and 19, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
f
1
SW
<<=
×
ττ
Eq. 20
If the voltage divider resistors R1 and R2 are in the k
range, a Cff of 1nF to 22nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
22nF if R1 and R2 are in k range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 21:
D)(1D
f
V
V
KSW
IN
FB(pp)
div −×
×
×=
τ
Eq. 21
Then the value of Rinj is obtained as:
1)
K
1
()//R(RR
div
21
inj −×= Eq. 22
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC28500 requires two resistors to set the output
voltage as shown in Figure 8.
Figure 8. Voltage-Divider Configuration
The output voltage is determined by Equation 23:
⎟
⎠
⎞
⎜
⎝
⎛+×=
2
1
FBO R
R
1VV Eq. 23
where, VFB = 0.8V. A typical value of R1 can be between
3k and 10k. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
FBOUT
1
FB
2VV
RV
R−
×
= Eq. 24
The inverting input voltage VINJ is clamped to 1.2V. As
the injected ripple increases, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected back as a DC error on the FB
terminal.
Thermal Measurements
Measuring the IC’s case temperature is recommended to
ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
Micrel, Inc. MIC28500
June 2011 21 M9999-060311-B
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
External VIN Limiter Circuit
The external VIN limiter circuit can be implemented either
on EN pin or VDD pin. Only one of these VIN limiter
circuits is required. The external VIN limiter circuit limits
the minimum input to 30V. If the minimum input in
certain applications is more than 30V then neither of
these limiter circuits is needed. Enabling the device
below 30V VIN and under maximum loading could heat
up the device beyond safe operating conditions. The
following figures show the external VIN limiter circuit on
EN and VDD pins Figure 9A and Figure 9B respectively.
The VIN limiter on EN consists of D5, R22, D6, R23 and
VIN limiter on the VDD pin along with VDD supply
regulator consists of D4, R14, D2, and Q1.
Figure 9A: VIN Limiter On EN pin
Figure 9B: VIN Limiter On VDD pin
Micrel, Inc. MIC28500
June 2011 22 M9999-060311-B
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths. Thickness of the copper planes
is also important in terms of dissipating heat. The 2oz
copper thickness is adequate from thermal point of view
and also thick copper plain helps in terms of noise
immunity. Keep in mind thinner planes can be easily
penetrated by noise
The following guidelines should be followed to insure
proper operation of the MIC28500 converter.
IC
• The 2.2µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
• The signal ground pin (SGND) must be connected
directly to the ground planes. Do not route the
SGND pin to the PGND Pad on the top layer.
• Place the IC close to the point of load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
• Place the input capacitor next to the power pins.
• Place the input capacitors on the same side of the
board and as close to the IC as possible.
• Keep both the PVIN pin and PGND connections
short.
• Place several vias to the ground plane close to the
input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close to
the inductor.
• Keep the switch node (SW) away from the feedback
(FB) pin.
• The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low-
side MOSFET.
• To minimize noise, place a ground plane underneath
the inductor.
• The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
RC Snubber
• Place the RC snubber on either side of the board
and as close to the SW pin as possible.
Micrel, Inc. MIC28500
June 2011 23 M9999-060311-B
Evaluation Board Schematic
Figure 10. Schematic of MIC28500 Evaluation Board
(J9, J10, J11, R13, R15 are for testing pu rposes)
Micrel, Inc. MIC28500
June 2011 24 M9999-060311-B
Bill of Materials
Item Part Number Manufacturer Description Qty.
C1 EEU-FC2A101B Panasonic
(1) 100µF Aluminum Capacitor, SMD, 100V 1
GRM32ER72A225KA35L Murata
(2)
C2, C3
C3225X7R2A225KT5 TDK(3)
2.2µF Ceramic Capacitor, X7R, Size 1210, 100V 2
GRM32ER60J107ME20L Murata
C13
12106D107MAT2A AVX(4) 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
06035C104KAT2A AVX
GRM188R71H104KA93D Murata
C6
C1608X7R1H104K TDK
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 1
GRM188R72A104KA35D Murata
C10
C1608X7S2A104K TDK
0.1µF Ceramic Capacitor, X7R, Size 0603, 100V 1
0805ZC225MAT2A AVX
GRM21BR71A225KA01L Murata
C8, C9
C2012X7R1A225K TDK
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 2
GRM188R72A102KA01D Murata
C1608X7R2A102K TDK
C11
06031C102KAT2A AVX
1nF Ceramic Capacitor, X7R, Size 0603, 100V 1
GRM188R71H472KA01D Murata
C1608X7R2A472K TDK
C12
06035C472KAT2A AVX
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1
GRM21BR71A105KA01L Murata
C16
C2012X7R1A105K TDK
1µF Ceramic Capacitor, X7R, Size 0805, 10V 1
C4, C5 Open
C7 Open
C14, C15 Open
BAT46W-TP MCC(5)
D1
BAT46W-7-F Diodes Inc.
(6)
Small Signal Schottky Diode 1
MMXZ5232B-TP MCC 1 D2
CMDZ5L6 Central Semi(7)
5.6V Zener Diode
1
D5 CMDZ24L-MIC Central Semi 24V Zener 1
D6 CMDZ3L6-MIC Central Semi 3.6V Zener 1
D4 Open
L1 DR125-100-R Cooper Bussmann(8) 10µH Inductor, 5.35A RMS, 7A Saturation Current 1
Q1 FCX493 Diodes Inc/ZETEX 100V NPN Transistor 1
R1 CRCW06034R75FKEA Vishay Dale 4.75 Resistor, Size 0603, 1% 1
R2, R16 CRCW08051R21FKEA Vishay Dale 1.21 Resistor, Size 0805, 1% 2
R3 CRCW060319K6FKEA Vishay Dale 19.6k Resistor, Size 0603, 1% 1
R4 CRCW060310K0FKEA
Vishay Dale 10k Resistor, Size 0603, 1% 1
R5 CRCW060380K6FKEA
Vishay Dale 80.6k Resistor, Size 0603, 1% 1
R6 CRCW060340K2FKEA
Vishay Dale 40.2k Resistor, Size 0603, 1% 1
Micrel, Inc. MIC28500
June 2011 25 M9999-060311-B
Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty.
R7 CRCW060320K0FKEA Vishay Dale 20k Resistor, Size 0603, 1% 1
R8 CRCW060311K5FKEA Vishay Dale 11.5k Resistor, Size 0603, 1% 1
R9 CRCW06038K06FKEA Vishay Dale 8.06k Resistor, Size 0603, 1% 1
R10, R23 CRCW06034K75FKEA Vishay Dale 4.75k Resistor, Size 0603, 1% 1
R11 CRCW06033K24FKEA Vishay Dale 3.24k Resistor, Size 0603, 1% 1
R12 CRCW06031K91FKEA Vishay Dale 1.91k Resistor, Size 0603, 1% 1
R13, R24 CRCW06030000Z0EAHP Vishay Dale 0 Resistor, Size 0603 2
R14 CRCW080510K0JNEA Vishay Dale 10k Resistor, Size 0805, 1% 1
R15 CRCW060349R9FKEA Vishay Dale 49.9 Resistor, Size 0603, 1% 1
R17 (OPEN) CRCW0603715RFKEA Vishay Dale 715 Resistor, Size 0603, 1%
R18, R19 CRCW0603100KFKEAHP Vishay Dale 100k Resistor, Size 0603, 1% 2
R20 CRCW06032R00FKEA Vishay Dale 2 Resistor, Size 0603, 1% 1
R21 (OPEN) CRCW060333K2FKEA Vishay Dale 33.2k Resistor, Size 0603, 1% 1
R22 CRCW060336K5FKEA Vishay Dale 36.5k Resistor, Size 0603, 1% 1
U1 MIC28500YJL Micrel. Inc.(9) 75V/4A Synchronous Buck DC-DC Regulator 1
Notes:
1. Panasonic: www.panasonic.com.
2. Murata: www.murata.com.
3. TDK: www.tdk.com.
4. AVX: www.avx.com.
5. MCC: www.mccsemi.com.
6. Diode Inc.: www.diodes.com.
7. Central Semi: www.centralsemi.com.
8. Cooper: www.cooperbussman.com.
9. Micrel, Inc.: www.micrel.com.
Micrel, Inc. MIC28500
June 2011 26 M9999-060311-B
PCB Layout
Figure 11. MIC28500 Evaluation Board Top Layer
Figure 12. MIC28500 Evaluation Board Mid-Layer 1 (Ground Plane)
Micrel, Inc. MIC28500
June 2011 27 M9999-060311-B
PCB Layout (Continued)
Figure 13. MIC28500 Evaluation Board Mid-Layer 2
Figure 14. MIC28500 Evaluation Board Bottom Layer
Micrel, Inc. MIC28500
June 2011 28 M9999-060311-B
Recommended Land Pattern
Micrel, Inc. MIC28500
June 2011 29 M9999-060311-B
Package Information
28-Lead 5mm x 6mm MLF® (YJL)
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