MP1471A
High-Efficiency, 3A
16V, 500kHz Synchronous,
Step-Down Converter In a 6-Pin TSOT 23
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DESCRIPTION
The MP1471A is a high-frequency,
synchronous, rectified, step-down, switch-mode
converter with internal power MOSFETs. It
offers a very compact solution to achieve a 3A
output current over a wide input supply range,
with excellent load and line regulation. The
MP1471A has synchronous-mode operation for
higher efficiency over the output current-load
range.
Current-mode operation provides fast transient
response and eases loop stabilization.
Protection features include over-current
protection and thermal shutdown.
The MP1471A requires a minimal number of
readily-available, standard, external
components and is available in a space-saving
6-pin TSOT23 package.
FEATURES
Wide 4.5V-to-16V Operating Input Range
110mΩ/57mΩ Low-RDS(ON) Internal Power
MOSFETs
Proprietary Switching-Loss–Reduction
Technology
High-Efficiency Synchronous-Mode
Operation
Fixed 500kHz Switching Frequency
Internal AAM Power-Save Mode for High
Efficiency at Light Load
Internal Soft-Start
Over-Current Protection and Hiccup
Thermal Shutdown
Output Adjustable from 0.8V
Available in a 6-pin TSOT-23 package
APPLICATIONS
Game Consoles
Digital Set-Top Boxes
Flat-Panel Television and Monitors
General Purposes
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Products, Quality Assurance page.
“MPS” and “The Future of Analog IC Technology” are registered trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
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ORDERING INFORMATION
Part Number*
Package
Top Marking
MP1471AGJ
TSOT23-6
AGM
* For Tape & Reel, add suffix –Z (e.g. MP1471AGJ–Z);
PACKAGE REFERENCE
ABSOLUTE MAXIMUM RATINGS (1)
VIN ................................................ -0.3V to 17V
VSW ....................................................................
-0.3V (-5V for <10ns) to 17V (19V for <10ns)
VBST ...................................................... VSW+6V
All Other Pins .................................. –0.3V to 6V
Continuous Power Dissipation (TA = +25°C) (2)
.......................................................... 1.25W
Junction Temperature .............................. 150°C
Lead Temperature ................................... 260°C
Storage Temperature ................. -65°C to 150°C
Recommended Operating Conditions (3)
Supply Voltage VIN .......................... 4.5V to 16V
Output Voltage VOUT ............... 0.8V to VIN*Dmax
Operating Junction Temp. (TJ) . -40°C to +125°C
Thermal Resistance (4) θJA θJC
TSOT-23-6 ............................ 100 ..... 55 ... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ (MAX), the junction-to-
ambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD (MAX) = (TJ
(MAX)-TA)/θJA. Exceeding the maximum allowable power
dissipation will cause excessive die temperature, and the
regulator will go into thermal shutdown. Internal thermal
shutdown circuitry protects the device from permanent
damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB.
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ELECTRICAL CHARACTERISTICS (5)
VIN = 12V, TA = 25°C, unless otherwise noted.
Parameter
Symbol
Condition
Min
Typ
Max
Units
Supply Current (Shutdown)
IIN
VEN = 0V
1
μA
Supply Current (Quiescent)
Iq
VEN = 2V, VFB = 1V
0.84
1.1
mA
HS Switch-On Resistance
HSRDS-ON
VBST-SW=5V
110
mΩ
LS Switch-On Resistance
LSRDS-ON
Vcc=5V
57
mΩ
Switch Leakage
SWLKG
VEN = 0V, VSW =12V
1
μA
Current Limit
ILIMIT
Duty=40%
3.9
4.6
A
Oscillator Frequency
fSW
VFB=0.75V
410
500
590
kHz
Maximum Duty Cycle
DMAX
VFB=700mV
88
92
%
Minimum On Time(5)
τON_MIN
60
ns
Feedback Voltage
VFB
788
804
820
mV
EN Rising Threshold
VEN_RISING
1.4
1.5
1.6
V
EN Falling Threshold
VEN_FALLING
1.23
1.32
1.41
V
EN Input Current
IEN
VEN=2V
1.8
μA
VEN=0
0
μA
VIN Under-Voltage Lockout
Threshold, Rising
INUVVth
3.9
4.15
4.4
V
VIN Under-Voltage Lockout
Threshold Hysteresis
INUVHYS
340
mV
Soft-Start Period
τSS
Vout from 0% to 100%
1.5
ms
Thermal Shutdown(5)
150
°C
Thermal Hysteresis(5)
20
°C
Notes:
5) Derived from bench characterization. Not tested in production.
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TYPICAL CHARACTERISTICS
VIN = 12V, VOUT = 3.3V, L = 4.7µH, TA = +25°C, unless otherwise noted.
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TYPICAL PERFORMANCE CHARACTERISTICS
Performance waveforms are tested on the evaluation board of the Design Example section.
VIN = 12V, VOUT = 3.3V, L = 4.7µH, TA = +25°C, unless otherwise noted.
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TYPICAL PERFORMANCE CHARACTERISTICS
(continued)
Performance waveforms are tested on the evaluation board of the Design Example section.
VIN = 12V, VOUT = 3.3V, L = 4.7µH, TA = +25°C, unless otherwise noted.
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TYPICAL PERFORMANCE CHARACTERISTICS
(continued)
Performance waveforms are tested on the evaluation board of the Design Example section.
VIN = 12V, VOUT = 3.3V, L = 4.7µH, TA = +25°C, unless otherwise noted.
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PIN FUNCTIONS
Package
Pin #
Name
Description
1
GND
System Ground. Reference ground of the regulated output voltage: requires extra care
during PCB layout. Connect to GND with copper traces and vias.
2
SW
Switch Output. Connect using wide a PCB trace.
3
IN
Supply Voltage. The MP1471A operates from a 4.5V-to-16V input rail. Requires C1 to
decouple the input rail. Connect using a wide PCB trace.
4
FB
Feedback. Connect to the tap of an external resistor divider from the output to GND to set
the output voltage. The frequency fold-back comparator lowers the oscillator frequency
when the FB voltage drops below 140mV to prevent current-limit runaway during a short
circuit fault.
5
EN
EN=HIGH to enable the MP1471A. For automatic start-up, connect EN to VIN using a
100kΩ resistor.
6
BST
Bootstrap. Connect a capacitor and a resistor between SW and BST pins to form a floating
supply across the high-side switch driver. Use a 1µF BST capacitor.
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BLOCK DIAGRAM
Figure 1: Functional Block Diagram
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OPERATION
The MP1471A is a high-frequency,
synchronous, rectified, step-down, switch-mode
converter with internal power MOSFETs. It
offers a very compact solution to achieve a 3A
output current over a wide input supply range,
with excellent load and line regulation.
The MP1471A operates in a fixed-frequency,
peak-current–control mode to regulate the
output voltage. An internal clock initiates the
PWM cycle to turn on the integrated high-side
power MOSFET. This MOSFET remains on
until its current reaches the value set by the
COMP voltage. When the power switch is off, it
remains off until the next clock cycle starts. If
the current in the power MOSFET does not
reach the COMP set current value within 90%
of one PWM period, the power MOSFET is
forced to turn off.
Internal Regulator
The 5V internal regulator powers most of the
internal circuits. This regulator takes VIN and
operates in the full VIN range. When VIN
exceeds 5.0V, the regulator output is in full
regulation. When VIN falls below 5.0V, the
output decreases.
Error Amplifier
The error amplifier compares the FB voltage
against the internal 0.8V reference (REF) and
outputs a current proportional to the difference
between the two. This output current charges or
discharges the internal compensation network
to form the COMP voltage, which is used to
control the power MOSFET current. The
optimized internal compensation network
minimizes the external component counts and
simplifies the control-loop design.
AAM Operation
The MP1471A has AAM (Advanced
Asynchronous Modulation) power-save mode
for light load. The AAM voltage is set at 0.5V
internally. Under the heavy load condition, the
VCOMP is higher than VAAM. When the clock goes
high, the high-side power MOSFET turns on
and remains on until VILsense reaches the value
set by the COMP voltage. The internal clock
resets every time when VCOMP exceed VAAM.
In light-load condition, the value of VCOMP is low.
When VCOMP is less than VAAM and VFB is less
than VREF, VCOMP ramps up until it exceeds VAAM.
During this time, the internal clock is blocked,
thus the MP1471A skips some pulses for PFM
(Pulse Frequency Modulation) mode and
achieves the light load power save.
Figure 2: Simplified AAM Control Circuit
When the load current is light, the inductor peak
current is set internally to about 650mA for
VIN=12V, VOUT=3.3V, and L=4.7μH. Figure 3
shows the inductor peak current vs. inductor
value curve.
Figure 3: Inductor Peak Current vs. Inductor
Value
Enable
EN is a digital control pin that turns the
regulator on and off: Drive EN HIGH to turn on
the regulator, drive it LOW to turn it off. An
internal 1MΩ resistor from EN to GND allows
EN to float to shut down the chip.
The EN pin is clamped internally using a 6.5V
series-Zener-diode as shown in Figure 4.
Connecting the EN input pin through a pullup
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resistor to the VIN voltage limits the EN input
current to less than 100μA.
For example, with 12V connected to Vin,
RPULLUP≥ (12V-6.5V)
100μA =55kΩ
Connecting the EN pin directly to a voltage
source without any pullup resistor requires
limiting the amplitude of the voltage source to ≤
6V to prevent damage to the Zener diode.
Figure 4: 6.5V Zener Diode
Under-Voltage Lockout (UVLO)
Under-voltage lockout (UVLO) protects the chip
from operating at an insufficient supply voltage.
The MP1471A UVLO comparator monitors the
output voltage of the internal regulator, VCC.
The UVLO rising threshold is about 4.15V while
its falling threshold is consistently 3.8V.
Internal Soft-Start
Soft-start prevents the converter output voltage
from overshooting during startup. When the
chip starts, the internal circuit generates a soft-
start voltage (SS) that ramps up from 0V to
1.2V: When SS falls below the internal
reference (REF), SS overrides REF so that the
error amplifier uses SS as the reference; when
SS exceeds REF, the error amplifier resumes
using REF as its reference. The SS time is
internally set to 1.5ms.
Over-Current-Protection and Hiccup
The MP1471A has a cycle-by-cycle over-
current limit for when the inductor current peak
value exceeds the set current-limit threshold.
First, when the output voltage drops until FB
falls below the Under-Voltage (UV) threshold
(typically 140mV) to trigger a UV event, the
MP1471A enters hiccup mode to periodically
restart the part. This protection mode is
especially useful when the output is dead-
shorted to ground. This greatly reduces the
average short-circuit current to alleviate thermal
issues and to protect the regulator. The
MP1471A exits hiccup mode once the over-
current condition is removed.
Thermal Shutdown
Thermal shutdown prevents the chip from
operating at exceedingly high temperatures.
When the silicon die temperature exceeds
150°C, it shuts down the whole chip. When the
temperature falls below its lower threshold
(typically 130°C) the chip is enabled again.
Floating Driver and Bootstrap Charging
An external bootstrap capacitor powers the
floating power MOSFET driver. This floating
driver has its own UVLO protection, with a
rising threshold of 2.2V and a hysteresis of
150mV. VIN regulates the bootstrap capacitor
voltage internally through D1, M1, R4, C4, L1
and C2 (Figure 5). If (VIN-VSW) exceeds 5V, U2
will regulate M1 to maintain a 5V BST voltage
across C4.
Figure 5 : Internal Bootstrap Charging Circuit
Start-Up and Shutdown
If both VIN and EN exceed their respective
thresholds, the chip starts. The reference block
starts first, generating a stable reference
voltage and currents, and then the internal
regulator is enabled. The regulator provides a
stable supply for the remaining circuits.
Three events can shut down the chip: EN low,
VIN low, and thermal shutdown. The shutdown
procedure starts by initially blocking the
signaling path to avoid any fault triggering. The
COMP voltage and the internal supply rail are
then pulled down. The floating driver is not
subject to this shutdown command.
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APPLICATION INFORMATION
Setting the Output Voltage
The external resistor divider sets the output
voltage. The feedback resistor R1 also sets the
feedback-loop bandwidth through the internal
compensation capacitor (see the Typical
Application circuit). Choose R1 around 10kΩ,
and R2 by:
OUT
R1
R2 V1
0.8V
Use a T-type network for when VOUT is low.
Figure 6: T-Type Network
Table 1 lists the recommended T-type resistors
value for common output voltages.
Table 1—Resistor Selection for Common Output
Voltages
VOUT
(V)
R1
(kΩ)
R2
(kΩ)
Rt
(kΩ)
LOUT
(μH)
COUT
(μF)
1.05
10
32.4
150
2.2
44
1.2
20.5
41.2
120
2.2
44
1.8
40.2
32.4
75
3.3
44
2.5
40.2
19.1
59
3.3
44
3.3
40.2
13
40.2
4.7
44
5
40.2
7.68
24.9
4.7
44
Selecting the Inductor
Use a 1µH-to-22µH inductor with a DC current
rating of at least 25% percent higher than the
maximum load current for most applications.
For highest efficiency, select an inductor with a
DC resistance less than 15mΩ. For most
designs, derive the inductance value from the
following equation.
OUT IN OUT
1IN L OSC
V (V V )
LV I f
Where ΔIL is the inductor ripple current. Choose
an inductor current approximately 30% of the
maximum load current. The maximum inductor
peak current is:
2
I
II L
LOAD)MAX(L
Under light-load conditions (below 100mA), use
a larger inductance for improved efficiency.
Selecting the Input Capacitor
The input current to the step-down converter is
discontinuous, and therefore requires a
capacitor to both supply the AC current to the
step-down converter and maintain the DC input
voltage. Use low ESR capacitors for the best
performance, such as ceramic capacitors with
X5R or X7R dielectrics of their low ESR and
small temperature coefficients. A 22µF
capacitor is sufficient for most applications.
The input capacitor (C1) requires an adequate
ripple current rating because it absorbs the
input switching. Estimate the RMS current in
the input capacitor with:
IN
OUT
IN
OUT
LOAD1C V
V
1
V
V
II
The worst-case condition occurs at VIN = 2VOUT,
where:
2
I
ILOAD
1C
For simplification, choose an input capacitor
with an RMS current rating greater than half the
maximum load current.
The input capacitor can be electrolytic, tantalum,
or ceramic. Place a small, high-quality, ceramic
capacitor (0.1μF) as close to the IC as possible
when using electrolytic or tantalum capacitors.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive input
voltage ripple. Estimate the input voltage ripple
caused by the capacitance with:
LOAD OUT OUT
IN IN
S IN
I V V
V1
f C1 V V
Selecting the Output Capacitor
The output capacitor (C2) maintains the DC
output voltage. Use ceramic, tantalum, or low-
ESR electrolytic capacitors. Use low ESR
capacitors to limit the output voltage ripple.
Estimate the output voltage ripple with:
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OUT OUT
OUT ESR
S 1 IN S
VV 1
V 1 R
f L V 8 f C2
Where L1 is the inductor value and RESR is the
equivalent series resistance (ESR) of the output
capacitor.
For ceramic capacitors, the capacitance
dominates the impedance at the switching
frequency and causes most of the output
voltage ripple. For simplification, estimate the
output voltage ripple with:
OUT OUT
OUT 2IN
S1
VV
ΔV 1 V
8 f L C2
For tantalum or electrolytic capacitors, the ESR
dominates the impedance at the switching
frequency. For simplification, the output ripple
can be approximated with:
OUT OUT
OUT ESR
IN
S1
VV
ΔV 1 R
f L V
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP1471A can be optimized for a wide range of
capacitance and ESR values.
External Bootstrap Diode
An external bootstrap (BST) diode can enhance
the efficiency of the regulator given the
following applicable conditions:
VOUT is 5V or 3.3V; and
Duty cycle is high: D=
IN
OUT
V
V
>65%
Connect the external BST diode from the output
of voltage regulator to the BST pin, as shown in
Figure 7.
Figure 7 : Optional External Bootstrap Diode
For most applications, use an IN4148 for the
external BST diode is IN4148, and a 1µF
capacitor for the BST capacitor.
PC Board Layout
PCB layout is very important to achieve stable
operation. For best results, use the following
guidelines and Figure 8 as reference.
1) Keep the connection between the input
ground and GND pin as short and wide as
possible.
2) Keep the connection between the input
capacitor and IN pin as short and wide as
possible.
3) Use short and direct feedback connections.
Place the feedback resistors and compensation
components as close to the chip as possible.
4) Route SW away from sensitive analog areas
such as FB.
Figure 8: Sample Layout
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Design Example
Below is a design example following the
application guidelines for the specifications:
Table 2—Design Example
VIN
12V
VOUT
3.3V
Io
3A
The detailed application schematic is shown in
Figure 10. The typical performance and circuit
waveforms have been shown in the Typical
Performance Characteristics section. For more
device applications, please refer to the related
Evaluation Board Datasheets.
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TYPICAL APPLICATION CIRCUITS
Figure 9: 12VIN, 5V/3A
Figure 10: 12VIN, 3.3V/3A
Figure 11: 12VIN, 2.5V/3A
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Figure 12: 12VIN, 1.8V/3A
Figure 13: 12VIN, 1.2V/3A
MP1471A – SYNCHRONOUS, STEP-DOWN CONVERTER WITH INTERNAL MOSFETS
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
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PACKAGE INFORMATION
TSOT23-6
FRONT VIEW
NOTE:
1) ALL DIMENSIONS ARE IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSION OR GATE BURR.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSION.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.10 MILLIMETERS MAX.
5) DRAWING CONFORMS TO JEDEC MO-193, VARIATION AB.
6) DRAWING IS NOT TO SCALE.
7) PIN 1 IS LOWER LEFT PIN WHEN READING TOP MARK
FROM LEFT TO RIGHT, (SEE EXAMPLE TOP MARK)
TOP VIEW RECOMMENDED LAND PATTERN
SEATING PLANE
SIDE VIEW
DETAIL "A"
SEE DETAIL '' A''
IAAAA
PIN 1 ID
See note 7
EXAMPLE
TOP MARK
