LM2596TVADJG - ON Semiconductor - Datasheet.Directory

November, 2008 − Rev. 0

1Publication Order Number:

LM2596/D

LM2596

3.0 A, Step-Down Switching

Regulator

The LM2596 regulator is monolithic integrated circuit ideally suited

for easy and convenient design of a step−down switching regulator

(buck converter). It is capable of driving a 3.0 A load with excellent

line and load regulation. This device is available in adjustable output

version and it is internally compensated to minimize the number of

external components to simplify the power supply design.

Since LM2596 converter is a switch−mode power supply, its

efficiency is significantly higher in comparison with popular

three−terminal linear regulators, especially with higher input voltages.

The LM2596 operates at a switching frequency of 150 kHz thus

allowing smaller sized filter components than what would be needed

with lower frequency switching regulators. Available in a standard

5−lead TO−220 package with several different lead bend options, and

D2PAK surface mount package.

The other features include a guaranteed $4% tolerance on output

voltage within specified input voltages and output load conditions, and

$15% on the oscillator frequency. External shutdown is included,

featuring 80 mA (typical) standby current. Self protection features

include switch cycle−by−cycle current limit for the output switch, as

well as thermal shutdown for complete protection under fault

conditions.

Features

•Adjustable Output Voltage Range 1.23 V − 37 V

•Guaranteed 3.0 A Output Load Current

•Wide Input Voltage Range up to 40 V

•150 kHz Fixed Frequency Internal Oscillator

•TTL Shutdown Capability

•Low Power Standby Mode, typ 80 mA

•Thermal Shutdown and Current Limit Protection

•Internal Loop Compensation

•Moisture Sensitivity Level (MSL) Equals 1

•Pb−Free Packages are Available

Applications

•Simple High−Efficiency Step−Down (Buck) Regulator

•Efficient Pre−Regulator for Linear Regulators

•On−Card Switching Regulators

•Positive to Negative Converter (Buck−Boost)

•Negative Step−Up Converters

•Power Supply for Battery Chargers

See detailed ordering and shipping information in the package

dimensions section on page 23 of this data sheet.

ORDERING INFORMATION

TO−220

TV SUFFIX

CASE 314B

Heatsink surface connected to Pin 3

TO−220

T SUFFIX

CASE 314D

Pin 1. Vin

2. Output

3. Ground

4. Feedback

5. ON/OFF

D2PAK

D2T SUFFIX

CASE 936A

Heatsink surface (shown as terminal 6 in

case outline drawing) is connected to Pin 3

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See general marking information in the device marking

section on page 23 of this data sheet.

DEVICE MARKING INFORMATION

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Figure 1. Typical Application and Internal Block Diagram

12 V

Unregulated

DC Input

33 mH

GND

+Vin

Cin

100 mF3ON

/OFF5

Output

Feedback

1N5822 Cout

220 mF

Typical Application (Adjustable Output Voltage Version)

Block Diagram

Unregulated

DC Input

+Vin

Cout

Feedback

Cin

Output

GND

ON/OFF

Reset

Latch

Thermal

Shutdown

150 kHz

Oscillator

1.235 V

Band-Gap

Reference

Freq

Shift

30 kHz

Comparator

Fixed Gain

Error Amplifier

Current

Limit

Driver

3.0 Amp

Switch

ON/OFF

3.1 V Internal

Regulator

Regulated

Output

Vout

Load

LM2596

5.0 V Regulated

Output 3.0 A Load

3.1k

1.0k

CFF

MAXIMUM RATINGS

Rating Symbol Value Unit

Maximum Supply Voltage Vin 45 V

ON/OFF Pin Input Voltage − −0.3 V ≤ V ≤ +Vin V

Output Voltage to Ground (Steady−State) − −1.0 V

Power Dissipation

Case 314B and 314D (TO−220, 5−Lead) PDInternally Limited W

Thermal Resistance, Junction−to−Ambient RqJA 65 °C/W

Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W

Case 936A (D2PAK) PDInternally Limited W

Thermal Resistance, Junction−to−Ambient RqJA 70 °C/W

Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W

Storage Temperature Range Tstg −65 to +150 °C

Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kW)−2.0 kV

Lead Temperature (Soldering, 10 seconds) −260 °C

Maximum Junction Temperature TJ150 °C

Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the

Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect

device reliability.

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PIN FUNCTION DESCRIPTION

Pin Symbol Description (Refer to Figure 1)

1 Vin This pin is the positive input supply for the LM2596 step−down switching regulator. In order to minimize voltage transi-

ents and to supply the switching currents needed by the regulator, a suitable input bypass capacitor must be present

(Cin in Figure 1).

2 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.5 V. It should be

kept in mind that the PCB area connected to this pin should be kept to a minimum in order to minimize coupling to

sensitive circuitry.

3 GND Circuit ground pin. See the information about the printed circuit board layout.

4 Feedback This pin is the direct input of the error amplifier and the resistor network R2, R1 is connected externally to allow pro-

gramming of the output voltage.

5 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total input supply

current to approximately 80 mA. The threshold voltage is typically 1.6 V. Applying a voltage above this value (up to

+Vin) shuts the regulator off. If the voltage applied to this pin is lower than 1.6 V or if this pin is left open, the regulator

will be in the “on” condition.

OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee

specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.)

Rating Symbol Value Unit

Operating Junction Temperature Range TJ−40 to +125 °C

Supply Voltage Vin 4.5 to 40 V

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SYSTEM PARAMETERS

ELECTRICAL CHARACTERISTICS Specifications with standard type face are for TJ = 25°C, and those with boldface type apply

over full Operating Temperature Range −40°C to +125°C

Characteristics Symbol Min Typ Max Unit

LM2596 (Note 1, Test Circuit Figure 15)

Feedback Voltage (Vin = 12 V, ILoad = 0.5 A, Vout = 5.0 V, ) VFB_nom 1.23 V

Feedback Voltage (8.5 V ≤ Vin ≤ 40 V, 0.5 A ≤ ILoad ≤ 3.0 A, Vout = 5.0 V) VFB 1.193

1.18

1.267

1.28

Efficiency (Vin = 12 V, ILoad = 3.0 A, Vout = 5.0 V) η−73 −%

Characteristics Symbol Min Typ Max Unit

Feedback Bias Current (Vout = 5.0 V) Ib25 100

200

Oscillator Frequency (Note 2) fosc 135

120

150 165

180

kHz

Saturation Voltage (Iout = 3.0 A, Notes 3 and 4) Vsat 1.5 1.8

2.0

Max Duty Cycle “ON” (Note 4) DC 95 %

Current Limit (Peak Current, Notes 2 and 3) ICL 4.2

3.5

5.6 6.9

7.5

Output Leakage Current (Notes 5 and 6)

Output = 0 V

Output = −1.0 V

IL0.5

6.0

2.0

Quiescent Current (Note 5) IQ5.0 10 mA

Standby Quiescent Current (ON/OFF Pin = 5.0 V (“OFF”))

(Note 6)

Istby 80 200

250

ON/OFF PIN LOGIC INPUT

Threshold Voltage 1.6 V

Vout = 0 V (Regulator OFF) VIH 2.2

2.4

Vout = Nominal Output Voltage (Regulator ON) VIL 1.0

0.8

ON/OFF Pin Input Current

ON/OFF Pin = 5.0 V (Regulator OFF) IIH −15 30 mA

ON/OFF Pin = 0 V (regulator ON) IIL −0.01 5.0 mA

1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.

When the LM2596 is used as shown in the Figure 15 test circuit, system performance will be as shown in system parameters section.

2. The oscillator frequency reduces to approximately 30 kHz in the event of an output short or an overload which causes the regulated output

voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by

lowering the minimum duty cycle from 5% down to approximately 2%.

3. No diode, inductor or capacitor connected to output (Pin 2) sourcing the current.

4. Feedback (Pin 4) removed from output and connected to 0 V.

5. Feedback (Pin 4) removed from output and connected to +12 V to force the output transistor “off”.

6. Vin = 40 V.

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IQ, QUIESCENT CURRENT (mA)

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)

out, OUTPUT VOLTAGE CHANGE (%)

, STANDBY QUIESCENT CURRENT (

TJ, JUNCTION TEMPERATURE (°C)

IO, OUTPUT CURRENT (A)

TJ, JUNCTION TEMPERATURE (°C)

Vin, INPUT VOLTAGE (V)

INPUT - OUTPUT DIFFERENTIAL (V)

TJ, JUNCTION TEMPERATURE (°C)

Figure 2. Normalized Output Voltage

TJ, JUNCTION TEMPERATURE (°C)

Figure 3. Line Regulation

Figure 4. Dropout Voltage Figure 5. Current Limit

Figure 6. Quiescent Current Figure 7. Standby Quiescent Current

ILoad = 200 mA

ILoad = 3.0 A

Vin = 12 V

Vin = 40 V

L1 = 33 mH

Rind = 0.1 W

ILoad = 500 mA

ILoad = 3.0 A

Vout = 5.0 V

Measured at

Ground Pin

TJ = 25°C

VON/OFF = 5.0 V

μA)

1.0

0.6

0.2

-0.2

-0.4

-1.0

1.4

1.2

1.0

0.8

0.6

0.4

0.2

-0.2

-0.4

-0.6

2.0

1.5

1.0

0.5

6.0

5.5

5.0

4.5

4.0

8.0

6.0

4.0

200

180

160

140

120

100

1251007550250-25-50 403530252015105.00

1251007550250-25-50 1251007550250-25-50

403530252015105.00 1251007550250-25-50

-0.8

-0.6

0.4

0.8 Vin = 20 V

ILoad = 500 mA

Normalized at TJ = 25°C

ILoad = 500 mA

TJ = 25°C

3.3 V and 5.0 V

12 V and 15 V

Istby

Vin = 25 V

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Vsat, SATURATION VOLTAGE (V)

2.0

2.5

3.0

4.0

Ib, FEEDBACK PIN CURRENT (nA)

, STANDBY QUIESCENT CURRENT (μA)I stby

, INPUT VOLTAGE (V)

TJ, JUNCTION TEMPERATURE (°C)

SWITCH CURRENT (A)

NORMALIZED FREQUENCY (%)

TJ, JUNCTION TEMPERATURE (°C)

Figure 8. Standby Quiescent Current

Vin, INPUT VOLTAGE (V)

Figure 9. Switch Saturation Voltage

Figure 10. Switching Frequency Figure 11. Minimum Supply Operating Voltage

Figure 12. Feedback Pin Current

TJ = 25°C

200

180

140

120

100

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

5.0

4.5

3.5

1.5

1.0

0.5

40302520151050 0 0.5 1.0 1.5 2.0 3.0

1251007550250-25

-50

TJ, JUNCTION TEMPERATURE (°C)

100

-20

-40

-60

-80

-100 1251007550250-25-50

160

35 2.5

-40°C

25°C

125°C

Vout ' 1.23 V

ILoad = 500 mA

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)

Vin

−9.0

−8.0

−7.0

−6.0

−5.0

−4.0

−3.0

−2.0

−1.0

0.0

1.0

−50 −25 0 25 50 75 100 125

VIN = 12 V Normalized

at 25°C

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2.0 A

100 ms/div2 ms/div

Figure 13. Switching Waveforms Figure 14. Load Transient Response

Vout = 5 V

A: Output Pin Voltage, 10 V/div

B: Switch Current, 2.0 A/div

C: Inductor Current, 2.0 A/div, AC−Coupled

D: Output Ripple Voltage, 50 mV/div, AC−Coupled

Horizontal Time Base: 5.0 ms/div

10 V

4.0 A

2.0 A

100 mV

Output

Voltage

Change

3.0 A

2.0 A

1.0 A

4.0 A

- 100 mV

Load

Current

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)

Figure 15. Typical Test Circuit

1N5822

33 mH

Output

Feedback

Cout

220 mF

Cin

100 mF

LM2596

53ON

/OFFGND

Vin

Load

Vout

5,000 V

Adjustable Output Voltage Versions

Vout +Vrefǒ1.0 )R2

R1Ǔ

R2 +R1ǒVout

Vref

1.0Ǔ

Where Vref = 1.23 V, R1

between 1.0 k and 5.0 k

8.5 V - 40 V

Unregulated

DC Input

CFF

LM2596

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PCB LAYOUT GUIDELINES

As in any switching regulator, the layout of the printed

circuit board is very important. Rapidly switching currents

associated with wiring inductance, stray capacitance and

parasitic inductance of the printed circuit board traces can

generate voltage transients which can generate

electromagnetic interferences (EMI) and affect the desired

operation. As indicated in the Figure 15, to minimize

inductance and ground loops, the length of the leads

indicated by heavy lines should be kept as short as possible.

For best results, single−point grounding (as indicated) or

ground plane construction should be used.

On the other hand, the PCB area connected to the Pin 2

(emitter of the internal switch) of the LM2596 should be

kept to a minimum in order to minimize coupling to sensitive

circuitry.

Another sensitive part of the circuit is the feedback. It is

important to keep the sensitive feedback wiring short. To

assure this, physically locate the programming resistors near

to the regulator, when using the adjustable version of the

LM2596 regulator.

DESIGN PROCEDURE

Buck Converter Basics

The LM2596 is a “Buck” or Step−Down Converter which

is the most elementary forward−mode converter. Its basic

schematic can be seen in Figure 16.

The operation of this regulator topology has two distinct

time periods. The first one occurs when the series switch is

on, the input voltage is connected to the input of the inductor.

The output of the inductor is the output voltage, and the

rectifier (or catch diode) is reverse biased. During this

period, since there is a constant voltage source connected

across the inductor, the inductor current begins to linearly

ramp upwards, as described by the following equation:

IL(on) +ǒVIN *VOUTǓton

During this “on” period, energy is stored within the core

material in the form of magnetic flux. If the inductor is

properly designed, there is sufficient energy stored to carry

the requirements of the load during the “off” period.

Figure 16. Basic Buck Converter

DVin RLoad

Cout

Power

Switch

The next period is the “off” period of the power switch.

When the power switch turns off, the voltage across the

inductor reverses its polarity and is clamped at one diode

voltage drop below ground by the catch diode. The current

now flows through the catch diode thus maintaining the load

current loop. This removes the stored energy from the

inductor. The inductor current during this time is:

IL(off) +ǒVOUT *VDǓtoff

This period ends when the power switch is once again

turned on. Regulation of the converter is accomplished by

varying the duty cycle of the power switch. It is possible to

describe the duty cycle as follows:

d+ton

T, where T is the period of switching.

For the buck converter with ideal components, the duty

cycle can also be described as:

d+Vout

Vin

Figure 17 shows the buck converter, idealized waveforms

of the catch diode voltage and the inductor current.

Power

Switch

Figure 17. Buck Converter Idealized Waveforms

Power

Switch

Off

Power

Switch

Off

Power

Switch

Power

Switch

Von(SW)

VD(FWD)

Time

ILoad(AV)

Imin

Ipk

Diode Diode

Power

Switch

Diode VoltageInductor Current

LM2596

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PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2596)

Procedure Example

Given Parameters:

Vout = Regulated Output Voltage

Vin(max) = Maximum DC Input Voltage

ILoad(max) = Maximum Load Current

Given Parameters:

Vout = 5.0 V

Vin(max) = 12 V

ILoad(max) = 3.0 A

1. Programming Output Voltage

To select the right programming resistor R1 and R2 value (see

Figure 1) use the following formula:

Resistor R1 can be between 1.0 k and 5.0 kW. (For best

temperature coefficient and stability with time, use 1% metal

film resistors).

Vout +Vref ǒ1.0 )R2

R1Ǔ

R2 +R1ǒVout

Vref

*1.0Ǔ

where Vref = 1.23 V

1. Programming Output Voltage (selecting R1 and R2)

Select R1 and R2:

R2 = 3.0 kW, choose a 3.0k metal film resistor.

R2 +R1ǒVout

Vref *1.0Ǔ+ǒ5V

1.23 V *1.0Ǔ

Vout +1.23ǒ1.0 )R2

R1ǓSelect R1 = 1.0 kW

2. Input Capacitor Selection (Cin)

To prevent large voltage transients from appearing at the input

and for stable operation of the converter, an aluminium or

tantalum electrolytic bypass capacitor is needed between the

input pin +Vin and ground pin GND This capacitor should be

located close to the IC using short leads. This capacitor should

have a low ESR (Equivalent Series Resistance) value.

For additional information see input capacitor section in the

“Application Information” section of this data sheet.

2. Input Capacitor Selection (Cin)

A 100 mF, 50 V aluminium electrolytic capacitor located near

the input and ground pin provides sufficient bypassing.

3. Catch Diode Selection (D1)

A. Since the diode maximum peak current exceeds the

regulator maximum load current the catch diode current

rating must be at least 1.2 times greater than the maximum

load current. For a robust design, the diode should have a

current rating equal to the maximum current limit of the

LM2596 to be able to withstand a continuous output short.

B. The reverse voltage rating of the diode should be at least

1.25 times the maximum input voltage.

3. Catch Diode Selection (D1)

A. For this example, a 3.0 A current rating is adequate.

B. For robust design use a 30 V 1N5824 Schottky diode or

any suggested fast recovery diode in the Table 2.

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PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2596) (CONTINUED)

Procedure Example

4. Inductor Selection (L1)

A. Use the following formula to calculate the inductor Volt x

microsecond [V x ms] constant:

B. Match the calculated E x T value with the corresponding

number on the vertical axis of the Inductor Value Selection

Guide shown in Figure 18. This E x T constant is a

measure of the energy handling capability of an inductor and

is dependent upon the type of core, the core area, the

number of turns, and the duty cycle.

C. Next step is to identify the inductance region intersected by

the E x T value and the maximum load current value on the

horizontal axis shown in Figure 18.

D. Select an appropriate inductor from Table 3.

The inductor chosen must be rated for a switching

frequency of 150 kHz and for a current rating of 1.15 x ILoad.

The inductor current rating can also be determined by

calculating the inductor peak current:

where ton is the “on” time of the power switch and

E T+ǒVIN *VOUT *VSATǓ

VOUT )VD

VIN *VSAT )VD

1000

150 kHz ǒV msǓ

Ip(max) +ILoad(max) )ǒVin *VoutǓton

ton +Vout

Vin

x1.0

fosc

4. Inductor Selection (L1)

A. Calculate E x T [V x ms] constant:

B. E x T = 27 [V x ms]

C. ILoad(max) = 3.0 A

Inductance Region = L40

D. Proper inductor value = 33 mH

Choose the inductor from Table 3.

E T+ǒ12 *5*1.5Ǔ 5)0.5

12 *5)0.5 1000

150 kHz ǒV msǓ

E T+ǒ5.5Ǔ 5.5

7.5 6.6ǒV msǓ

5. Output Capacitor Selection (Cout)

A. Since the LM2596 is a forward−mode switching regulator

with voltage mode control, its open loop has 2−pole−1−zero

frequency characteristic. The loop stability is determined by

the output capacitor (capacitance, ESR) and inductance

values.

For stable operation use recommended values of the output

capacitors in Table 1.

Low ESR electrolytic capacitors between 220uFand 1500uF

provide best results.

B. The capacitors voltage rating should be at least 1.5 times

greater than the output voltage, and often much higher

voltage rating is needed to satisfy low ESR requirement

5. Output Capacitor Selection (Cout)

A. In this example is recommended Nichicon PM

capacitors: 470 mF/35 V or 220 mF/35 V

6. Feedforward Capacitor (CFF)

It provides additional stability mainly for higher input voltages. For

Cff selection use Table 1. The compensation capacitor between

0.6 nF and 40 nF is wired in parallel with the output voltage setting

resistor R2, The capacitor type can be ceramic, plastic, etc..

6. Feedforward Capacitor (CFF)

In this example is recommended feedforward capacitor

15 nF or 5 nF.

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LM2596 Series Buck Regulator Design Procedures (continued)

Table 1. RECOMMENDED VALUES OF THE OUTPUT CAPACITOR AND FEEDFORWARD CAPACITOR

(Iload = 3 A)

Nichicon PM Capacitors

Vin (V) Capacity/Voltage Range/ESR (mF/V/mW)

40 1500/35/24 1000/35/29 1000/35/29 680/35/36 560/25/55 560/25/55 470/35/46 470/35/46

26 1200/35/26 820/35 680/35/36 560/35/41 470/25/65 470/25/65 330/35/60

22 1000/35/29 680/35/36 560/35/41 330/25/85 330/25/85 220/35/85

20 820/35/32 470/35/46 470/25/65 330/25/85 330/25/85 220/35/85

18 820/35/32 470/35/46 470/25/65 330/25/85 330/25/85 220/35/85

12 820/35/32 470/35/46 220/35/85 220/25/111

10 820/35/32 470/35/46 220/35/85

Vout (V) 2 4 6 9 12 15 24 28

CFF (nF] 40 15 5 2 1.5 1 0.6 0.6

15uH

22uH

33uH

47uH

68uH

100uH

150uH

220uH

L35L27

L36

L27

L42 L43

L44

L37

L38

L30

L29

L21

L22

L31

L39

L40

L32

L23

L15

L24

L40

L25

L34

0.6 0.8 1.0 1.5 2.0 2.5 3.0

E*T(V*us)

Maximum load current (A)

Figure 18. Inductor Value Selection Guides (For Continuous Mode Operation)

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Table 2. DIODE SELECTION

Schottky Fast Recovery

3.0 A 4.0 − 6.0 A 3.0 A 4.0 − 6.0 A

Through

Hole

Surface

Mount

Through

Hole

Surface

Mount

Through

Hole

Surface

Mount

Through

Hole

Surface

Mount

20 V 1N5820

MBR320P

SR302

SK32 1N5823

SR502

SB520

MUR320

31DF1

HER302

(all diodes

rated

to at least

100 V)

MURS320T3

MURD320

30WF10

(all diodes

rated

to at least

100 V)

MUR420

HER602

(all diodes

rated

to at least

100 V)

MURD620CT

50WF10

(all diodes

rated

to at least

100 V)

30 V 1N5821

MBR330

SR303

31DQ03

SK33

30WQ03

1N5824

SR503

SB530

50WQ03

40 V 1N5822

MBR340

SR304

31DQ04

SK34

30WQ04

MBRS340T3

MBRD340

1N5825

SR504

SB540

MBRD640CT

50WQ04

50 V MBR350

31DQ05

SR305

SK35

30WQ05

SB550 50WQ05

60 V MBR360

DQ06

SR306

MBRS360T3

MBRD360

50SQ080 MBRD660CT

NOTE: Diodes listed in bold are available from ON Semiconductor.

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Table 3. INDUCTOR MANUFACTURERS PART NUMBERS

Inductance

(mH)

Current

(A)

Schott Renco Pulse Engineering Coilcraft

Through

Hole

Surface

Mount

Through

Hole

Surface

Mount

Through

Hole

Surface

Mount Surface Mount

L15 22 0.99 67148350 67148460 RL−1284−22−43 RL1500−2

PE−53815 PE−53815−S DO3308−223

L21 68 0.99 67144070 67144450 RL−5471−5 RL1500−6

PE−53821 PE−53821−S DO3316−683

L22 47 1.17 67144080 67144460 RL−5471−6−PE−53822 PE−53822−S DO3316−473

L23 33 1.40 67144090 67144470 RL−5471−7−PE−53823 PE−53823−S DO3316−333

L24 22 1.70 67148370 67148480 RL−1283−22−43 −PE−53824 PE−53825−S DO3316−223

L25 15 2.10 67148380 67148490 RL−1283−15−43 −PE−53825 PE−53824−S DO3316−153

L26 330 0.80 67144100 67144480 RL−5471−1−PE−53826 PE−53826−S DO5022P−334

L27 220 1.00 67144110 67144490 RL−5471−2−PE−53827 PE−53827−S DO5022P−224

L28 150 1.20 67144120 67144500 RL−5471−3−PE−53828 PE−53828−S DO5022P−154

L29 100 1.47 67144130 67144510 RL−5471−4−PE−53829 PE−53829−S DO5022P−104

L30 68 1.78 67144140 67144520 RL−5471−5−PE−53830 PE−53830−S DO5022P−683

L31 47 2.20 67144150 67144530 RL−5471−6−PE−53831 PE−53831−S DO5022P−473

L32 33 2.50 67144160 67144540 RL−5471−7−PE−53932 PE−53932−S DO5022P−333

L33 22 3.10 67148390 67148500 RL−1283−22−43 −PE−53933 PE−53933−S DO5022P−223

L34 15 3.40 67148400 67148790 RL−1283−15−43 −PE−53934 PE−53934−S DO5022P−153

L35 220 1.70 67144170 −RL−5473−1−PE−53935 PE−53935−S−

L36 150 2.10 67144180 −RL−5473−4−PE−54036 PE−54036−S−

L37 100 2.50 67144190 −RL−5472−1−PE−54037 PE−54037−S−

L38 68 3.10 67144200 −RL−5472−2−PE−54038 PE−54038−S DO5040H−683ML

L39 47 3.50 67144210 −RL−5472−3−PE−54039 PE−54039−S DO5040H−473ML

L40 33 3.50 67144220 67148290 RL−5472−4−PE−54040 PE−54040−S DO5040H−333ML

L41 22 3.50 67144230 67148300 RL−5472−5−PE−54041 PE−54041−S DO5040H−223ML

L42 150 2.70 67148410 −RL−5473−4−PE−54042 PE−54042−S−

L43 100 3.40 67144240 −RL−5473−2−PE−54043 -

L44 68 3.40 67144250 −RL−5473−3−PE−54044 DO5040H−683ML

LM2596

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APPLICATION INFORMATION

EXTERNAL COMPONENTS

Input Capacitor (Cin)

The Input Capacitor Should Have a Low ESR

For stable operation of the switch mode converter a low

ESR (Equivalent Series Resistance) aluminium or solid

tantalum bypass capacitor is needed between the input pin

and the ground pin, to prevent large voltage transients from

appearing at the input. It must be located near the regulator

and use short leads. With most electrolytic capacitors, the

capacitance value decreases and the ESR increases with

lower temperatures. For reliable operation in temperatures

below −25°C larger values of the input capacitor may be

needed. Also paralleling a ceramic or solid tantalum

capacitor will increase the regulator stability at cold

temperatures.

RMS Current Rating of Cin

The important parameter of the input capacitor is the RMS

current rating. Capacitors that are physically large and have

large surface area will typically have higher RMS current

ratings. For a given capacitor value, a higher voltage

electrolytic capacitor will be physically larger than a lower

voltage capacitor, and thus be able to dissipate more heat to

the surrounding air, and therefore will have a higher RMS

current rating. The consequence of operating an electrolytic

capacitor beyond the RMS current rating is a shortened

operating life. In order to assure maximum capacitor

operating lifetime, the capacitor’s RMS ripple current rating

should be:

Irms > 1.2 x d x ILoad

where d is the duty cycle, for a buck regulator

d+ton

T+Vout

Vin

and d +ton

T+

|Vout|

|Vout|)V

for a buck*boost regulator.

Output Capacitor (Cout)

For low output ripple voltage and good stability, low ESR

output capacitors are recommended. An output capacitor

has two main functions: it filters the output and provides

regulator loop stability. The ESR of the output capacitor and

the peak−to−peak value of the inductor ripple current are the

main factors contributing to the output ripple voltage value.

Standard aluminium electrolytics could be adequate for

some applications but for quality design, low ESR types are

recommended.

An aluminium electrolytic capacitor’s ESR value is

related to many factors such as the capacitance value, the

voltage rating, the physical size and the type of construction.

In most cases, the higher voltage electrolytic capacitors have

lower ESR value. Often capacitors with much higher

voltage ratings may be needed to provide low ESR values

that, are required for low output ripple voltage.

Feedfoward Capacitor

(Adjustable Output Voltage Version)

This capacitor adds lead compensation to the feedback

loop and increases the phase margin for better loop stability.

For CFF selection, see the design procedure section.

The Output Capacitor Requires an ESR Value

That Has an Upper and Lower Limit

As mentioned above, a low ESR value is needed for low

output ripple voltage, typically 1% to 2% of the output

voltage. But if the selected capacitor’s ESR is extremely low

(below 0.05 W), there is a possibility of an unstable feedback

loop, resulting in oscillation at the output. This situation can

occur when a tantalum capacitor, that can have a very low

ESR, is used as the only output capacitor.

At Low Temperatures, Put in Parallel Aluminium

Electrolytic Capacitors with Tantalum Capacitors

Electrolytic capacitors are not recommended for

temperatures below −25°C. The ESR rises dramatically at

cold temperatures and typically rises 3 times at −25°C and

as much as 10 times at −40°C. Solid tantalum capacitors

have much better ESR spec at cold temperatures and are

recommended for temperatures below −25°C. They can be

also used in parallel with aluminium electrolytics. The value

of the tantalum capacitor should be about 10% or 20% of the

total capacitance. The output capacitor should have at least

50% higher RMS ripple current rating at 150 kHz than the

peak−to−peak inductor ripple current.

LM2596

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Catch Diode

Locate the Catch Diode Close to the LM2596

The LM2596 is a step−down buck converter; it requires a

fast diode to provide a return path for the inductor current

when the switch turns off. This diode must be located close

to the LM2596 using short leads and short printed circuit

traces to avoid EMI problems.

Use a Schottky or a Soft Switching

Ultra−Fast Recovery Diode

Since the rectifier diodes are very significant sources of

losses within switching power supplies, choosing the

rectifier that best fits into the converter design is an

important process. Schottky diodes provide the best

performance because of their fast switching speed and low

forward voltage drop.

They provide the best efficiency especially in low output

voltage applications (5.0 V and lower). Another choice

could be Fast−Recovery, or Ultra−Fast Recovery diodes. It

has to be noted, that some types of these diodes with an

abrupt turnoff characteristic may cause instability or

EMI troubles.

A fast−recovery diode with soft recovery characteristics

can better fulfill some quality, low noise design requirements.

Table 2 provides a list of suitable diodes for the LM2596

regulator. Standard 50/60 Hz rectifier diodes, such as the

1N4001 series or 1N5400 series are NOT suitable.

Inductor

The magnetic components are the cornerstone of all

switching power supply designs. The style of the core and

the winding technique used in the magnetic component’s

design has a great influence on the reliability of the overall

power supply.

Using an improper or poorly designed inductor can cause

high voltage spikes generated by the rate of transitions in

current within the switching power supply, and the

possibility of core saturation can arise during an abnormal

operational mode. Voltage spikes can cause the

semiconductors to enter avalanche breakdown and the part

can instantly fail if enough energy is applied. It can also

cause significant RFI (Radio Frequency Interference) and

EMI (Electro−Magnetic Interference) problems.

Continuous and Discontinuous Mode of Operation

The LM2596 step−down converter can operate in both the

continuous and the discontinuous modes of operation. The

regulator works in the continuous mode when loads are

relatively heavy, the current flows through the inductor

continuously and never falls to zero. Under light load

conditions, the circuit will be forced to the discontinuous

mode when inductor current falls to zero for certain period

of time (see Figure 19 and Figure 20). Each mode has

distinctively different operating characteristics, which can

affect the regulator performance and requirements. In many

cases the preferred mode of operation is the continuous

mode. It offers greater output power, lower peak currents in

the switch, inductor and diode, and can have a lower output

ripple voltage. On the other hand it does require larger

inductor values to keep the inductor current flowing

continuously, especially at low output load currents and/or

high input voltages.

To simplify the inductor selection process, an inductor

selection guide for the LM2596 regulator was added to this

data sheet (Figure 18). This guide assumes that the regulator

is operating in the continuous mode, and selects an inductor

that will allow a peak−to−peak inductor ripple current to be

a certain percentage of the maximum design load current.

This percentage is allowed to change as different design load

currents are selected. For light loads (less than

approximately 300 mA) it may be desirable to operate the

regulator in the discontinuous mode, because the inductor

value and size can be kept relatively low. Consequently, the

percentage of inductor peak−to−peak current increases. This

discontinuous mode of operation is perfectly acceptable for

this type of switching converter. Any buck regulator will be

forced to enter discontinuous mode if the load current is light

enough.

HORIZONTAL TIME BASE: 2.0 ms/DIV

Figure 19. Continuous Mode Switching Current

Waveforms

VERTRICAL RESOLUTION 1.0 A/DIV

2.0 A

0 A

2.0 A

0 A

Inductor

Current

Waveform

Power

Switch

Current

Waveform

Selecting the Right Inductor Style

Some important considerations when selecting a core type

are core material, cost, the output power of the power supply,

the physical volume the inductor must fit within, and the

amount of EMI (Electro−Magnetic Interference) shielding

that the core must provide. The inductor selection guide

covers different styles of inductors, such as pot core, E−core,

toroid and bobbin core, as well as different core materials

such as ferrites and powdered iron from different

manufacturers.

For high quality design regulators the toroid core seems to

be the best choice. Since the magnetic flux is contained

within the core, it generates less EMI, reducing noise

problems in sensitive circuits. The least expensive is the

bobbin core type, which consists of wire wound on a ferrite

rod core. This type of inductor generates more EMI due to

the fact that its core is open, and the magnetic flux is not

contained within the core.

When multiple switching regulators are located on the

same printed circuit board, open core magnetics can cause

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interference between two or more of the regulator circuits,

especially at high currents due to mutual coupling. A toroid,

pot core or E−core (closed magnetic structure) should be

used in such applications.

Do Not Operate an Inductor Beyond its

Maximum Rated Current

Exceeding an inductor’s maximum current rating may

cause the inductor to overheat because of the copper wire

losses, or the core may saturate. Core saturation occurs when

the flux density is too high and consequently the cross

sectional area of the core can no longer support additional

lines of magnetic flux.

This causes the permeability of the core to drop, the

inductance value decreases rapidly and the inductor begins

to look mainly resistive. It has only the DC resistance of the

winding. This can cause the switch current to rise very

rapidly and force the LM2596 internal switch into

cycle−by−cycle current limit, thus reducing the DC output

load current. This can also result in overheating of the

inductor and/or the LM2596. Different inductor types have

different saturation characteristics, and this should be kept

in mind when selecting an inductor.

0.4 A

0 A

0.4 A

0 A

Inductor

Current

Waveform

Power

Switch

Current

Waveform

Figure 20. Discontinuous Mode Switching Current

Waveforms

VERTICAL RESOLUTION 200 mA/DIV

HORIZONTAL TIME BASE: 2.0 ms/DIV

GENERAL RECOMMENDATIONS

Output Voltage Ripple and Transients

Source of the Output Ripple

Since the LM2596 is a switch mode power supply

regulator, its output voltage, if left unfiltered, will contain a

sawtooth ripple voltage at the switching frequency. The

output ripple voltage value ranges from 0.5% to 3% of the

output voltage. It is caused mainly by the inductor sawtooth

ripple current multiplied by the ESR of the output capacitor.

Short Voltage Spikes and How to Reduce Them

The regulator output voltage may also contain short

voltage spikes at the peaks of the sawtooth waveform (see

Figure 21). These voltage spikes are present because of the

fast switching action of the output switch, and the parasitic

inductance of the output filter capacitor. There are some

other important factors such as wiring inductance, stray

capacitance, as well as the scope probe used to evaluate these

transients, all these contribute to the amplitude of these

spikes. To minimize these voltage spikes, low inductance

capacitors should be used, and their lead lengths must be

kept short. The importance of quality printed circuit board

layout design should also be highlighted.

Unfiltered

Output

Voltage

Filtered

Output

Voltage

HORIZONTAL TIME BASE: 5.0 ms/DIV

Figure 21. Output Ripple Voltage Waveforms

VERTRICAL

Voltage spikes

caused by

switching action

of the output

switch and the

parasitic

inductance of the

output capacitor

RESOLUTION

20 mV/DIV

Minimizing the Output Ripple

In order to minimize the output ripple voltage it is possible

to enlarge the inductance value of the inductor L1 and/or to

use a larger value output capacitor. There is also another way

to smooth the output by means of an additional LC filter (20

mH, 100 mF), that can be added to the output (see Figure 30)

to further reduce the amount of output ripple and transients.

With such a filter it is possible to reduce the output ripple

voltage transients 10 times or more. Figure 21 shows the

difference between filtered and unfiltered output waveforms

of the regulator shown in Figure 30.

The lower waveform is from the normal unfiltered output

of the converter, while the upper waveform shows the output

ripple voltage filtered by an additional LC filter.

Heatsinking and Thermal Considerations

The Through−Hole Package TO−220

The LM2596 is available in two packages, a 5−pin

TO−220(T, TV) and a 5−pin surface mount D2PAK(D2T).

Although the TO−220(T) package needs a heatsink under

most conditions, there are some applications that require no

heatsink to keep the LM2596 junction temperature within

the allowed operating range. Higher ambient temperatures

require some heat sinking, either to the printed circuit (PC)

board or an external heatsink.

The Surface Mount Package D2PAK and its

Heatsinking

The other type of package, the surface mount D2PAK, is

designed to be soldered to the copper on the PC board. The

copper and the board are the heatsink for this package and

the other heat producing components, such as the catch

diode and inductor. The PC board copper area that the

package is soldered to should be at least 0.4 in2 (or 260 mm2)

and ideally should have 2 or more square inches (1300 mm2)

of 0.0028 inch copper. Additional increases of copper area

beyond approximately 6.0 in2 (4000 mm2) will not improve

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heat dissipation significantly. If further thermal

improvements are needed, double sided or multilayer PC

boards with large copper areas should be considered. In

order to achieve the best thermal performance, it is highly

recommended to use wide copper traces as well as large

areas of copper in the printed circuit board layout. The only

exception to this is the OUTPUT (switch) pin, which should

not have large areas of copper (see page 8 ‘PCB Layout

Guideline’).

Thermal Analysis and Design

The following procedure must be performed to determine

whether or not a heatsink will be required. First determine:

1. PD(max) maximum regulator power dissipation in the

application.

2. TA(max) maximum ambient temperature in the

application.

3. TJ(max) maximum allowed junction temperature

(125°C for the LM2596). For a conservative

design, the maximum junction temperature

should not exceed 110°C to assure safe

operation. For every additional +10°C

temperature rise that the junction must

withstand, the estimated operating lifetime

of the component is halved.

4. RqJC package thermal resistance junction−case.

5. RqJA package thermal resistance junction−ambient.

(Refer to Maximum Ratings on page 2 of this data sheet or

RqJC and RqJA values).

The following formula is to calculate the approximate

total power dissipated by the LM2596:

PD = (Vin x IQ) + d x ILoad x Vsat

where d is the duty cycle and for buck converter

d+ton

T+

Vin

IQ(quiescent current) and Vsat can be found in the

LM2596 data sheet,

Vin is minimum input voltage applied,

VOis the regulator output voltage,

ILoad is the load current.

The dynamic switching losses during turn−on and

turn−off can be neglected if proper type catch diode is used.

Packages Not on a Heatsink (Free−Standing)

For a free−standing application when no heatsink is used,

the junction temperature can be determined by the following

expression:

TJ = (RqJA) (PD) + TA

where (RqJA)(PD) represents the junction temperature rise

caused by the dissipated power and TA is the maximum

ambient temperature.

Packages on a Heatsink

If the actual operating junction temperature is greater than

the selected safe operating junction temperature determined

in step 3, than a heatsink is required. The junction

temperature will be calculated as follows:

TJ = PD (RqJA + RqCS + RqSA) + TA

where RqJC is the thermal resistance junction−case,

RqCS is the thermal resistance case−heatsink,

RqSA is the thermal resistance heatsink−ambient.

If the actual operating temperature is greater than the

selected safe operating junction temperature, then a larger

heatsink is required.

Some Aspects That can Influence Thermal Design

It should be noted that the package thermal resistance and

the junction temperature rise numbers are all approximate,

and there are many factors that will affect these numbers,

such as PC board size, shape, thickness, physical position,

location, board temperature, as well as whether the

surrounding air is moving or still.

Other factors are trace width, total printed circuit copper

area, copper thickness, single− or double−sided, multilayer

board, the amount of solder on the board or even color of the

traces.

The size, quantity and spacing of other components on the

board can also influence its effectiveness to dissipate the heat.

Figure 22. Inverting Buck−Boost Develops −12 V

1N5822

33 mH

Feedback

12 to 40 V

Unregulated

DC Input

Cin

100 mF/50 V GND

ON/OFF

+Vin

−12 V @ 0.7 A

Regulated

Output

Cout

220 mF

LM2596−ADJ

LM2596

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ADDITIONAL APPLICATIONS

Inverting Regulator

An inverting buck−boost regulator using the

LM2596−ADJ is shown in Figure 22. This circuit converts

a positive input voltage to a negative output voltage with a

common ground by bootstrapping the regulators ground to

the negative output voltage. By grounding the feedback pin,

the regulator senses the inverted output voltage and

regulates it.

In this example the LM2596−12 is used to generate a

−12 V output. The maximum input voltage in this case

cannot exceed +28 V because the maximum voltage

appearing across the regulator is the absolute sum of the

input and output voltages and this must be limited to a

maximum of 40 V.

This circuit configuration is able to deliver approximately

0.7 A to the output when the input voltage is 12 V or higher.

At lighter loads the minimum input voltage required drops

to approximately 4.7 V, because the buck−boost regulator

topology can produce an output voltage that, in its absolute

value, is either greater or less than the input voltage.

Since the switch currents in this buck−boost configuration

are higher than in the standard buck converter topology, the

available output current is lower.

This type of buck−boost inverting regulator can also

require a larger amount of startup input current, even for

light loads. This may overload an input power source with

a current limit less than 5.0 A.

Such an amount of input startup current is needed for at

least 2.0 ms or more. The actual time depends on the output

voltage and size of the output capacitor.

Because of the relatively high startup currents required by

this inverting regulator topology, the use of a delayed startup

or an undervoltage lockout circuit is recommended.

Using a delayed startup arrangement, the input capacitor

can charge up to a higher voltage before the switch−mode

regulator begins to operate.

The high input current needed for startup is now partially

supplied by the input capacitor Cin.

It has been already mentioned above, that in some

situations, the delayed startup or the undervoltage lockout

features could be very useful. A delayed startup circuit

applied to a buck−boost converter is shown in Figure 27.

Figure 29 in the “Undervoltage Lockout” section describes

an undervoltage lockout feature for the same converter

topology.

Design Recommendations:

The inverting regulator operates in a different manner

than the buck converter and so a different design procedure

has to be used to select the inductor L1 or the output

capacitor Cout.

The output capacitor values must be larger than what is

normally required for buck converter designs. Low input

voltages or high output currents require a large value output

capacitor (in the range of thousands of mF).

The recommended range of inductor values for the

inverting converter design is between 68 mH and 220 mH. To

select an inductor with an appropriate current rating, the

inductor peak current has to be calculated.

The following formula is used to obtain the peak inductor

current:

where ton +

|VO|

Vin )|VO|x1.0

fosc , and fosc +52 kHz.

Ipeak [

ILoad (Vin )|VO|)

Vin )

Vin xt

2L1

Under normal continuous inductor current operating

conditions, the worst case occurs when Vin is minimal.

Figure 23. Inverting Buck−Boost Develops −12 V

1N5822

33 mH

Feedback

12 to 40 V

Unregulated

DC Input

Cin

100 mF/50 V GND

ON/OFF

+Vin

−12 V @ 0.7 A

Regulated

Output

Cout

220 mF

LM2596−ADJ

0.1 mF

47k

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Figure 24. Inverting Buck−Boost Regulator Shutdown

Circuit Using an Optocoupler

LM2596−XX

35GN

ON/OFF

+Vin

47 k

Cin

100 mF

NOTE: This picture does not show the complete circuit.

47 k

470

Shutdown

Input

MOC8101

-Vou

Off

5.0 V

+Vin

With the inverting configuration, the use of the ON/OFF

pin requires some level shifting techniques. This is caused

by the fact, that the ground pin of the converter IC is no

longer at ground. Now, the ON/OFF pin threshold voltage

(1.3 V approximately) has to be related to the negative

output voltage level. There are many different possible shut

down methods, two of them are shown in Figures 24 and 25.

Figure 25. Inverting Buck−Boost Regulator Shutdown

Circuit Using a PNP Transistor

NOTE: This picture does not show the complete circuit.

5.6 k

2N3906

LM2596−XX

35GN

ON/OFF

12 k -Vout

+Vin

Shutdown

Input

Off

+Vin

Cin

100 mF

Negative Boost Regulator

This example is a variation of the buck−boost topology

and it is called negative boost regulator. This regulator

experiences relatively high switch current, especially at low

input voltages. The internal switch current limiting results in

lower output load current capability.

The circuit in Figure 26 shows the negative boost

configuration. The input voltage in this application ranges

from −5.0 V to −12 V and provides a regulated −12 V output.

If the input voltage is greater than −12 V, the output will rise

above −12 V accordingly, but will not damage the regulator.

Figure 26. Negative Boost Regulator

1N5822

33 mH

Feedback

−12 V

Unregulated

DC Input

Cin

100 mF/

50 V GND

ON/OFF

+Vin

−12 V @ 0.7 A

Regulated

Output

LM2596−ADJ

Cout

470 mF

Design Recommendations:

The same design rules as for the previous inverting

buck−boost converter can be applied. The output capacitor

Cout must be chosen larger than would be required for a what

standard buck converter. Low input voltages or high output

currents require a large value output capacitor (in the range

of thousands of mF). The recommended range of inductor

values for the negative boost regulator is the same as for

inverting converter design.

Another important point is that these negative boost

converters cannot provide current limiting load protection in

the event of a short in the output so some other means, such

as a fuse, may be necessary to provide the load protection.

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Delayed Startup

There are some applications, like the inverting regulator

already mentioned above, which require a higher amount of

startup current. In such cases, if the input power source is

limited, this delayed startup feature becomes very useful.

To provide a time delay between the time when the input

voltage is applied and the time when the output voltage

comes up, the circuit in Figure 27 can be used. As the input

voltage is applied, the capacitor C1 charges up, and the

voltage across the resistor R2 falls down. When the voltage

on the ON/OFF pin falls below the threshold value 1.3 V, the

regulator starts up. Resistor R1 is included to limit the

maximum voltage applied to the ON/OFF pin. It reduces the

power supply noise sensitivity, and also limits the capacitor

C1 discharge current, but its use is not mandatory.

When a high 50 Hz or 60 Hz (100 Hz or 120 Hz

respectively) ripple voltage exists, a long delay time can

cause some problems by coupling the ripple into the

ON/OFF pin, the regulator could be switched periodically

on and off with the line (or double) frequency.

Figure 27. Delayed Startup Circuitry

47 k

LM2596−XX

35GN

ON/OFF

47 k

+Vin

0.1 mF

Cin

100 mF

NOTE: This picture does not show the complete circuit.

Undervoltage Lockout

Some applications require the regulator to remain off until

the input voltage reaches a certain threshold level. Figure 28

shows an undervoltage lockout circuit applied to a buck

regulator. A version of this circuit for buck−boost converter

is shown in Figure 29. Resistor R3 pulls the ON/OFF pin

high and keeps the regulator off until the input voltage

reaches a predetermined threshold level with respect to the

ground Pin 3, which is determined by the following

expression:

Vth [VZ1 )ǒ1.0 )R2

R1ǓVBE (Q1)

Figure 28. Undervoltage Lockout Circuit for

Buck Converter

10 k

1N5242B

10 k

2N3904

47 k

Vth ≈ 13 V

Cin

100 mF

LM2596−XX

35GN

ON/OFF

+Vin

NOTE: This picture does not show the complete circuit.

The following formula is used to obtain the peak inductor

current:

where ton +

|VO|

Vin )|VO|x1.0

fosc , and fosc +52 kHz.

Ipeak [

ILoad (Vin )|VO|)

Vin )

Vin xt

2L1

Under normal continuous inductor current operating

conditions, the worst case occurs when Vin is minimal.

Figure 29. Undervoltage Lockout Circuit for

Buck−Boost Converter

15 k

1N5242B

15 k

2N3904

47 k

Vth ≈ 13 V

Cin

100 mF

LM2596−XX

35GN

ON/OFF

+Vin

Vout

NOTE: This picture does not show the complete circuit.

Adjustable Output, Low−Ripple Power Supply

A 3.0 A output current capability power supply that

features an adjustable output voltage is shown in Figure 30.

This regulator delivers 3.0 A into 1.2 V to 35 V output.

The input voltage ranges from roughly 3.0 V to 40 V. In order

to achieve a 10 or more times reduction of output ripple, an

additional L−C filter is included in this circuit.

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Figure 30. 1.2 to 35 V Adjustable 3.0 A Power Supply with Low Output Ripple

1N5822

33 mH

Output

Feedback

50 k

1.21 k

20 mHOutput

Voltage

1.2 to 35 V @ 3.0 A

Optional Output

Ripple Filter

40 V Max

Unregulated

DC Input

Cout

220 mFC1

100 mF

Cin

100 mF

LM2596−Adj

53ON

/OFFGN

+Vin

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THE LM2596 STEP−DOWN VOLTAGE REGULATOR WITH 5.0 V @ 3.0 A OUTPUT POWER CAPABILITY.

TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT

C1 −100 mF, 50 V, Aluminium Electrolytic

C2 −220 mF, 25 V, Aluminium Electrolytic

D1 −3.0 A, 40 V, Schottky Rectifier, 1N5822

L1 −33 mH, DO5040H, Coilcraft

R1 −1.0 kW, 0.25 W

R2 −3.0 kW, 0.25 W

Figure 31. Schematic Diagram of the 5.0 V @ 3.0 A Step−Down Converter Using the LM2596−ADJ

Vref = 1.23 V

R1 is between 1.0 k and 5.0 k

1N5822

33 mH

Output

2R2

3.0 k

1.0 k

Regulated

Output Filtered

Vout2 = 5.0 V @ 3.0 A

Unregulated

DC Input

220 mF

/16 V

100 mF

/50 V

LM2596−ADJ

53ON

/OFFGN

+Vin

+Vin = 10 V to 40 V

4 Feedback

Vout +Vref )ǒ1.0 )R2

R1Ǔ

ON/OFF

Figure 32. Printed Circuit Board Layout

Component Side

Figure 33. Printed Circuit Board Layout

Copper Side

NOTE: Not to scale. NOTE: Not to scale.

CFF

References

•National Semiconductor LM2596 Data Sheet and Application Note

•National Semiconductor LM2595 Data Sheet and Application Note

•Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990

•Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995

LM2596

http://onsemi.com

ORDERING INFORMATION

Device Package Shipping†

LM2596TADJG TO−220

(Pb−Free)

50 Units / Rail

LM2596TVADJG TO−220 (F)

(Pb−Free)

50 Units / Rail

LM2596DSADJG D2PAK

(Pb−Free)

50 Units / Rail

LM2596DSADJR4G D2PAK

(Pb−Free)

800 / Tape & Reel

†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging

Specifications Brochure, BRD8011/D.

A = Assembly Location

WL = Wafer Lot

Y = Year

WW = Work Week

G = Pb−Free Package

TO−220

TV SUFFIX

CASE 314B

MARKING DIAGRAMS

TO−220

T SUFFIX

CASE 314D

D2PAK

DS SUFFIX

CASE 936A

2596T−ADJ

AWLYWWG

2596T−ADJ

AWLYWWG

2596−ADJ

AWLYWWG

LM2596

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PACKAGE DIMENSIONS

TO−220

TV SUFFIX

CASE 314B−05

ISSUE L

−P−

0.10 (0.254) P M

5X J

0.24 (0.610) T

OPTIONAL

CHAMFER

SLW

−T−SEATING

PLANE

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION D DOES NOT INCLUDE

INTERCONNECT BAR (DAMBAR) PROTRUSION.

DIMENSION D INCLUDING PROTRUSION SHALL

NOT EXCEED 0.043 (1.092) MAXIMUM.

DIM MIN MAX MIN MAX

MILLIMETERSINCHES

A0.572 0.613 14.529 15.570

B0.390 0.415 9.906 10.541

C0.170 0.180 4.318 4.572

D0.025 0.038 0.635 0.965

E0.048 0.055 1.219 1.397

F0.850 0.935 21.590 23.749

G0.067 BSC 1.702 BSC

H0.166 BSC 4.216 BSC

J0.015 0.025 0.381 0.635

K0.900 1.100 22.860 27.940

L0.320 0.365 8.128 9.271

N0.320 BSC 8.128 BSC

Q0.140 0.153 3.556 3.886

S--- 0.620 --- 15.748

U0.468 0.505 11.888 12.827

V--- 0.735 --- 18.669

W0.090 0.110 2.286 2.794

5X D

TO−220

T SUFFIX

CASE 314D−04

ISSUE F

−Q−

12345

5 PL

0.356 (0.014) T

SEATING

PLANE

−T−

DIM MIN MAX MIN MAX

MILLIMETERSINCHES

A0.572 0.613 14.529 15.570

B0.390 0.415 9.906 10.541

C0.170 0.180 4.318 4.572

D0.025 0.038 0.635 0.965

E0.048 0.055 1.219 1.397

G0.067 BSC 1.702 BSC

H0.087 0.112 2.210 2.845

J0.015 0.025 0.381 0.635

K0.977 1.045 24.810 26.543

L0.320 0.365 8.128 9.271

Q0.140 0.153 3.556 3.886

U0.105 0.117 2.667 2.972

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION D DOES NOT INCLUDE

INTERCONNECT BAR (DAMBAR) PROTRUSION.

DIMENSION D INCLUDING PROTRUSION SHALL

NOT EXCEED 10.92 (0.043) MAXIMUM.

B1 0.375 0.415 9.525 10.541

DETAIL A-A

DETAIL A−A

LM2596

http://onsemi.com

PACKAGE DIMENSIONS

D2PAK

D2T SUFFIX

CASE 936A−02

ISSUE C

5 REF

123

TERMINAL 6 NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A

AND K.

4. DIMENSIONS U AND V ESTABLISH A MINIMUM

MOUNTING SURFACE FOR TERMINAL 6.

5. DIMENSIONS A AND B DO NOT INCLUDE MOLD

FLASH OR GATE PROTRUSIONS. MOLD FLASH

AND GATE PROTRUSIONS NOT TO EXCEED 0.025

(0.635) MAXIMUM.

DIM

MIN MAX MIN MAX

MILLIMETERS

0.386 0.403 9.804 10.236

INCHES

B0.356 0.368 9.042 9.347

C0.170 0.180 4.318 4.572

D0.026 0.036 0.660 0.914

E0.045 0.055 1.143 1.397

G0.067 BSC 1.702 BSC

H0.539 0.579 13.691 14.707

K0.050 REF 1.270 REF

L0.000 0.010 0.000 0.254

M0.088 0.102 2.235 2.591

N0.018 0.026 0.457 0.660

P0.058 0.078 1.473 1.981

R5 REF

S0.116 REF 2.946 REF

U0.200 MIN 5.080 MIN

V0.250 MIN 6.350 MIN

0.010 (0.254) T

−T−

OPTIONAL

CHAMFER

8.38

0.33

1.016

0.04

16.02

0.63

10.66

0.42

3.05

0.12

1.702

0.067

SCALE 3:1 ǒmm

inchesǓ

*For additional information on our Pb−Free strategy and soldering

details, please download the ON Semiconductor Soldering and

Mounting Techniques Reference Manual, SOLDERRM/D.

SOLDERING FOOTPRINT*

ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice

to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability

arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.

“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All

operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights

nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications

intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should

Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,

and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death

associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal

Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.

LM2596/D

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