DPA423RTL - Power Integrations, Inc.

DPA422-426

DPA-Switch™ Family

www.powerint.com December 2012

Highly Integrated DC-DC Converter ICs for

Power over Ethernet & Telecom Applications

Output Power Table

36-75 VDC Input Range (Forward)2

Total Device

Dissipation3

Product4

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA4227,8 7.5 W 10 W - - - 10 W

DPA423 12 W 16 W - - - 18 W

DPA424 16 W 23 W 35 W - - 35 W

DPA425 23 W 32 W 50 W 62 W - 70 W

DPA426525 W 35 W 55 W 70 W 83 W 100 W

36-75 VDC Input Range (Flyback)2

Total Device

Dissipation3

Product4

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA4227,8 6.5 W 9.0 W - - 9.0 W

DPA423 9 W 13 W - - 13 W

DPA424 10 W 14.5 W 18 W 24 W 26 W

DPA425 -6-6-625.5 W 52 W

Table 1. Output Power Table.

Notes:

1. Maximum output power is limited by device internal current limit.

2. See Applications Considerations section for complete description of assumptions

and for output powers with other input voltage ranges.

3. For device dissipation of 1.5 W or below, use P or G packages. Device

dissipation

above 1.5 W is possible with R package.

4. Packages: P: DIP-8, G: SMD-8, R: TO-263-7C. For lead-free package options,

see Part Ordering Information.

5. Available in R package only.

6. Due to higher switching losses, the DPA425 may not deliver additional power

compared to a smaller device.

7. Available in P and G package only.

8. DPA422 not recommended for new designs, use DPA423 instead.

Product Highlights

Highly Integrated Solution

• Eliminates up to 50 external components–saves space, cost

• Integrates 220 V high frequency MOSFET, PWM control

• Lower cost plastic DIP surface mount (G package) and

through-hole (P package) options for designs ≤35 W

• Thermally efﬁcient TO-263-7C (R package) option for high

power applications

Superior Performance and Flexibility

• Eliminates all external current sensing circuitry

• Built-in auto-restart for output overload/open loop protection

• Pin selectable 300/400 kHz ﬁxed frequency

• Wide input (line) voltage range: 16-75 VDC

• Externally programmable current limit

• Source connected tab reduces EMI

• Line undervoltage (UV) detection: meets ETSI standards

• Line overvoltage (OV) shutdown protection

• UV/OV limits gate drive voltage for synchronous rectiﬁcation

• Fully integrated soft-start for minimum stress/overshoot

• Supports forward or ﬂyback topology

• Cycle skipping: regulation to zero load without pre-load

• Hysteretic thermal shutdown for automatic fault recovery

• RoHS compliant P and G package options

EcoSmart™– Energy Efﬁcient

• Extremely low consumption at no-load

• Cycle skipping at light load for high standby efﬁciency

Applications

• PoE applications, VoIP phones, WLAN, security cameras

• Telco central ofﬁce equipment: xDSL, ISDN, PABX

• Distributed power architectures (24 V/48 V bus)

• Industrial controls

Description

The DPA-Switch IC family is a highly integrated solution for

DC-DC conversion applications with 16-75 VDC input.

DPA-Switch uses the same proven topology as TOPSwitch®,

cost effectively integrating a power MOSFET, PWM control, fault

protection and other control circuitry onto a single CMOS chip.

High performance features are enabled with three user

conﬁgurable pins. Hysteretic thermal shutdown is also provided.

In addition, all critical parameters (i.e. current limit, frequency,

PWM gain) have tight temperature and absolute tolerance, to

simplify design and reduce system cost.

Figure 1. Typical Forward Converter Application.

PI-2770-032002

DPA-Switch

VIN

CONTROL

SENSE

CIRCUIT

RESET/

CLAMP

CIRCUIT

Rev. T 12/12

DPA422-426

www.powerint.com

Section List

Functional Block Diagram ....................................................................................................................................... 3

Pin Functional Description ...................................................................................................................................... 3

DPA-Switch Family Functional Description ............................................................................................................ 4

CONTROL (C) Pin Operation .................................................................................................................................... 4

Oscillator and Switching Frequency .......................................................................................................................... 5

Pulse Width Modulator & Maximum Duty Cycle ........................................................................................................ 5

Minimum Duty Cycle and Cycle Skipping ................................................................................................................. 6

Error Ampliﬁer .......................................................................................................................................................... 6

On-chip Current Limit with External Programmability ................................................................................................ 6

Line Undervoltage Detection (UV) ............................................................................................................................. 6

Line Overvoltage Shutdown (OV) .............................................................................................................................. 6

Line Feed-Forward with DCMAX Reduction ................................................................................................................ 6

Remote ON/OFF ...................................................................................................................................................... 7

Synchronization ........................................................................................................................................................ 7

Soft-Start ................................................................................................................................................................. 8

Shutdown/Auto-Restart ........................................................................................................................................... 8

Hysteretic Over-Temperature Protection ................................................................................................................... 8

Bandgap Reference ................................................................................................................................................. 8

High-Voltage Bias Current Source ............................................................................................................................ 8

Using Feature Pins ..................................................................................................................................................... 8

FREQUENCY (F) Pin Operation ................................................................................................................................ 8

LINE-SENSE (L) Pin Operation ................................................................................................................................. 9

EXTERNAL CURRENT LIMIT (X) Pin Operation ......................................................................................................... 9

Typical Uses of FREQUENCY (F) Pin ...................................................................................................................... 11

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins ....................................................... 11

Application Examples .............................................................................................................................................. 14

Key Application Considerations .............................................................................................................................. 16

DPA-Switch Design Considerations ........................................................................................................................ 16

DPA-Switch Layout Considerations ........................................................................................................................ 18

Quick Design Checklist .......................................................................................................................................... 19

Design Tools .......................................................................................................................................................... 20

Product Speciﬁcations and Test Conditions .......................................................................................................... 21

Typical Performance Characteristics .................................................................................................................. 27

Part Ordering Information ....................................................................................................................................... 30

Package Outlines .................................................................................................................................................... 31

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 2. Functional Block Diagram.

Pin Functional Description

DRAIN (D) Pin:

High-voltage power MOSFET drain output. The internal startup

bias current is drawn from this pin through a switched high-

voltage current source. Internal current limit sense point for

drain current.

CONTROL (C) Pin:

Error ampliﬁer and feedback current input pin for duty cycle

control. Internal shunt regulator connection to provide internal

bias current during normal operation. It is also used as the

connection point for the supply bypass and auto-restart/

compensation capacitor.

LINE-SENSE (L) Pin:

Input pin for overvoltage (OV), undervoltage (UV) lock out, line

feed-forward with the maximum duty cycle (DCMAX) reduction,

remote ON/OFF and synchronization. A connection to

SOURCE pin disables all functions on this pin.

EXTERNAL CURRENT LIMIT (X) Pin:

Input pin for external current limit adjustment and remote

ON/OFF. A connection to SOURCE pin disables all functions

on this pin.

FREQUENCY (F) Pin:

Input pin for selecting switching frequency: 400 kHz if

connected to SOURCE pin and 300 kHz if connected to

CONTROL pin.

SOURCE (S) Pin:

Output MOSFET source connection for the power return.

Primary side control circuit common and reference point.

PI-2760-070501

SHUTDOWN/

AUTO-RESTART

PWM

COMPARATOR

CLOCK

SAW

300/400 kHz

CONTROLLED

TURN-ON

GATE DRIVER

CURRENT LIMIT

COMPARATOR

INTERNAL UV

COMPARATOR

INTERNAL

SUPPLY

5.8 V

4.8 V

SOURCE (S)

DMAX

STOP SOFT-

START

CONTROL (C)

LINE-SENSE (L)

EXTERNAL

CURRENT LIMIT (X)

FREQUENCY (F)

5.8 V

1 V

IFB

LEADING

EDGE

BLANKING

÷ 8

HYSTERETIC

THERMAL

SHUTDOWN

SHUNT REGULATOR/

ERROR AMPLIFIER +

DRAIN (D)

ON/OFF

SOFT

START

DCMAX

VBG

DCMAX

VBG + VT

OV/UV

VI (LIMIT)

CURRENT

LIMIT

ADJUST

LINE

SENSE

SOFT START

CYCLE

SKIPPING

STOP LOGIC

OSCILLATOR

Tab internally connected

to SOURCE pin

(See layout considerations)

R Package

(TO-263-7C)

123457

CLXSFD

P Package (DIP-8)

G Package (SMD-8)

PI-4030-110507

Figure 2. Pin Conﬁguration (Top View).

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 4. Relationship of Duty Cycle to CONTROL Pin Current.

DPA-Switch Family Functional Description

DPA-Switch is an integrated switched mode power supply chip

that converts a current at the control input to a duty cycle at the

open drain output of a high-voltage power MOSFET. During

normal operation the duty cycle of the power MOSFET

decreases linearly with increasing CONTROL pin current as

shown in Figure 4. A patented high-voltage CMOS technology

allows both the high-voltage power MOSFET and all the low

voltage control circuitry to be cost effectively integrated onto a

single monolithic chip.

In addition to the standard TOPSwitch features, such as the

high-voltage start-up, the cycle-by-cycle current limiting, loop

compensation circuitry, auto-restart and thermal shutdown,

DPA-Switch also offers many advanced features that reduce

system cost and increase power supply performance and

design ﬂexibility. Following is a summary of the advanced

features:

1. A fully integrated 5 ms soft-start limits peak currents and

voltages during start-up and reduces or eliminates output

overshoot in most applications.

2. A 75% maximum duty cycle (DCMAX) together with the line

feed-forward with DCMAX reduction feature makes

DPA-Switch well suited for both ﬂyback and forward

topologies.

3. High switching frequency (400 kHz/300 kHz, pin selectable)

allows the use of smaller size transformers and offers high

bandwidth for power supply control loop.

4. Cycle skipping operation at light load minimizes standby

power consumption (typically <10 mA input current).

5. Line undervoltage ensures glitch free operations at both

power-up and power-down and is tightly toleranced over

process and temperature to meet system level requirements

common in DC to DC converters (e.g. ETSI).

6. Line overvoltage protects DPA-Switch against excessive

input voltage and line surge.

7. External current limit adjustment allows the setting of the

current limit externally to a lower level near the operating

peak current and, if desired, further adjusts the level gradu-

ally as line voltage rises. This makes possible an ideal

implementation of overload power limiting.

8. Synchronization function allows the synchronization of

DPA-Switch operation to an external lower frequency.

9. Remote ON/OFF feature permits DPA-Switch based power

supplies to be easily switched on/off using logic signals.

Maximum input current consumption is 2 mA in remote OFF.

10. Hysteretic over-temperature shutdown provides automatic

recovery from thermal fault.

11. Tight absolute tolerances and small temperature variations

on switching frequency, current limit, and undervoltage lock

out threshold (UV).

Three pins, LINE-SENSE (L), EXTERNAL CURRENT LIMIT (X)

and FREQUENCY (F), are used to implement all the pin -

controllable features. A resistor from the LINE-SENSE pin to DC

input bus implements line UV, line OV and line feed-forward with

DCMAX reduction. A resistor from the EXTERNAL CURRENT

LIMIT pin to the SOURCE pin sets current limit externally. In

addition, remote ON/OFF may be implemented through either

the LINE-SENSE pin or the EXTERNAL CURRENT LIMIT pin

depending on the polarity of the logic signal available as well as

other system speciﬁc considerations. Shorting both the LINE-

SENSE and the EXTERNAL CURRENT LIMIT pins to the

SOURCE pin disables line OV, line UV, line feed-forward with

DCMAX reduction, external current limit, remote ON/OFF and

synchronization. The FREQUENCY pin sets the switching

frequency to 400 kHz if connected to the SOURCE pin, or

300 kHz if connected to the CONTROL pin. This pin should not

be left open. Please refer to “Using Feature Pins” section for

detailed information regarding the proper use of those pins.

CONTROL (C) Pin Operation

The CONTROL pin is a low impedance node that is capable of

receiving a combined supply and feedback current. During

normal operation, a shunt regulator is used to separate the

feedback signal from the supply current. CONTROL pin voltage

VC is the supply voltage for the control circuitry including the

MOSFET gate driver. An external bypass capacitor closely

connected between the CONTROL and SOURCE pins is

required to supply the instantaneous gate drive current. The

total amount of capacitance connected to this pin also sets the

auto-restart timing as well as control loop compensation.

When the DC input voltage is applied to the DRAIN pin during

start-up, the MOSFET is initially off, and the CONTROL pin

capacitor is charged through the switched high-voltage current

source connected internally between the DRAIN and

CONTROL pins. When the CONTROL pin voltage VC reaches

approximately 5.8 V, the control circuitry is activated and the

soft-start begins. The soft-start circuit gradually increases the

duty cycle of the MOSFET from zero to the maximum value

over approximately 5 ms. The high-voltage current source is

turned off at the end of the soft-start. If no external feedback/

supply current is fed into the CONTROL pin by the end of the

soft-start, the CONTROL pin will start discharging in response

to the supply current drawn by the control circuitry and the gate

current of the switching MOSFET driver. If the power supply is

designed properly, and no fault condition such as open loop or

PI-2761-112102

Duty Cycle (%)

IC (mA)

IL = 115 µA

IL < IL(DC) IC (SKIP)

Slope = PWM Gain

ICD1 IB

Auto-restart

Rev. T 12/12

DPA422-426

www.powerint.com

overloaded output exists, the feedback loop will close, providing

external CONTROL pin current, before the CONTROL pin

voltage has had a chance to discharge to the lower threshold

voltage of approximately 4.8 V (internal supply undervoltage

lockout threshold). When the externally fed current charges the

CONTROL pin to the shunt regulator voltage of 5.8 V, current in

excess of the consumption of the chip is shunted to SOURCE

through resistor RE as shown in Figure 2. This current ﬂowing

through RE controls the duty cycle of the power MOSFET to

provide closed loop regulation. The shunt regulator has a ﬁnite

low output impedance ZC that sets the gain of the error ampliﬁer

when used in a primary feedback conﬁguration. The dynamic

impedance ZC of the CONTROL pin together with the external

CONTROL pin capacitance sets the dominant pole for the

control loop.

When a fault condition such as an open loop or overloaded

output prevents the ﬂow of an external current into the CONTROL

pin, the capacitor on the CONTROL pin discharges towards 4.8

V. At 4.8 V auto-restart is activated which turns the output

MOSFET off and puts the control circuitry in a low current

standby mode. The high-voltage current source turns on and

charges the external capacitance again. A hysteretic internal

supply undervoltage comparator keeps VC within a window of

typically 4.8 V to 5.8 V by turning the high-voltage current

source on and off as shown in Figure 5. The auto-restart circuit

has a divide-by-8 counter that prevents the output MOSFET

from turning on again until eight discharge/charge cycles have

elapsed. This is accomplished by enabling the output MOSFET

only when the divide-by-8 counter reaches full count (S7). The

counter effectively limits DPA-Switch power dissipation as well

as the maximum power delivered to the power supply output by

reducing the auto-restart duty cycle to typically 4%. Auto-

restart mode continues until output voltage regulation is again

achieved through closure of the feedback loop.

Oscillator and Switching Frequency

The internal oscillator linearly charges and discharges an

internal capacitance between two voltage levels to create a

sawtooth waveform for the pulse width modulator. The

oscillator sets both the pulse width modulator latch and the

current limit latch at the beginning of each cycle.

The nominal switching frequency of 400 kHz was chosen to

minimize the transformer size and to allow faster power supply

loop response. The FREQUENCY pin, when shorted to the

CONTROL pin, lowers the switching frequency to 300 kHz,

which may be preferable in some applications such as those

employing secondary synchronous rectiﬁcation. Otherwise, the

FREQUENCY pin should be connected to the SOURCE pin for

the default 400 kHz.

Pulse Width Modulator and Maximum Duty Cycle

The pulse width modulator implements voltage mode control by

driving the output MOSFET with a duty cycle inversely

proportional to the current into the CONTROL pin that is in

excess of the internal supply current of the chip (see Figure 4).

The excess current is the feedback error signal that appears

across RE (see Figure 2). This signal is ﬁltered by an RC network

with a typical corner frequency of 30 kHz to reduce the effect of

switching noise in the chip supply current generated by the

MOSFET gate driver. The ﬁltered error signal is compared with

the internal oscillator sawtooth waveform to generate the duty

cycle waveform. As the control current increases, the duty cycle

decreases. A clock signal from the oscillator sets a latch that

turns on the output MOSFET. The pulse width modulator resets

the latch, turning off the output MOSFET. Note that a minimum

current must be driven into the CONTROL pin before the duty

cycle begins to change.

PI-3867-050602

S1 S2 S6 S7 S1 S2 S6 S7S0 S1 S7

S0 S0 5.8 V

4.8 V

0 V

VLINE

VDRAIN

VOUT

Note: S0 through S7 are the output states of the auto-restart counter

1234

0 V

S6 S7

VUV

Figure 5. Typical Waveforms for (1) Power Up, (2) Normal Operation, (3) Auto-restart and (4) Power Down.

Rev. T 12/12

DPA422-426

www.powerint.com

The maximum duty cycle, DCMAX is set at a default maximum

value of 75% (typical). However, by connecting the LINE-SENSE

to the DC input bus through a resistor with appropriate value,

the maximum duty cycle can be made to decrease from 75% to

33% (typical) as shown in Figure 7 when input line voltage

increases (see Line Feed-Forward with DCMAX Reduction).

Minimum Duty Cycle and Cycle Skipping

To maintain power supply output regulation, the pulse width

modulator reduces duty cycle as the load at the power supply

output decreases. This reduction in duty cycle is proportional to

the current ﬂowing into the CONTROL pin. As the CONTROL

pin current increases, the duty cycle reduces linearly towards a

minimum value speciﬁed as minimum duty cycle, DCMIN. After

reaching DCMIN, if CONTROL pin current is increased further by

approximately 2 mA, the pulse width modulator will force the

duty cycle from DCMIN to zero in a discrete step (refer to Figure

4). This feature allows a power supply to operate in a cycle

skipping mode when the load consumes less power than the

DPA-Switch delivers at minimum duty cycle, DCMIN. No

additional control is needed for the transition between normal

operation and cycle skipping. As the load increases or

decreases, the power supply automatically switches between

normal and cycle skipping mode as necessary.

Cycle skipping may be avoided, if so desired, by connecting a

minimum load at the power supply output such that the duty

cycle remains at a level higher than DCMIN at all times.

Error Ampliﬁer

The shunt regulator can also perform the function of an error

ampliﬁer in primary side feedback applications. The shunt

regulator voltage is accurately derived from a temperature-

compensated bandgap reference. The gain of the error ampliﬁer

is set by the CONTROL pin dynamic impedance. The

CONTROL pin clamps external circuit signals to the VC voltage

level. The CONTROL pin current in excess of the supply current

is separated by the shunt regulator and ﬂows through RE as a

voltage error signal.

On-chip Current Limit with External Programmability

The cycle-by-cycle peak drain current limit circuit uses the

output MOSFET ON-resistance as a sense resistor. A current

limit comparator compares the output MOSFET on-state drain

to source voltage, VDS(ON) with a threshold voltage. At the current

limit, VDS(ON) exceeds the threshold voltage and the MOSFET is

turned off until the start of the next clock cycle. The current limit

comparator threshold voltage is temperature compensated to

minimize the variation of the current limit due to temperature

related changes in RDS(ON) of the output MOSFET. The default

current limit of DPA-Switch is preset internally. However, with a

resistor connected between EXTERNAL CURRENT LIMIT pin

and SOURCE pin, the current limit can be programmed

externally to a lower level between 25% and 100% of the default

current limit. Please refer to the graphs in the Typical

Performance Characteristics section for the selection of the

resistor value. By setting current limit low, a larger DPA-Switch

than necessary for the power required can be used to take

advantage of the lower RDS(ON) for higher efﬁciency/smaller heat

sinking requirements. With a second resistor connected

between the EXTERNAL CURRENT LIMIT pin and the DC input

bus, the current limit is reduced with increasing line voltage,

allowing a true power limiting operation against line variation to

be implemented in a ﬂyback conﬁguration.

The leading edge blanking circuit inhibits the current limit

comparator for a short time after the output MOSFET is turned

on. The leading edge blanking time has been set so that, if a

power supply is designed properly, current spikes caused by

primary-side capacitance and secondary-side rectiﬁer reverse

recovery time should not cause premature termination of the

switching pulse.

The current limit after the leading edge blanking time is as

shown in Figure 31. To avoid triggering the current limit in

normal operation, the drain current waveform should stay within

the envelope shown.

Line Undervoltage Detection (UV)

At power up, UV keeps DPA-Switch off until the input line

voltage reaches the under voltage upper threshold. At power

down, UV holds DPA-Switch on until the input voltage falls

below the under voltage lower threshold. A single resistor

connected from the LINE-SENSE pin to the DC input bus sets

UV upper and lower thresholds. To avoid false triggering by

noise, a hysteresis is implemented which sets the UV lower

threshold typically at 94% of the UV upper threshold. If the UV

lower threshold is reached during operation without the power

supply losing regulation and the condition stays longer than

10 µs (typical), the device will turn off and stay off until the UV

upper threshold has been reached again. Then, a soft-start will

be initiated the next time CONTROL pin voltage reaches 5.8.V. If

the power supply loses regulation before reaching the UV lower

threshold, the device will enter auto-restart. At the end of each

auto-restart cycle (S7), the UV comparator is enabled. If the UV

upper threshold is not exceeded, the MOSFET will be disabled

during the next cycle (see Figure 5). The UV feature can be

disabled independent of OV feature.

Line Overvoltage Shutdown (OV)

The same resistor used for UV also sets an overvoltage

threshold which, once exceeded, will force the DPA-Switch

output into the off-state within one switching cycle. The ratio of

OV and UV thresholds is preset at 2.7 as can be seen in Figure 7.

When the MOSFET is off, the input voltage surge capability is

increased to the voltage rating of the MOSFET (220 V), due to

the absence of the reﬂected voltage and leakage spikes on the

drain. A small amount of hysteresis is provided on the OV

threshold to prevent noise triggering. The OV feature can be

disabled independent of the UV feature as shown in Figure 13.

Line Feed-Forward with DCMAX Reduction

The same resistor used for UV and OV also implements line

voltage feed-forward that minimizes output line ripple and

reduces power supply output sensitivity to line transients. This

feed-forward operation is illustrated in Figure 4 by the different

values of IL. Note that for the same CONTROL pin current,

higher line voltage results in smaller operating duty cycle. As an

added feature, the maximum duty cycle DCMAX is also reduced

from 75% (typical) at a voltage slightly higher than the UV

threshold to 33% (typical) at the OV threshold (see Figures 4, 7).

Limiting DCMAX at higher line voltages helps prevent transformer

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 6. Synchronization Timing Diagram.

saturation due to large load transients in forward converter

applications. DCMAX of 33% at the OV threshold was chosen to

ensure that the power capability of the DPA-Switch is not

restricted by this feature under normal operation.

Remote ON/OFF

Remote ON/OFF control describes operation where the IC is

turned on or off for long periods as opposed to the cycle-by-

cycle on/off control, which is described in the Synchronization

section below.

DPA-Switch can be turned on or off by controlling the current

into the LINE-SENSE pin or out from the EXTERNAL CURRENT

LIMIT pin (see Figure 7). This allows easy implementation of

remote ON/OFF control of DPA-Switch in several different ways.

A transistor or an optocoupler output connected between the

EXTERNAL CURRENT LIMIT pin and the SOURCE pin

implements this function with “active-on” (Figures 17, 19 and 21)

while a transistor or an optocoupler output connected between

the LINE-SENSE pin and the CONTROL pin implements the

function with “active-off” (Figures 18, 20 and 22).

When a signal is received at the LINE-SENSE pin or the

EXTERNAL CURRENT LIMIT pin to disable the output through

any of the pin functions such as OV, UV and remote ON/OFF,

DPA-Switch always completes its current switching cycle before

the output is forced off. The internal oscillator is stopped at the

end of the current cycle and stays there as long as the disable

signal exists. When the signal at the above pins changes state

from disable to enable, the internal oscillator starts the next

switching cycle.

The remote ON/OFF feature can be used as a standby or power

switch to turn off the DPA-Switch and keep it in a very low

power consumption state for indeﬁnitely long periods. If the

DPA-Switch is held in remote-off state for longer than 10 µs

(typical), the CONTROL pin goes into the hysteretic mode of

operation. In this mode, the CONTROL pin goes through

alternate charge and discharge cycles between 4.8 V and 5.8 V

(see CONTROL Pin Operation section above) and the IC runs

entirely off the high-voltage DC input, but with very low power

consumption (30 mW typical at 48 V with LINE-SENSE and

EXTERNAL CURRENT LIMIT pins open). When the DPA-Switch

is remotely turned on after entering this mode, it will initiate a

normal start-up sequence with soft-start the next time the

CONTROL pin reaches 5.8 V. In the worst case, the delay from

remote on to start-up can be equal to the full discharge/charge

cycle time of the CONTROL pin, which is approximately 36 ms

for a 22 µF CONTROL pin capacitor. This reduced-consumption

remote-off mode can eliminate expensive and unreliable in-line

mechanical switches. It also allows for microprocessor-

controlled turn-on and turn-off sequences that may be required

in certain applications.

Synchronization

In addition to sensing incoming current for OV, UV and remote

ON/OFF, the LINE-SENSE pin also monitors its pin voltage

through a 1 V threshold comparator. A pin voltage below 1 V

turns on DPA-Switch. When the voltage at LINE-SENSE pin

rises beyond 1 V to disable the output, DPA-Switch completes

its current switching cycle before the output is forced off (similar

to remote ON/OFF operation). The internal oscillator is stopped

at the end of the current cycle awaiting the LINE-SENSE pin

voltage to go low to start the next cycle. This allows the use of

the 1 V threshold to synchronize DPA-Switch to an external

signal with a frequency lower than its internal switching

frequency. A transistor or an optocoupler output connected

between the LINE-SENSE pin and the SOURCE pin implements

this function (see Figure 24). Please refer to Figure 6 for the

timing waveforms of synchronization operation.

PI-2762-070501

Oscillator

(SAW)

DMAX

2 V

0 V

tON

fSYNC ≥ 128 kHz; tOFF £ 7.7 µs; 120 ns £ tON £ 2250 ns for fOSC = 400 kHz

£ 3080 ns for fOSC = 300 kHz

tOFF

ON OFF SYNC

Time

Rev. T 12/12

DPA422-426

www.powerint.com

Table 2. Typical LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Conﬁgurations.

In order to be recognized as a synchronization pulse, the

LINE-SENSE pin needs to stay low (on-time) for at least

120 ns but no more than 2250 ns for 400 kHz (or 3080 ns for

300 kHz) internal switching frequency. In addition, the off-time

must be kept below 7.7 µs, which is a limitation set by the

lowest synchronization frequency of 128 kHz allowed by the

chip. The effective DCMAX for synchronization operation can be

calculated as 0.75 × fSYNC/fOSC. An off-time longer than

7.7 µs may force the CONTROL pin to go into the hysteretic

mode and initiate a soft-start cycle at next turn-on.

Soft-Start

Two on-chip soft-start functions are activated at start-up with a

duration of 5 ms (typical). Maximum duty cycle starts from 0%

and linearly increases to the default maximum of 75% at the

end of the 5 ms duration and the current limit starts from about

85% and linearly increases to 100% at the end of the

5 ms duration. In addition to start-up, soft-start is also activated

at each restart attempt during auto-restart and when restarting

after being in hysteretic regulation of CONTROL pin voltage (VC),

due to remote off or thermal shutdown conditions. This

effectively minimizes current and voltage stresses on the output

MOSFET, the clamp circuit and the output rectiﬁer during start-

up. This feature also helps minimize output overshoot and

prevents saturation of the transformer during start-up.

Shutdown/Auto-Restart

To minimize DPA-Switch power dissipation under fault

conditions, the shutdown/auto-restart circuit turns the power

supply on and off at an auto-restart duty cycle of typically 4% if

an out of regulation condition persists. Loss of regulation

interrupts the external current into the CONTROL pin. VC

regulation changes from shunt mode to the hysteretic auto-

restart mode as described in CONTROL pin operation section.

When the fault condition is removed, the power supply output

becomes regulated, VC regulation returns to shunt mode, and

normal operation of the power supply resumes.

Hysteretic Over-Temperature Protection

Over temperature protection is provided by a precision analog

circuit that turns the output MOSFET off when the junction

temperature exceeds the thermal shutdown temperature

(137 °C typical). When the junction temperature cools to below

the hysteretic temperature (110 °C typical), normal operation

resumes providing automatic recovery. VC is regulated in

hysteretic mode and a 4.8 V to 5.8 V (typical) sawtooth

waveform is present on the CONTROL pin while in thermal

shutdown.

Bandgap Reference

All critical DPA-Switch internal voltages are derived from a

temperature-compensated bandgap reference. This reference

is also used to generate a temperature-compensated current

reference that is trimmed to accurately set the switching

frequency, current limit, and the line OV/UV thresholds.

DPA-Switch has improved circuitry to maintain all of the above

critical parameters within very tight absolute and temperature

tolerances.

High-Voltage Bias Current Source

This current source biases DPA-Switch from the DRAIN pin and

charges the CONTROL pin external capacitance during start-up

or hysteretic operation. Hysteretic operation occurs during

auto-restart, remote off and over-temperature shutdown. In this

mode of operation, the current source is switched on and off

with an effective duty cycle of approximately 20%. This duty

cycle is determined by the ratio of CONTROL pin charge (IC(CH))

and discharge currents (ICD1 and ICD2). This current source is

turned off during normal operation when the output MOSFET is

switching. The effect of the current source switching may be

seen on the DRAIN voltage waveform as small disturbances,

which is normal.

Using Feature Pins

FREQUENCY (F) Pin Operation

The FREQUENCY pin is a digital input pin. Shorting the

FREQUENCY pin to SOURCE pin selects the nominal switching

frequency of 400 kHz (Figure 9) which is suited for most

applications. For other applications that may beneﬁt from lower

switching frequency, a 300 kHz switching frequency can be

selected by shorting the FREQUENCY pin to the CONTROL pin

(Figure 10). This pin should not be left open.

Line-Sense and External Current Limit Pin Table*

Figure Number ➧11 12 13 14 15 16 17 18 19 20 21 22 23 24

Three Terminal Operation ✓

Undervoltage ✓✓ ✓✓✓

Overvoltage ✓ ✓ ✓ ✓ ✓

Line Feed-Forward (DCMAX) ✓ ✓ ✓ ✓

Overload Power Limiting ✓

External Current Limit ✓ ✓ ✓ ✓ ✓ ✓

Remote ON/OFF ✓✓✓✓✓✓

Synchronization ✓

*This table is only a partial list of many LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Conﬁgurations that are possible.

Rev. T 12/12

DPA422-426

www.powerint.com

LINE-SENSE (L) Pin Operation

When current is fed into the LINE-SENSE pin, it works as a

voltage source of approximately 2.6 V up to a maximum current

of +240 µA (typical). At +240 µA, this pin turns into a constant

current sink. Refer to Figure 8. In addition, a comparator with a

threshold of 1 V is connected at the pin and is used to detect

when the pin is shorted to the SOURCE pin.

There are a total of ﬁve functions available through the use of

the LINE-SENSE pin: OV, UV, line feed-forward with DCMAX

reduction, remote ON/OFF and synchronization. Shorting the

LINE-SENSE pin to the SOURCE pin disables all ﬁve functions.

The LINE-SENSE pin is typically used for line sensing by

connecting a resistor from this pin to the positive input DC

voltage bus to implement OV, UV and line feed-forward with

DCMAX reduction over line voltage. In this mode, the value of the

resistor determines the line OV/UV thresholds, and the DCMAX is

reduced linearly with input DC high-voltage starting from just

above the UV threshold. This pin can also be used as the input

pin for remote ON/OFF and synchronization. An external

transistor placed between the LINE-SENSE pin and the

CONTROL pin realizes remote ON/OFF via UV or OV threshold.

Synchronization is available by connecting an open drain

external MOSFET between the LINE-SENSE pin and the

SOURCE pin to generate synchronization pulse. Each time the

MOSFET turns on, the falling edge of the LINE-SENSE pin

voltage initiates a new switching cycle. The lowest

synchronization frequency guaranteed by DPA-Switch is

128 kHz. Refer to Table 2 for possible combinations of the

functions with example circuits shown in Figure 11 through

Figure 24. A description of speciﬁc functions in terms of the

LINE-SENSE pin I/V characteristic is shown in Figure 7 (right

hand side). The horizontal axis represents LINE-SENSE pin

current with positive polarity indicating currents ﬂowing into the

pin. The meaning of the vertical axes varies with functions. For

those that control the on/off states of the output such as UV, OV

and remote ON/OFF, the vertical axis represents the enable/

disable states of the output. UV triggers at IUV (+50 µA typical

with 4 µA hysteresis) and OV triggers at IOV (+13 5 µA typical with

4 µA hysteresis). Between the UV and OV thresholds, the

output is enabled. For line feed-forward with DCMAX reduction,

the vertical axis represents the magnitude of the DCMAX Line

feed-forward with DCMAX reduction lowers maximum duty cycle

from 75% at IL(DC) (+55 µA typical) to 33% at IOV (+13 5 µA).

EXTERNAL CURRENT LIMIT (X) Pin Operation

When current is drawn out of the EXTERNAL CURRENT LIMIT pin,

it works as a voltage source of approximately 1.3 V up to a

maximum current of -230 µA (typical). At -230 µA, it turns into a

constant current source (refer to Figure 8).

There are two functions available through the use of the

EXTERNAL CURRENT LIMIT pin: external current limit and

remote ON/OFF. Shorting the EXTERNAL CURRENT LIMIT pin

and SOURCE pin disables both functions. In high efﬁciency

applications, this pin can be used to reduce the current limit

externally to a value close to the operating peak current, by

connecting the pin to the SOURCE pin through a resistor. The

pin can also be used as a remote ON/OFF control input.

Table 2 shows several different ways of using this pin. See

Figure 7 for a description of the functions where the horizontal

axis (left hand side) represents the EXTERNAL CURRENT LIMIT

pin current. The meaning of the vertical axes varies with

function. For those that control the on/off states of the output

such as remote ON/OFF, the vertical axis represents the enable/

disable states of the output. For external current limit, the

vertical axis represents the magnitude of the ILIMIT. Please see

graphs in the Typical Performance Characteristics section for

the current limit programming range and the selection of the

appropriate resistor value.

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 7. LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Characteristics.

-250 -200 -150 -100 -50 50 100 150 200 2500 0

PI-2778-080801

Output

MOSFET

Switching

(Enabled)

(Disabled)

(Enabled)

(Disabled)

ILIMIT (Default)

Current

Limit

VBG

-21.5 µA

-25.5 µA

VBG + VTP

IUV(U)

IREM(U) IOV(U)

X Pin Voltage

Output

MOSFET

Switching

Maximum

Duty Cycle

L Pin Voltage

DCMAX (75%)

Note: These gures provide idealized functional characteristics with typical performance values. Please refer to the

parametric table and typical performance characteristics sections of the data sheet for measured data.

X Pin Current (µA) L Pin Current (µA)

131 µA

135 µA

47 µA

VBG + VT

1 V

VBG

230 ∝A

240 ∝A

CONTROL (C)

(Voltage Sense)

(Positive Current Sense - Under-Voltage,

Overvoltage, ON/OFF Maximum Duty

Cycle Reduction)

(Negative Current Sense - ON/OFF,

Current Limit Adjustment)

PI-2765-061704

DPA-Switch

LINE-SENSE (L)

EXTERNAL CURRENT LIMIT (X)

Figure 8. LINE-SENSE (L), and EXTERNAL CURRENT LIMIT (X) Pin Input Simpliﬁed Schematic

Rev. T 12/12

DPA422-426

www.powerint.com

Typical Uses of FREQUENCY (F) Pin

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins

Figure 9. 400 kHz Frequency Operation.

PI-2654-071700

Input

Voltage

CONTROL

X F

PI-2766-070901

Input

Voltage

CONTROL

LC S D

C L X S F D

Figure 11. Three Terminal Operation (LINE-SENSE and EXTERNAL CURRENT

LIMIT Features Disabled. FREQUENCY Pin can be tied to SOURCE or

CONTROL Pin).

Figure 13. Line-Sensing for Undervoltage Only (Overvoltage Disabled).

PI-2852-091302

Input

Voltage

VUV = RLS x IUV +

VL (I

L = IUV)

For Values Shown

VUV = 33.3 V

RLS

6.2 V

576 k

42.2 k

CONTROL

Figure 14. Line-Sensing for Overvoltage Only (Undervoltage Disabled).

Maximum Duty Cycle will be reduced at Low Line.

PI-2853-091302

Input

Voltage

CONTROL

1N4148

= I

L (I

= I

)

For Values Shown

= 86.2 V

590 k

30 k

Figure 12. Line-Sensing for Undervoltage, Overvoltage and Line Feed-forward.

PI-2767-091302

Input

Voltage

CONTROL

619 k

VUV = IUV x RLS + VL (I

= I

)

VOV = IOV x RLS + VL (I

= I

)

For RLS = 619 k

VUV = 33.3 V

VOV = 86.0 V

Figure 10. 300 kHz Frequency Operation.

PI-2655-071700

Input

Voltage

CONTROL

Rev. T 12/12

DPA422-426

www.powerint.com

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)

Figure 15. Externally Set Current Limit. Figure 16. Current Limit Reduction with Line Voltage.

PI-2836-050602

Input

Voltage

For R

= 12 k

LIMIT

= 64%

See Figure 32 for other

resistor values (R

)

For R

= 25 k

LIMIT

= 34%

CONTROL

PI-2854-050602

Input

Voltage

363 k

4.2 k

100% @ 36 VDC

64% @ 72 VDC

LIMIT

CONTROL

Figure 17. Active-on (Fail Safe) Remote ON/OFF. Figure 18. Active-off Remote ON/OFF. Maximum Duty Cycle Will Be Reduced.

PI-2855-050602

Input

Voltage

CONTROL

47 k

37.4 k

can be an

optocoupler output or

can be replaced

by a manual switch.

ON/OFF

PI-2625-040501

Input

Voltage

ON/OFF

47 K

can be an optocoupler

output or can be replaced by

a manual switch.

CONTROL

Figure 19. Active-on Remote ON/OFF with Externally Set Current Limit. Figure 20. Active-off Remote ON/OFF with Externally Set Current Limit.

ON/OFF

Input

Voltage

CONTROL

PI-2857-050602

Input

Voltage

CONTROL

47 k

37.4 k

can be an

optocoupler output

or can be replaced

by a manual switch.

ON/OFF

For R

= 12 k

LIMIT

= 64%

For R

= 25 k

LIMIT

= 34%

Rev. T 12/12

DPA422-426

www.powerint.com

Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)

Input

Voltage

CONTROL

ON/OFF

47 k

PI-2859-050602

Input

Voltage

CONTROL

619 k

MAX

@36 V = 75%

MAX

@72 V = 42%

For R

= 12 k

LIMIT

= 64%

can be an optocoupler

output or can be replaced

by a manual switch.

Figure 21. Active-on Remote ON/OFF with LINE-SENSE and

EXTERNAL CURRENT LIMIT.

Figure 22. Active-off Remote ON/OFF with LINE-SENSE.

Figure 23. Line-Sensing and Externally Set Current Limit.

PI-3868-050602

Input

Voltage

CONTROL

ON/OFF

47 kΩ

can be an optocoupler

output.

For timing requirements,

see Figure 6.

Figure 24. Synchronization.

PI-2837-050602

Input

Voltage

CONTROL

12 k

619 K

For R

= 619 k

MAX

@36 V = 75%

MAX

@72 V = 42%

For R

= 12 k

LIMIT

= 64%

See Figure 32 for other

resistor values (R

)

to select different I

LIMIT

values

= 33.3 V

= 86.0 V

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 25. A High Efﬁciency 30 W, 5 V, Telecom Input DC-DC Converter.

DPA425R

BAV

19WS

220 nF

VR1

SMBJ

150 C6

68 µF

10 V

1 nF

1.5 kV R14

10 Ω

18.2 kΩ

619 kΩ

1.0 Ω

C10

100 µF

10 V

C11

100 µF

10 V

C12

1 µF

10 V

36-75 VDC

1 µH

2.5 A L2

C1, C2 & C3

1 µF

100 V

SXF

CONTROL

PC357N1T

PI-3472-040903

DPA-Switch

5 V, 6

RTN

Si4888

R15

10 ΩR16

10 kΩ

BAV19WS

R17

10 ΩQ1

Si4888

4.7 µF

20 V U2

BAV19WS

LM431AIM3

220 Ω

R11

10.0 kΩ

R10

10.0 kΩ

R12

5.1 Ω

10 kΩ

C14

1 µF

C13

10 µF

10 V

C16

100 nF

150 Ω

C17

3300 pF

High Efﬁciency 30 W Forward Converter

The circuit shown in Figure 25 is a typical implementation of a

single output DC-DC converter using DPA-Switch in a forward

conﬁguration with synchronous rectiﬁcation. This design

delivers 30 W at 5 V, from a 36 VDC to 75 VDC input with a

nominal efﬁciency at 48 VDC of 90% using the DPA425R.

By taking advantage of many of the built-in features of the

DPA-Switch, the design is greatly simpliﬁed compared to a

discrete implementation. Resistor R1 programs the input under-

voltage and overvoltage thresholds to typically 33 V and 86 V

respectively. This resistor also linearly reduces the maximum

duty from the internal maximum of 75% at 36 V to 42% at

72 V to prevent core saturation during load transients at high

input voltages. The DPA-Switch internal thresholds are

toleranced and characterized so the designer can guarantee

the converter will begin operation at 36 V, necessary to meet

ETSI standards, without the cost of an external reference IC.

The current limit is externally set by resistor R3 to just above the

drain current level needed for maximum load regulation to limit

the maximum overload power of the converter. The externally

programmable current limit feature also allows a larger

DPA-Switch family member to be selected. Using the X pin, the

current limit can be adjusted to the same level. A large device

reduces conduction losses and improves efﬁciency without

requiring any other circuit changes. This has been used here to

replace the DPA424R with a DPA425R.

Application Examples

The selectable 300/400 kHz switching frequency is set to 300 kHz

by connecting the FREQUENCY (F) pin to CONTROL (C).

DRAIN voltage clamping is provided by VR1, which keeps the

peak DRAIN voltage within acceptable limits. Transformer core

reset is provided by the gate capacitance of Q1 with R17 in

series. Optional reset capacitance C9 with R5 can be added if

necessary to supplement the gate capacitance of Q1.

The output of the transformer is rectiﬁed using MOSFETs to

provide synchronous rectiﬁcation. The UV/OV function, together

with the turns ratio of the transformer, deﬁnes the maximum

MOSFET gate voltage, allowing the very simple gate drive

arrangement, without the need for drive windings or a drive IC.

During primary on-time, capacitor C17 couples charge through

resistor R15 to drive the gate of the forward MOSFET, Q2.

Capacitor C17 provides a DC isolated drive for Q2, preventing

gate overstress on Q1 during power down. The time constant

formed by R16 and C17 is selected to be much longer than one

switching cycle. Diode D4 resets the voltage on capacitor C17

before the next switching cycle. During the primary off-time, the

diode D2 provides a conduction path for the energy in inductor

L2 while Q1 is still off. The transformer reset voltage on the

secondary winding directly drives a positive voltage on the gate

of catch MOSFET, Q1. MOSFET Q1 provides a low loss

conduction path for a substantial portion of the primary off-time.

An isolated auxiliary winding on L2, rectiﬁed and ﬁltered by D1

and C4, provides the bias supply for the optocoupler transistor.

Rev. T 12/12

DPA422-426

www.powerint.com

CONTROL

DPA-Switch

22 µF

10 V

0.1 µF

50 V C10

0.33 µF

C11

0.1 µF

1 µF

50 V

1 µF

10 V

330 µF

6 V

1 µH, 2A

330 µF

6 V

330 µF

6 V

1 µF

100 V

47 pF

200 V

VR1

SMAJ

150A

1 MΩ

8.66 kΩ

5.1 Ω

20 kΩ

34 kΩ

1 kΩ

100 Ω

51 Ω

619 kΩ

9, 101

6, 7

DPA423G U2

PC357

CAT431L,

SOT23

BAV19, SOD323

SL43

+VIN

36 - 57 VDC

J1-1

3.3 V, 2

J2-1

-VIN

J1-2

RTN

J2-2

PI-3806-061704

Figure 26. A Cost Effective 6.6 W, 3.3 V Flyback DC-DC Converter.

Output regulation is achieved by using secondary side voltage

reference, U3. The resistor divider formed by R10 and R11,

together with the reference voltage, determines the output

voltage. Diode D3 and C13 form a soft-ﬁnish network that,

together with the internal duty cycle and current limit soft-start

of the DPA-Switch, prevent output overshoot at start-up.

Resistor R7 ensures that the soft-ﬁnish capacitor is discharged

quickly when the output falls out of regulation. Control loop

response is shaped by R6, C16, R12, C14, R9, R4 and C5,

providing a wide bandwidth and good phase margin at gain

crossover. Since the PWM control in DPA-Switch is voltage mode,

no slope compensation is required for duty cycles above 50%.

Cost Effective 6.6 W Flyback Converter

The DPA-Switch ﬂyback power supply provides a cost effective

solution for high density PoE and VoIP DC-DC applications.

Figure 26 shows a typical implementation of a single output

ﬂyback converter using the DPA423G. For applications that

require input to output isolation, this simple, low component

count design delivers 6.6 W at 3.3 V from a 36 VDC to 57 VDC

input with a nominal efﬁciency at 48 VDC of 80%.

Resistor R2 programs the input undervoltage and overvoltage

thresholds to 33 V and 86 V respectively. Resistors R1 and R3

program the internal device current limit. The addition of line

sense resistor R1 reduces the current limit with increasing input

voltage, preventing excessive overload output current. In this

design the overload output current varies less than ±2.5%

across the entire input voltage range. Controlling the current

limit also reduces secondary component stress and leakage

inductance spikes, allowing the use of a lower VRRM (30 V rather

than 40 V) Schottky output diode, D2.

The primary side Zener clamp VR1 ensures the peak drain

voltage is kept below the 220 V BVDSS rating of U1 under input

surge and overvoltage events. During normal operation, VR1

does not conduct and C2 is sufﬁcient to limit the peak drain

voltage.

The primary bias winding provides CONTROL pin current after

start-up. Diode D3 rectiﬁes the bias winding, while components

R5 and C8 reduce high frequency switching noise and prevent

peak charging of the bias voltage. Capacitor C3 provides local

decoupling of U1 and should be physically close to the

CONTROL and SOURCE pins. Energy storage for start-up and

auto-restart timing is provided by C4.

The secondary is rectiﬁed by D2 and the Low ESR tantalum

output capacitors, C5-C7, minimizing switching ripple and

maximizing efﬁciency. A small footprint secondary output choke

L1 and ceramic output capacitor C9 are adequate to reduce

high frequency noise and ripple to below 35 mV peak-peak

under full load conditions.

The output voltage is sensed by the voltage divider formed by

resistors R8 and R9 and is fed to the low voltage 1.24 V

reference U3. Feedback compensation is provided by R6, R7

and C11 together with C4 and R4. Capacitor C10 provides a

soft-ﬁnish characteristic, preventing output overshoot during

start-up.

Low Cost PoE VoIP Phone Converter

The basic circuitry to support IEEE standard 802.3af Power

over Ethernet (PoE) is straightforward. Class 0 signature and

classiﬁcation circuits can be implemented with a single resistor

and the required undervoltage lockout function is a voltage

Rev. T 12/12

DPA422-426

www.powerint.com

Figure 27. PoE Interface Circuit Using a Bipolar Transistor Pass-Switch and DPA424P.

DPA424PN

BAV

19WS

Q22

Si4804

D21

SL13

15 V

Q21

Si4804

1 µF

100 V

VR1

SMAJ

150 C5

47 µF

10 V

D41

BAV19WS

D31

20CJQ060

R21

10 Ω

R22

10 Ω

R23

10 kΩ

VR21

C21

2.2 nF

649 kΩ

1.0 Ω

C22-C24

100 µF 5 V

VR41

6.8 V

D42

IN4148

C31

100 µF

10 V

C41

4.7 µF, 35 V

C25

1 µF

10 V

1 µH 2.5 A

PoE Interface

16 µH 4 A

4.7 µF

20 V

D11

BAV19WS

LM431AIM3

C11

2.2 µF

10 V

R14

1 kΩ

R15

10 kΩ

R16

10 kΩ

160 Ω

R12

150 Ω

R13

11 Ω

1 µF

100 V

PC357

N1T

R11

10 kΩ

PI-3824-031008

DPA-Switch

5 V, 2.4 A

RTN

7.5 V, 0.4 A

20 V, 10 mA

C13

68 nF

Ethernet

(RJ-45)

Connector

13.3 kΩ

R52

20 kΩ

D101

DL4002

D102

D103

DL4002

D104

D105

DL4002

D106

D107

DL4002

D108

R22

10 kΩ

R23

174 k

R21

10 k

R51

24.9 kΩ

1% 1/4 W

D51

BAV19

D52

BAV19

C51

1 nF

50 V

R54

20 Ω

VR51

28 V

(1,2)

(4,5)

(3,6)

(7,8)

CONTROL

C12

100 nF

R53

20 kΩ

Q51

TIP29C (100 V/1 A)

or MMBTA06

220 nF

Q20

MMBTS3906

controlled pass-switch. By adding this circuitry to the front end

of a DPA converter, a low cost and low component count PoE

powered device (PD) power supply can be implemented.

Figure 27 shows a typical PD solution.

The PoE speciﬁcation requires the PD to provide three

fundamental functions: discovery, classiﬁcation, and pass-

switch connection.

When input voltage is applied to the PD, it must present the

correct discovery signature impedance in the voltage range of

2.5 VDC to 10 VDC. This impedance is provided by R51 in

Figure 27.

The second “classiﬁcation” phase occurs at input voltages

15 VDC to 20 VDC. The PD must draw a speciﬁed current to

identify the device class (Class 0 speciﬁes 0.5 mA to 4 mA).

This is again accomplished by resistor R51.

In the third phase, the bipolar pass-switch (Q51 in Figure 27)

connects the input voltage to the power supply at voltages

above approximately 30 VDC (28 V+VR52). Zener diode VR51

conducts, driving the current through resistor R52 to the base

of Q51. Resistor R53 prevents turn-on under other conditions.

Once the Power supply has started, components D51, D52,

C51 and R54 enhance the base-current drive by coupling

power from the power supply bias winding.

Once the three start up phases have been successfully

completed, the DPA-Switch is allowed to function as a forward

converter (described in Figure 25 and accompanying text).

Key Application Considerations

DPA-Switch Design Considerations

Power Table

This section provides a description of the assumptions used to

generate the power tables (Tables 1 and 3 through 6) and

explains how to use the information provided by them.

All Power tables: Tables 1 and 3 through 6

• Maximum output power is limited by the device internal

current limit. This is the peak output power which could

become the continuous output power, provided adequate

heat sinking is used.

• Data assumes adequate heat sinking to keep the junction

temperature at or below 100 °C and worst case RDS(ON) at

TJ = 100 °C.

• The use of P and G packages are recommended for device

dissipation equal to or less than 1.5 W only due to package

thermal limitation. For device dissipation above 1.5 W, use

R package.

Rev. T 12/12

DPA422-426

www.powerint.com

Forward power tables: Tables 1 (upper half), 3 and 4

• Output power ﬁgures are based on forward topology using

Schottky diode rectiﬁcation. Up to 5% higher output power is

possible using synchronous rectiﬁcation.

• Dissipation data assumes a diode loss representing 6% of the

total output power and combined loss in magnetic compo-

nents representing 6% of the total output power.

DPA-Switch losses are based on a ratio between conduction

and switching losses of approximately 3:1. These assump-

tions are typical for a single 5 V output forward converter

design using Schottky rectiﬁcation and adequately designed

magnetic components.

Flyback power tables: Tables 1 (lower half), 5 and 6

• Output power and dissipation ﬁgures are based on a 5 V

output using Schottky diode rectiﬁcation with an efﬁciency of

85%. Values are generated by calculation based on

I2 × RDS(ON) losses and characterization of switching losses,

correlated to bench measurement of each DPA-Switch device.

• Device dissipations above 1.5 W are possible using the

R package. However the forward converter topology is

recommended for such higher power designs.

The power tables provide two types of information. The ﬁrst is

the expected device dissipation for a given output power. The

second is the maximum power output. Each table speciﬁes the

input voltage range and assumes a single 5 V output using

Schottky diode rectiﬁcation.

For example, referring to Table 1, for 36 VDC to 75 VDC input

range, DPA424 would typically dissipate 1 W in a 23 W forward

converter and has a maximum power capacity of 35 W. In the

same converter, DPA425 would dissipate 0.5 W. Selecting

DPA425 with associated reduced dissipation would increase

overall converter efﬁciency by approximately 2%.

Issues Affecting Dissipation:

1. Using synchronous rectiﬁcation will tend to reduce device

dissipation.

2. Designs with lower output voltages and higher currents will

tend to increase the device dissipation listed in the power

table.

3. Reduced input voltage decreases the available output power

for the same device dissipation. Tables 3 to 6 are the power

tables for 16 VDC and 24 VDC input voltages. Input voltages

below 16 V are possible, but since the internal start-up

current source is not speciﬁed at voltages below 16 V, an

external chip supply current should be fed into the

CONTROL pin approximately equal to but less than ICD1.

DPA-Switch Selection

Use Tables 1 and 3 through 6 to select the DPA-Switch based

on device dissipation. Selecting the optimum DPA-Switch

depends upon required maximum output power, efﬁciency,

heat sinking constraints and cost goals. With the option to

externally reduce current limit, a larger DPA-Switch may be

used for lower power applications where higher efﬁciency is

needed or minimal heat sinking is available. Generally, selecting

the next larger device, than is required for power delivery will

give the highest efﬁciency. Selecting even larger devices may

Output Power Table

16-32 VDC Range (Forward)2

Total Device

Dissipation

Product3

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA422 3.5 W 4.5 W - - - 5.0 W

DPA423 5 W 7 W - - - 7.5 W

DPA424 7 W 10 W 15 W - - 15.5 W

DPA425 10 W 14 W 22 W 27 W - 31 W

DPA426 12 W 16.5 W 25 W 31 W 37 W 43 W

Table 3. Output Power Table for 16-32 VDC Input Voltage.

Notes:

1. Limited by device internal current limit.

2. See text in this section for a complete description of assumptions.

3. See Part Ordering Information.

Output Power Table

16-32 VDC Input Range (Flyback)2

Total Device

Dissipation

Product3

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA422 3 W - - - 4.5 W

DPA423 5 W - - - 6 W

DPA424 6.5 W 8.5 W 10 W - 11 W

DPA425 7 W 10 W 12 W 15 W 22 W

Output Power Table

24-48 VDC Range (Forward)2

Total Device

Dissipation

Product3

0.5 W 1 W 2.5 W 4 W 6 W

Max

Power

Output1

DPA422 5.5 W 7 W - - - 8 W

DPA423 8 W 11 W - - - 11.5 W

DPA424 11 W 16 W 23.5 W - - 25 W

DPA425 16 W 22 W 35 W 43 W - 47 W

DPA426 18 W 25 W 39 W 48 W 58 W 65 W

Table 4. Output Power Table for 24-48 VDC Input Voltage

(See Table 3 for Notes).

Table 5. Flyback Output Power Table for 16-32 VDC Input Voltage

(See Table 3 for Notes).

Output Power Table

24-48 VDC Input Range (Flyback)2

Total Device

Dissipation

Product3,4

0.5 W 0.75 W 1 W 1.5 W

Max

Power

Output1

DPA422 3 W - - - 7 W

DPA423 7 W - - - 8.5 W

DPA424 8.5 W 11.5 W 14 W - 17 W

Table 6. Flyback Output Power Table for 24-48 VDC Input Voltage.

Notes:

1. Maximum output power is limited by device internal current limit.

2. See text in this section for a complete description of assumptions.

3. See Part Ordering Information.

4. Higher switching losses may prevent DPA425 from delivering more power than a

smaller device.

Rev. T 12/12

DPA422-426

www.powerint.com

give little or no improvement in efﬁciency due to the

improvement in conduction losses being negated by larger

device switching losses. Figure 50 provides information on

switching losses. This together with conduction loss

calculations give an estimate of device dissipation.

Primary Clamp

A primary clamp network is recommended to keep the peak

DRAIN voltage due to primary leakage inductance to below the

BVDSS speciﬁcation. A Zener diode combined with a small value

capacitor connected across the primary winding is a low cost

and low part count implementation.

Output Rectiﬁcation

Rectiﬁcation of the secondary is typically performed using

Schottky diodes or synchronous rectiﬁcation. Schottky diodes

are selected for peak inverse voltage, output current, forward

drop and thermal conditions. Synchronous rectiﬁcation

requires the additional complication of providing gate drive. The

speciﬁed line undervoltage and line overvoltage thresholds of

DPA-Switch simpliﬁes deriving gate drive directly from the

transformer secondary winding for many applications. The

turns ratio of the transformer together with the under/over-

voltage thresholds deﬁnes the minimum and maximum gate

voltages, removing the need for Zeners to clamp the gate

voltage.

Soft-Start

Generally a power supply experiences maximum stress at start-

up before the feedback loop achieves regulation. For a period

of 5 ms the on-chip soft-start linearly increases the duty cycle

from zero to the default DCMAX at turn-on. In addition, the

primary current limit increases from 85% to 100% over the

same period. This causes the output voltage to rise in an

orderly manner allowing time for the feedback loop to take

control of the duty cycle. This integrated soft-start reduces the

stress on the DPA-Switch MOSFET, clamp circuit and output

diode(s), and helps prevent transformer saturation during start-

up. Also, soft-start limits the amount of output voltage

overshoot, and in many applications eliminates the need for a

soft-ﬁnish capacitor. If necessary, to remove output overshoot,

a soft-ﬁnish capacitor may be added to the secondary

reference.

Switching Frequency

The FREQUENCY pin of DPA-Switch offers a switching

frequency option of 400 kHz or 300 kHz. Operating at 300 kHz

will increase the amount of magnetization energy stored in the

transformer. This is ideal for applications using synchronous

rectiﬁcation driven directly from the transformer secondary

where this energy can be used to drive the catch MOSFET gate.

Transformer Design

It is recommended that the forward converter transformer be

designed for maximum operating ﬂux swing of 1500 Gauss and

a peak ﬂux density of 3500 Gauss. When operating at the

maximum current limit of the selected DPA-Switch (during

overload conditions), neither magnetic component (transformer

and output inductor) should be allowed to saturate. When a

larger device than necessary has been selected, reducing the

internal current limit close to the operating peak current limits

overload power and minimizes the size of the secondary

components.

No-load and Standby Consumption

Cycle skipping operation at light or no-load can signiﬁcantly

reduce power loss. In addition this operating mode ensures

that the output maintains regulation even without an external

minimum load. However, if cycle skipping is undesirable in a

particular application, it can be avoided by adding sufﬁcient

pre-load.

DPA-Switch Layout Considerations

The DPA-Switch can operate with large DRAIN current, the

following guidelines should be carefully followed.

Primary Side Connections

The tab of DPA-Switch R package is the intended return path

for the high switching currents. Therefore, the tab should be

connected by wide, low impedance traces back to the input

decoupling capacitor. The SOURCE pin should not be used to

return the power currents; incorrect operation of the device may

result. The SOURCE is only intended as a signal ground. The

device tab (SOURCE) is the correct connection for high current

with the R package.

The CONTROL pin bypass capacitor should be located as

close as possible to the SOURCE and CONTROL pins and its

SOURCE connection trace should not be shared by the main

MOSFET switching currents. All SOURCE pin referenced

components connected to the LINE-SENSE or EXTERNAL

CURRENT LIMIT pins should also be located closely between

their respective pin and SOURCE. Once again, the SOURCE

connection trace of these components should not be shared by

the main MOSFET switching currents. It is critical that the tab

(SOURCE) power switching currents are returned to the input

capacitor through a separate trace that is not shared by the

components connected to CONTROL, LINE-SENSE or

EXTERNAL CURRENT LIMIT pins.

Any traces to the L or X pins should be kept as short as

possible and away from the DRAIN trace to prevent noise

coupling. LINE-SENSE resistor (R1 in Figure 25) should be

located close to the LINE-SENSE pin to minimize the trace

length on the LINE-SENSE pin side.

In addition to the CONTROL pin capacitor (C6 in Figure 25), a

high frequency bypass capacitor in parallel is recommended as

close as possible between SOURCE and CONTROL pins for

better noise immunity. The feedback optocoupler output

should also be located close to the CONTROL and SOURCE

pins of DPA-Switch.

Heat Sinking

To maximize heat sinking of the DPA-Switch R or G package

and the other power components, special thermally conductive

PC board material (aluminum clad PC board) is recommended.

This has an aluminum sheet bonded to the PC board during the

manufacturing process to provide heat sinking directly and

allow the attachment of an external heat sink. If normal PC

board material is used (such as FR4), providing copper areas on

Rev. T 12/12

DPA422-426

www.powerint.com

both sides of the board and using thicker copper will improve

heat sinking.

If an aluminum clad board is used then shielding of switching

nodes is recommended. This consists of an area of copper

placed directly underneath switching nodes such as the drain

node, and output diode to provide an electrostatic shield to

prevent coupling to the aluminum substrate. These areas are

connected to input negative in the case of the primary and

output return for secondary. This reduces the amount of

capacitive coupling into the insulated aluminum substrate that

can then appear on the output as ripple and high frequency

noise.

Quick Design Checklist

As with any power supply design, all DPA-Switch designs

should be veriﬁed on the bench to make sure that component

speciﬁcations limits are not exceeded under worst case

conditions. The following minimum set of tests for DPA-Switch

forward converters is strongly recommended:

1. Maximum drain voltage – Verify that peak VDS does not

exceed minimum BVDSS at highest input voltage and maxi-

mum overload output power. It is normal, however, to have

additional margin of approximately 25 V below BVDSS to allow

for other power supply component unit-to-unit variations.

Maximum overload output power occurs when the output is

loaded to a level just before the power supply goes into auto-

restart (loss of regulation).

2. Transformer reset margin – Drain voltage should also be

checked at highest input voltage with a severe load step

(50-100%) to verify adequate transformer reset margin. This

test shows the duty cycle at high input voltage, placing the

most demand on the transformer reset circuit.

3. Maximum drain current – At maximum ambient temperature,

maximum input voltage and maximum output load, verify

drain current waveforms at start-up for any signs of trans-

former or output inductor saturation and excessive leading

edge current spikes. DPA-Switch has a leading edge

blanking time of 100 ns to prevent premature termination of

the on cycle. Verify that the leading edge current spike does

not extend beyond the blanking period.

4. Thermal check – At maximum output power, minimum input

voltage and maximum ambient temperature, verify that

temperature speciﬁcations are not exceeded for the

transformer, output diodes, output choke(s) and output

capacitors. The DPA-Switch is fully protected against over-

temperature conditions by its thermal shutdown feature. It is

recommended that sufﬁcient heat sinking is provided to keep

the tab temperature at or below 115 °C (R package),

SOURCE pins at or below 100 °C (P/G packages) under

worst case continuous load conditions (at low input voltage,

maximum ambient and full load). This provides adequate

margin to minimum thermal shutdown temperature

(130 °C) to account for part-to-part RDS(ON) variation. When

monitoring device temperatures, note that the junction-to-

case thermal resistance should be accounted for when

estimating die temperature.

Design Tools

Up-to-date information on design tools is available at the Power

Integrations website: www.powerint.com.

Rev. T 12/12

DPA422-426

www.powerint.com

PI-2883-060602

Solder Side

Component Side

TOP VIEW

Transformer

Out

Maximize hatched copper area for optimum heat sinking

Via between board layers

Opto-

coupler

Inductor

(Coupled)

Output

Diode

DPA-Switch

PI-3805-012904

Opto-

coupler

Out

DPA-Switch Heat Sink

Bottom Diode

Heat Sink

V V

C L X F

S D S

DPA-Switch

Top Side PCB

Bottom Side PCB

(Two-sided printed circuit board)

TOP VIEW

Maximize hatched copper area for optimum heat sinking

Via between board layers

Figure 28. Layout Considerations for DPA-Switch Using R Package.

Figure 29. Layout Considerations for DPA-Switch Using G Package.

Rev. T 12/12

DPA422-426

www.powerint.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 33

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Control Functions

Switching

Frequency fOSC TJ = 25 °C

FREQUENCY Pin

Connected to SOURCE 375 400 425

kHz

FREQUENCY Pin

Connected to CONTROL 282 300 317

Duty Cycle (Prior to

Cycle Skipping) DCMIN 4 6 %

Maximum Duty Cycle DCMAX IC = ICD1

VL = 0 V 71 75 79

%IL = 80 µA 52 62 71

IL = 115 µA 32 42 57

Control Current at

Start of Cycle Skipping IC(SKIP) TJ = 25 °C; fOSC = 400 kHz

DPA422 6.3 8.0

DPA423 7.2 9.0

DPA424 8.2 10.0

DPA425 10.0 12.0

DPA426 11.5 14.0

External Bias

Current IBTJ = 25 °C; fOSC = 400 kHz

DPA422 1.8 2.5 3.1

DPA423 2 2.8 3.5

DPA424 2.5 3.5 4.4

DPA425 3.6 4.8 6.0

DPA426 4.4 5.7 7.1

Absolute Maximum Ratings(1,5)

DRAIN Voltage .....................................................-0.3 V to 220 V

DRAIN Peak Current: DPA422......................................1.31 A

DPA423......................................1.75 A

DPA424....................................... 3.5 A

DPA425...........................................7 A

DPA426....................................... 9.6 A

CONTROL Voltage ................................................ -0.3 V to 9 V

CONTROL Current .........................................................100 mA

LINE SENSE Pin Voltage ...................................... -0.3 V to 9 V

EXTERNAL CURRENT LIMIT Pin Voltage ........ -0.3 V to 9 V

FREQUENCY Pin Voltage ....................................... -0.3 V to 9 V

Storage Temperature-65 °C to 150 °C

Operating Junction Temperature(2) ....................-40 °C to 150 °C

Lead Temperature(3) ....................................................... 260 °C

Notes:

1. All voltages referenced to SOURCE, TA = 25 °C.

2. Normally limited by internal circuitry.

3. 1/16” from case for 5 seconds.

4. Maximum ratings speciﬁed may be applied, one at a time,

without causing permanent damage to the product.

Exposure to Absolute Maximum Rating conditions for

extended periods of time may affect product reliability.

Thermal Resistance

Thermal Resistance: P or G Package:

(qJA) ........................... 70 °C/W(1); 60 °C/W(2)

(qJC)(3) ............................................11 °C/W

R Package:

(qJA) ................................................. 40 °C/W(4)

(qJA) ................................................. 30 °C/W(5)

(qJC)(6) ................................................2 °C/W

Notes:

1. Soldered to 0.36 sq. in. (232 mm2), 2 oz. (610 g/m2) copper clad.

2. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.

3. Measured on pin 7 (SOURCE) close to plastic interface.

4. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.

5. Soldered to 3 sq. in. (1935 mm2), 2 oz. (610 g/m2) copper clad.

6. Measured at the back surface of tab.

Rev. T 12/12

DPA422-426

www.powerint.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 33

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Control Functions (cont.)

Softstart Time tSOFT TJ = 25 °C; DCMIN to DCMAX 5 7.2 ms

PWM Gain DCREG TJ = 25 °C; IC = IC(SKIP) + IB)/2 -28 -22 -18 %/mA

PWM Gain

Temperature Drift See Note A -0.01 %/mA/°C

Dynamic Impedance ZCTJ = 25 °C; IC = IC(SKIP) + IB)/2 10 15 22 W

Dynamic Impedance

Temperature Drift 0.18 %/°C

CONTROL Pin Internal

Filter Pole 30 kHz

Shutdown/Auto-Restart

CONTROL Pin

Charging Current IC(CH))

During Startup and Auto-Restart:

VC = 5.0 V; VD = 16 V & 40 V; TJ = 25 °C -5.2 -4 -3

Average Current at the Beginning of

Softstart: VC = 5.0 V; VD = 16 V & 40 V;

TJ = 25 °C

-19

Charging Current

Temperature Drift See Note A -0.6 %/ °C

Auto-Restart Upper

Threshold Voltage VC(AR)U 5.8 V

Auto-Restart Lower

Threshold Voltage VC(AR)L 4.5 4.8 5.1 V

Auto-Restart

Hysteresis Voltage VC(AR)Hyst 0.8 1 V

Auto-Restart

Duty Cycle DC(AR)

CCONTROL = 22 µF; fOSC = 400 kHz;

VX = 0 V 10 %

Auto-Restart

Frequency f(AR)

CCONTROL = 22 µF; fOSC = 400 kHz;

VX = 0 V 3.8 Hz

Line-Sense (L) and External Current Limit (X) Inputs

Line Undervoltage

Threshold Current and

Hysteresis (L Pin)

IUV TJ = 25 °C

Threshold from Off to On 48 50 52

µAThreshold from On to Off 44.5 47 49.5

Hysteresis 2 3

Line Overvoltage or

Remote ON/OFF

Threshold Current and

Hysteresis (L Pin)

IOV TJ = 25 °C

Threshold from On to Off 135 149

µAThreshold from Off to On 117 131

Hysteresis 4

Remote ON/OFF

Negative Threshold

Current and

Hysteresis (X Pin)

IREM TJ = 25 °C

Threshold from On to Off -27 -21.5 -16

µAThreshold from Off to On -25.5

Hysteresis 4.5

Rev. T 12/12

DPA422-426

www.powerint.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 33

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Line-Sense (L) and External Current Limit (X) Inputs (cont.)

LINE-SENSE Pin

Short-Circuit Current IL(SC)

VL = VC175 240 380 µA

VL = 0 V -230 -170

X Pin Short-Circuit

Current IX(SC) VX = 0 V Normal Mode -270 -230 -185 µA

Remote OFF Using L Pin -105 -85 -65

Line Pin Voltage

(Positive Current) VL

IL = IUV 2.05 2.35 2.6

IL = IOV 2.1 2.5 2.9

X Pin Voltage

(Negative Current) VX

IX = -50 µA 1.35

IX = -150 µA 1.25

Maximum Duty Cycle

Reduction Onset

Threshold Current

IL(DC) TJ = 25 °C 55 µA

Remote OFF DRAIN

Supply Current ID(RMT)

VD = 40 V

VX = 0 V

VL = Floating 0.6 1.1 mA

VL = VC0.9 1.5

LINE-SENSE Pin

Voltage Turn-On

Threshold in

Synchronous Mode

VL(TH) 0.6 1 1.4 V

On-Time Pulse Width

for Synchronization tON(SYNC)

fOSC = 400 kHz 120 2250

fOSC = 300 kHz 120 3080

Off-Time Pulse Width

for Synchronization tOFF(SYNC) 0.25 7.7 µs

Synchronous

Turn-On Delay tDELAY(SYNC) From Synchronous On to Drain Turn-On 250 ns

Frequency (F) Input

FREQUENCY Pin

Threshold Voltage VF1.1 4 V

FREQUENCY Pin Input

Current IF

VF = 0 V -0.38 µA

VF = VC17 120

FREQUENCY Pin

Delay Time tDELAY(VF) 2µs

Circuit Protection

Self Protection

Current Limit

(See Note B)

ILIMIT TJ = 25 °C

DPA422 di/dt = 225 mA/µs 0.86 0.935 1.0

DPA423 di/dt = 300 mA/µs 1.16 1.25 1.34

DPA424 di/dt = 600 mA/µs 2.32 2.50 2.68

DPA425 di/dt = 1.25 A/µs 4.65 5.00 5.35

DPA426 di/dt = 1.75 A/µs 6.50 7.00 7.50

Rev. T 12/12

DPA422-426

www.powerint.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 33

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Circuit Protection (cont.)

Initial Current Limit IINIT VD = 35 V 0.9 ×

ILIMIT(MIN)

Leading Edge

Blanking Time tLEB TJ = 25 °C 100 ns

Current Limit Delay TIL(D) IC = (IC(SKIP) + IB)/2 100 ns

Thermal Shutdown

Temperature TJ(SD) 130 137 145 °C

Thermal Shutdown

Hysteresis TJ(SD)HYST 27 °C

Power-Up Reset

Threshold Voltage VC(RESET) 1.5 2.75 4 V

Output

ON-State Resistance RDS(ON)

DPA422

ID = 150 mA

TJ = 25 °C 2.60 3.00

TJ = 100 °C 4.00 4.60

DPA423

ID = 300 mA

TJ = 25 °C 1.30 1.50

TJ = 100 °C 2.00 2.30

DPA424

ID = 600 mA

TJ = 25 °C 0.65 0.75

TJ = 100 °C 1.00 1.15

DPA425

ID = 1.25 A

TJ = 25 °C 0.33 0.38

TJ = 100 °C 0.50 0.58

DPA426

ID = 1.75 A

TJ = 25 °C 0.24 0.28

TJ = 100 °C 0.37 0.43

OFF-State Drain

Leakage Current IDSS

VX, VL = Floating;

VD = 150 V;

TJ = 125 °C;

IC = (IC(SKIP) + IB)/2

DPA422 33

µA

DPA423 65

DPA424 130

DPA425 260

DPA426 360

Breakdown Voltage BVDSS

VX, VL = Floating; TJ = 25 °C;

IC = (IC(SKIP) + IB)/2; See Note C 220 V

Rise Time tRMeasured in a Typical Application 10 ns

Fall Time tFMeasured in a Typical Application 10 ns

Rev. T 12/12

DPA422-426

www.powerint.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 33

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Supply Voltage Characteristics

DRAIN Supply Voltage See Note D 16 V

Shunt Regulator

Voltage VC(SHUNT) IC = (IC(SKIP) + IB)/2; TJ = 25 °C 5.6 5.85 6.0 V

Shunt Regulator

Temperature Drift IC = (IC(SKIP) + IB)/2 ±50 PPM/°C

CONTROL Supply/

Discharge Current

ICD1

Output

MOSFET Enabled

VL = 0 V;

fOSC = 400 kHz

DPA422 1.6 2.0 2.4

DPA423 1.9 2.3 2.7

DPA424 2.6 3.0 3.4

DPA425 3.7 4.3 4.8

DPA426 4.8 5.4 6

ICD2

Output MOSFET Disabled

VL = 0 V; fOSC = 400 kHz 0.4 0.73 1.2

NOTES:

A. For speciﬁcations with negative values, a negative temperature coefﬁcient corresponds to an increase in magnitude with increasing

temperature, and a positive temperature coefﬁcient corresponds to a decrease in magnitude with increasing temperature.

B. For externally adjusted current limit values, please refer to Figure 35 (Current Limit vs. External Current Limit Resistance) in the

Typical Performance Characteristics section.

C. Breakdown voltage may be checked against minimum BVDSS speciﬁcation by ramping the DRAIN pin voltage up to but not

exceeding minimum BVDSS.

D. It is possible to start up and operate DPA-Switch at DRAIN voltages well below 16 V. However, the CONTROL pin charging

current is reduced, which affects start-up time, auto-restart frequency, and auto-restart duty cycle. Refer to Figure 45, the

characteristic graph on CONTROL pin charge current (IC) vs. DRAIN voltage for low voltage operation characteristics.

Rev. T 12/12

DPA422-426

www.powerint.com

1.4

1.2

0.6

0.2

0.4

0.8

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2

Time (µs)

DRAIN Current (Normalized)

PI-2889-061302

tLEB (Blanking Time) +

tIL(D) (Current Limit Delay)

Temperature Range:

-40 °C to 125 °C

120

100

0 2 4 6 8 10

CONTROL Pin Voltage (V)

CONTROL Pin Current (mA)

Slope

Dynamic

Impedance =

PI-2880-060302

PI-2039-033001

DRAIN

VOLTAGE

0 V

90%

10%

90%

D = t1

Figure 30. Duty Cycle Measurement.

Figure 32. Typical Drain Operation Current Envelope.

PI-2860-050602

5-50 V

40 V

0-15 V

0.1 µF47 µF

470 Ω

5 W

470 Ω

0-200 kΩ

0-70 kΩ

NOTE: This test circuit is not applicable for current limit or output characteristic measurements.

SF X

CONTROL DPA-Switch

Figure 33. DPA-Switch General Test Circuit.

Figure 31. CONTROL Pin I-V Characteristic.

Rev. T 12/12

DPA422-426

www.powerint.com

The following precautions should be followed when testing

DPA-Switch by itself outside of a power supply. The schematic

shown in Figure 33 is suggested for laboratory testing of

DPA-Switch.

When the DRAIN pin supply is turned on, the part will be in the

auto-restart mode. The CONTROL pin voltage will be

oscillating at a low frequency between 4.8 V and 5.8 V and the

drain is turned on every eighth cycle of the CONTROL pin

oscillation. If the CONTROL pin power supply is turned on

while in this auto-restart mode, there is only a 12.5% chance

that the CONTROL pin oscillation will be in the correct state

(drain active state) so that the continuous drain voltage

waveform may be observed. It is recommended that the VC

power supply be turned on ﬁrst and the DRAIN pin power

supply second if continuous drain voltage waveforms are to be

observed. The 12.5% chance of being in the correct state is

due to the divide-by-8 counter. Temporarily shorting the

CONTROL pin to the SOURCE pin will reset DPA-Switch, which

then will come up in the correct state.

BENCH TEST PRECAUTIONS FOR EVALUATION OF ELECTRICAL CHARACTERISTICS

Typical Performance Characteristics

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

1.0

-230 -180 -130 -80 -30

IX (µA)

Current Limit

(Normalized to Internal Current Limit)

PI-2838-032202

Figure 34. Current Limit vs. EXTERNAL CURRENT LIMIT Pin Current.

0.3

0.2

0.1

0.5

0.4

0.6

0.7

0.8

0.9

1.0

1.1

0 5 10 15 20 25 30 35

External Current Limit Resistor RIL (kΩ)

Current Limit

(Normalized to Internal Current Limit)

PI-2839-032202

Minimum

Typical

Maximum

Figure 35. Current Limit vs. External Current Limit Resistance.

Rev. T 12/12

DPA422-426

www.powerint.com

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2872-051602

Current Limit

(Normalized to 25 °C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2865-051702

Overvoltage Threshold

(Normalized to 25 °C)

Figure 38. External Current Limit vs. Temperature with RIL = 12 kW.Figure 39. Overvoltage Threshold vs. Temperature.

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2867-051502

Under-Voltage Threshold

(Normalized to 25 °C)

7.0

4.0

6.0

5.0

0.0

-200 -100 -50-150 050 100 150 200 250

LINE-SENSE Pin Current ( A)

LINE-SENSE Pin Voltage (V)

PI-2871-051502

2.0

1.0

3.0

Figure 40. Undervoltage Threshold vs. Temperature. Figure 41. LINE-SENSE Pin Voltage vs. Current.

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2868-051502

Output Frequency

(Normalized to 25 °C)

Figure 36. Frequency vs. Temperature.

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2866-051702

Current Limit

(Normalized to 25 °C)

Figure 37. Internal Current Limit vs. Temperature.

Typical Performance Characteristics (cont.)

Rev. T 12/12

DPA422-426

www.powerint.com

Typical Performance Characteristics (cont.)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2863-011904

CONTROL Pin Current

(Normalized to 25 °C)

Figure 43. CONTROL Pin Current at Minimum Duty

Cycle vs. Temperature.

1.6

1.0

1.4

1.2

0-225

-250 -150

-175

-200 -125 -50 -25

-100-75 0

EXTERNAL CURRENT LIMIT Pin Current (

µA)

EXTERNAL CURRENT LIMIT

Pin Voltage (V)

PI-2869-042408

0.4

0.2

0.6

0.8

VX = 1.375 - IXx 843 Ω

-200 µA ≤ IX ≤ -25 µA

Figure 42. EXTERNAL CURRENT LIMIT Pin Voltage vs. Current

5.0

3.5

3.0

4.0

4.5

0.0

010 20 30 40 50 60

DRAIN Voltage (V)

CONTROL Pin

Charging Current (mA)

PI-2841-051502

1.0

0.5

1.5

2.5

2.0

VC = 5 V

Figure 45. IC vs. DRAIN Voltage.

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-2864-051702

Onset Threshold Current

(Normalized to 25 °C)

Figure 44. Max. Duty Cycle Reduction Onset Threshold

Current vs. Temperature.

1.1

1.0

0.9

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

Breakdown Voltage

(Normalized to 25 °C)

PI-176B-033001

Figure 47. Breakdown Voltage vs. Temperature.

1.2

0.8

1.0

-50 -20 0255075 100 125 150

Junction Temperature ( C)

Remote OFF DRAIN Supply Current

(Normalized to 25 C)

PI-2870-051502

0.2

0.4

0.6

Figure 46. Remote OFF DRAIN Supply Current vs. Temperature.

Rev. T 12/12

DPA422-426

www.powerint.com

Part Ordering Information

• DPA-Switch Product Family

• Series Number

• Package Identier

G Plastic Surface Mount DIP (422, 423, 424 & 425 only)

P Plastic DIP

R Plastic TO-263-7C (available only with TL option)

• Lead Finish

Blank Standard (Sn Pb)

N Pure Matte Tin (Pb-Free) (P & G)

• Tape & Reel and Other Options

Blank Standard Conﬁgurations

TL Tape & Reel, (G package: 1000 min./mult., R package: 750 min./mult.)

Typical Performance Characteristics (cont.)

0 1 2 3 4 5

Drain Voltage (V)

PI-2849-050302

Drain Current (A)

Scaling Factors:

DPA423 = 0.18

DPA424 = 0.36

DPA425 = 0.72

DPA426 = 1.00

TCASE = 25 °C

TCASE = 100 °C

Figure 48. Output Characteristics.

2.5

1.5

0.5

-50 -25 0 25 7550 100 125 150

Temperature (°C)

PI-2850-011904

On-resistance

(Normalized to 25 °C)

Figure 49. On-Resistance vs. Temperature.

0 20 40 60 80 100 120 140 160

100

1000

10000

PI-2851-011904

DRAIN Voltage (V)

Capacitance (pF)

Scaling Factors:

DPA423 = 0.18

DPA424 = 0.36

DPA425 = 0.72

DPA426 = 1.00

Figure 50. COSS vs. DRAIN Voltage.

2.5

1.5

0.5

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

In-Circuit Peak Drain Current Normalized to

Nominal Device Internal Current Limit

PI-2877-061704

Switching Loss (W) at 400 kHz

Scaling Factors:

DPA423 = 0.08

DPA424 = 0.20

DPA425 = 0.55

DPA426 = 1.00

Data taken from a typical

forward converter design.

Figure 51. Typical Switching Loss.

DPA 423 G N - TL

Rev. T 12/12

DPA422-426

www.powerint.com

PI-2076-040110

8 5

PDIP-8 (P Package)

D S .004 (.10)

-E-

-D-

-F-

DIM

Inches

0.367-0.387

0.240-0.260

0.125-0.145

0.015-0.040

0.120-0.140

0.057-0.068

0.014-0.022

0.008-0.015

0.100 BSC

0.030 (MIN)

0.300-0.320

0.300-0.390

0.300 BSC

9.32-9.83

6.10-6.60

3.18-3.68

0.38-1.02

3.05-3.56

1.45-1.73

0.36-0.56

0.20-0.38

2.54 BSC

0.76 (MIN)

7.62-8.13

7.62-9.91

7.62 BSC

Notes:

1. Package dimensions conform to JEDEC

specification MS-001-AB for standard dual in-line

(DIP) package .300 inch row spacing (PLASTIC)

8 leads (issue B, 7/85).

2. Controlling dimensions are inches.

3. Dimensions shown do not include mold flash

or other protrusions. Mold flash or protrusions

shall not exceed .006 (.15) on any side.

4. D, E and F are reference datums on the molded

body.

Q P08A

PI-2077-040110

G08A

SMD-8 (G Package)

D S .004 (.10)

E S .010 (.25)

-E-

-D-

-F-

DIM

Inches

0.367-0.387

0.240-0.260

0.125-0.145

0.004-0.012

0.036-0.044

0.057-0.068

0.048-0.053

0.032-0.037

0.007-0.011

0.010-0.012

0.100 BSC

0.030 (MIN)

0.372-0.388

0-8°

9.32-9.83

6.10-6.60

3.18-3.68

0.10-0.30

0.91-1.12

1.45-1.73

1.22-1.35

0.81-0.94

0.18-0.28

0.25-0.30

2.54 BSC

0.76 (MIN)

9.45-9.86

0-8°

Notes:

1. Package dimensions conform to JEDEC

specification MS-001-AB (issue B, 7/85)

except for lead shape and size.

2. Controlling dimensions are inches.

3. Dimensions shown do not include mold

flash or other protrusions. Mold flash or

protrusions shall not exceed .006 (.15) on

any side.

4. D, E and F are reference datums on the

molded body.

.004 (.10)

.010 (.25) M A S

.420

.046 .060 .060 .046

.080

Pin 1

.086

.186

.286

Solder Pad Dimensions

Rev. T 12/12

DPA422-426

www.powerint.com

.165 (4.19)

.185 (4.70)

R07C

TO-263-7C

-A-

.580 (14.73)

.620 (15.75)

.396 (10.06)

.415 (10.54)

.326 (8.28)

.336 (8.53)

.055 (1.40)

.066 (1.68)

.100 (2.54)

Ref.

.208 (5.28)

Ref.

.026 (0.66)

.032 (0.81)

.225 (5.72)

min.

.245 (6.22)

min.

.000 (0.00)

.010 (0.25)

.090 (2.29)

.110 (2.79)

.017 (0.43)

.023 (0.58)

°8 -°0

.045 (1.14)

.055 (1.40)

.050 (1.27)

Notes:

1. Package Outline Exclusive of Mold Flash & Metal Burr.

2. Package Outline Inclusive of Plating Thickness.

3. Foot Length Measured at Intercept Point Between

Datum A Lead Surface.

4. Controlling Dimensions are in Inches. Millimeter

Dimensions are shown in Parentheses.

.004 (0.10)

.315 (8.00)

.128 (3.25)

.038 (0.97)

.050 (1.27)

.380 (9.65)

.638 (16.21)

Rev. T 12/12

DPA422-426

www.powerint.com

Revision Notes Date

F Final release data sheet. 6/02

G Updated Figure 25 and text description. 9/02

H Corrected missing text on page 9 and corrected Table 4.

Updated R package description.

Revised thermal impedances (qJA), DCMIN, VF, IF, and BVDSS.

Updated Figure 25 and description in text.

4/03

I Corrected text errors on pp. 1, 7 and 20. 5/03

J Figure 25 and description in text updated. 5/03

K Added P and G packages. 1/04

L Corrected Figure 3. 4/04

M Added package information to Table 1.

Revised Figure 13.

Added lead-free ordering information.

12/04

N Minor error corrections. 2/05

O Added S-PAK. 7/05

P Added notes to Table 6. 7/05

Q Updated Figure 27 to best reﬂect current requirements for PoE. 5/06

R Added DPA422. 2/07

S Removed S-Pack. 12/07

T Added Note 8, page 1. 12/12

For the latest updates, visit our website: www.powerint.com

Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power

Integrations does not assume any liability arising from the use of any device or circuit described herein. POWER INTEGRATIONS MAKES

NO WARRANTY HEREIN AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED

WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.

Patent Information

The products and applications illustrated herein (including transformer construction and circuits external to the products) may be covered

by one or more U.S. and foreign patents, or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A

complete list of Power Integrations patents may be found at www.powerint.com. Power Integrations grants its customers a license under

certain patent rights as set forth at http://www.powerint.com/ip.htm.

Life Support Policy

POWER INTEGRATIONS PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:

1. A Life support device or system is one which, (i) is intended for surgical implant into the body, or (ii) supports or sustains life, and (iii)

whose failure to perform, when properly used in accordance with instructions for use, can be reasonably expected to result in signiﬁcant

injury or death to the user.

2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause

the failure of the life support device or system, or to affect its safety or effectiveness.

The PI logo, TOPSwitch, TinySwitch, LinkSwitch, LYTSwitch, DPA-Switch, PeakSwitch, CAPZero, SENZero, LinkZero, HiperPFS, HiperTFS,

HiperLCS, Qspeed, EcoSmart, Clampless, E-Shield, Filterfuse, StakFET, PI Expert and PI FACTS are trademarks of Power Integrations, Inc.

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