ISL6545CBZ-T - INTERSIL - Datasheet.Directory

FN6305.6

ISL6545, ISL6545A

5V or 12V Single Synchronous Buck

Pulse-Width Modulation (PWM) Controller

The ISL6545, ISL6545A makes simple work out of

implementing a complete control and protection scheme for a

DC/DC stepdown converter driving N-Channel MOSFETs in a

synchronous buck topology. Since it can work with either 5V or

12V supplies, this one IC can be used in a wide variety of

applications within a system. The ISL6545, ISL6545A

(hereafter referred to as “ISL6545x”, except as needed)

integrat es the contro l, ga te drivers, output adjustment,

monitoring and protection functions into a single 8 Ld SOIC or

10 Ld DFN package.

The ISL6545x provides single feedback loop, voltage-mode

control with fast transient response. The output voltage can be

precisely regulated to as low as 0.6V, with a maximum

tolerance of ±1.0% over-temperature and line voltage

variations. A selectable fixed frequency oscillator (ISL6545 for

300kHz; ISL6545A for 600kHz) reduces design complexity,

while balancing typical application cost and efficiency.

The error amplifier features a 20MHz gain-bandwidth

product and 9V/µs slew rate which enables high converter

bandwidth for fast transient performance. The resulting

PWM duty cycles range from 0% to 100%.

Protection from overcurrent conditions is provided by

monitoring the rDS(ON) of the lower MOSFET to inhibit PWM

operation appropriate ly. This approach simplifies the

implementation and improves efficiency by eliminating the

need for a current sense resistor.

Pinout ISL6545, ISL6545A

(8 LD SOIC)

TOP VIEW

ISL6545, ISL6545A

(10 LD 3x3 DFN)

TOP VIEW

Features

• Operates from +5V or +12V Supply Voltage (for bias)

- 1.0V to 12V VIN Input Rang e (up to 20V possible with

restrictions; see Input V o ltage Considerations)

- 0.6V to VIN Output Range

- Integrated Gate Drivers use VCC (5V to 12V)

- 0.6V Internal Reference; ± 1.0% tolerance

• Simple Single-Loop Control Design

- Voltage-Mode PWM Control

- Drives N-Channel MOSFETs

- Traditional Dual Edge Modulator

• Fast Tr ansient Response

- High-Band width Error Amplifier

- Full 0% to 100% Duty Cycle

• Lossless, Programmable Overcurrent Protection

- Uses Lower MOSFET’s rDS(ON)

• Small Converter Size in 8 Ld SOIC or 10 Ld DFN

- 300kHz or 600kHz Fixed Freq uency Oscillator

- Fixed Internal Soft-Start, Capable into a Pre-biased

Load

- Integrated Boot Diode

- Enable/Shutdown Fun c tion on COMP/SD Pin

- Output Current Sourcing and Sinking

• Pb-Free (RoHS Compliant)

Applications

• Power Supplies for Microprocessors or Peripherals

- PCs, Embedded Controllers, Memory Supplies

- DSP and Core Communications Processor Supplies

• Subsystem Power Supplies

- PCI, AGP; Graphics Cards; Digital TV

- SSTL-2 and DDR/DDR2/DDR3 SDRAM Bus

Termination Supply

• Cable Modems, Set Top Boxes, and DSL Modems

• Industrial Power Supplies; General Purpose Supplies

• 5V or 12V-Input DC/DC Regulators

• Low-Voltage Distributed Power Supplies

UGATE

GND

PHASE

VCC

COMP/SD

BOOT

LGATE/OCSET

BOOT

UGATE

N/C

GND

PHASE

COMP/SD

N/C

LGATE/OCSET

6VCC

GND

Data Sheet March 3, 2011

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.

All other trademarks mentioned are the property of their respective owners.

2FN6305.6

March 3, 2011

Ordering Information

PART NUMBER

(Notes 1, 2, 3) PART MARKING

FIXED FREQUENCY

OSCILLATOR

(kHz) TEMP. RANGE

(°C) PACKAGE

(Pb-Free) PKG.

DWG. #

ISL6545CBZ 6545 CBZ 300 0 to +70 8 Ld SOIC M8.15

ISL6545ACBZ 6545 ACBZ 600 0 to +70 8 Ld SOIC M8.15

ISL6545IBZ 6545 IBZ 300 -40 to +85 8 Ld SOIC M8.15

ISL6545AIBZ 6545 AIBZ 600 -40 to +85 8 Ld SOIC M8.15

ISL6545CRZ 545Z 300 0 to +70 10 Ld DFN L10.3x3C

ISL6545ACRZ 45AZ 600 0 to +70 10 Ld DFN L10.3x3C

ISL6545IRZ 45IZ 300 -40 to +85 10 Ld DFN L10.3x3C

ISL6545AIRZ 5ARZ 600 -40 to +85 10 Ld DFN L10.3x3C

NOTES:

1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.

2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte

tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil

Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of

IPC/JEDEC J STD-020.

3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6545, ISL6545A. For more information on MSL please see

techbrief TB363.

ISL6545, ISL6545A

3FN6305.6

March 3, 2011

Block Diagram

Typical Application

OSCILLATOR

INHIBIT

PWM

COMPARATOR

ERROR

AMP

VCC

PWM

GND

COMP/SD

COMPARATOR

GATE

CONTROL

LOGIC

BOOT

UGATE

PHASE

21.5µA

FIXED 300 (or 600)kHz

LGATE/OCSET

VCC

SOFT-START

POR AND

SAMPLE

AND

HOLD

DBOOT

-DIS

0.4V

INTERNAL

REGULATOR

DIS

0.6V

LGATE/OCSET

20μA

20kΩ

5V int.

VCC

+VO

COMP/SD UGATE

PHASE

BOOT

VCC

GND

LGATE/OCSET

ISL6545,

ROFFSET

LOUT

CBOOT

CBULK

CDCPL

CHF

COUT

VIN

5V OR 12V 1V TO 12V

ROCSET

Type II

compensation

shown

ISL6545A

ISL6545, ISL6545A

4FN6305.6

March 3, 2011

Absolute Maximum Ratings Thermal Information

Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 15V

BOOT Voltage, VBOOT . . . . . . . . . . . . . . . . . . . .GND - 0.3V to 36V

UGATE Voltage VUGATE . . . . . . . .VPHASE - 0.3V to VBOOT + 0.3V

VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT +0.3V

LGATE/OCSET Voltage, VLGATE/OCSET GND - 0.3V to VCC + 0.3V

GND - 5V (<100ns Pulse Width, 2µJ) to VCC + 0.3V

PHASE Voltage, VPHASE . . . . . . . . . .GND - 0.3V to VBOOT + 0.3V

GND - 8V (<400ns, 20µJ) to 30V (<200ns, VBOOT-GND <36V)

Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . . .15V

-0.3V to 16V (<10ns, 10µJ)

Clamp Voltage, VBOOT - VCC . . . . . . . . . . . . . . . . . . . . . . . . . . .24V

FB, COMP/SD Voltage. . . . . . . . . . . . . . . . . . . . . .GND - 0.3V to 6V

ESD Ratings

Human Body Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5kV

Machine Model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .150V

Charged Device Model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.0kV

Thermal Resistance θJA (°C/W) θJC (°C/W)

SOIC Package (Note 4). . . . . . . . . . . . . . 95 N/A

DFN Package (Notes 5, 6). . . . . . . . . . . . 44 5.5

Maximum Junction Temperature

(Plastic Package) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C

Maximum Storage Temperature Range. . . . . . . . . -65°C to +150°C

Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below

http://www.intersil.com/pbfree/Pb-FreeReflow.asp

Operating Conditions

Supply Voltage, VCC . . . . +5V ±10%, +12V ±20%, or 6.5V to 14.4V

Ambient Temperature Range

ISL6545C, ISL6545AC. . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C

ISL6545I, ISL6545AI . . . . . . . . . . . . . . . . . . . . . . .-40°C to +85°C

Junction Temperature Range. . . . . . . . . . . . . . . . . .-40°C to +125°C

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and

result in failures not covered by warranty.

NOTES:

4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.

5. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air , with “direct attach” features. See

Tech Brief TB379 for details.

6. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.

Electrical Specifications VCC = 12V, TJ = 0 to +85°C. Boldface limits apply over the operating temperature range,

-40°C to +85°C.

PARAMETER SYMBOL TEST CONDITIONS MIN

(Note 7) TYP MAX

(Note 7) UNITS

VCC SUPPLY CURRENT

Input Bias Supply Current IVCC VCC = 12V; disabled 45.2 7mA

POWER-ON RESET

Rising VCC POR Threshold VPOR 3.9 4.1 4.3 V

VCC POR Threshold Hysteresis 0.30 0.35 0.40 V

OSCILLATOR

Switching Frequency fOSC ISL6545C 270 300 330 kHz

ISL6545I 240 300 330 kHz

fOSC ISL6545AC 540 600 660 kHz

ISL6545AI 510 600 660 kHz

Ramp Amplitude ΔVOSC 1.5 VP-P

REFERENCE

Reference Voltage Tolerance ISL6545C -1.0 -+1.0 %

ISL6545I -1.5 -+1.5 %

Nominal Reference Voltage VREF 0.600 V

ERROR AMPLIFIER

DC Gain GAIN 96 dB

Gain-Bandwidth Product GBWP 20 MHz

Slew Rate SR 9 V/µs

GATE DRIVERS

Upper Gate Source Impedance RUG-SRCh VCC = 14.5V; I = 50mA 3.0 Ω

Upper Gate Sink Impedance RUG-SNKh VCC = 14.5V; I = 50mA 2.7 Ω

Lower Gate Source Impedance RLG-SRCh VCC = 14.5V; I = 50mA 2.4 Ω

Lower Gate Sink Impedance RLG-SNKh VCC = 14.5V; I = 50mA 2.0 Ω

ISL6545, ISL6545A

5FN6305.6

March 3, 2011

Functional Pin Description (SOIC, DFN)

VCC (SOIC Pin 5, DFN Pin 6)

This pin provides the bias supply for the ISL6545x, as well

as the lower MOSFET’s gate, and the BOOT voltage for the

upper MOSFET’s gate. An internal 5V regulator will supply

bias if VCC rises above 6.5V (but the LGATE/OCSET and

BOOT will still be sourced by VCC). Connect a well-

decoupled 5V or 12V supply to this pin.

FB (SOIC Pin 6, DFN Pin 8)

This pin is the inverting input of the internal error amplifier. Use

FB, in combination with the COMP/SD pin, to compensate the

voltage-control feedback loop of the converter. A resistor divider

from the output to GND is used to set the regulation voltage.

GND (SOIC Pin 3, DFN Pin 4)

This pin represents the signal and power ground for the IC.

Tie this pin to the ground island/plane through the lowest

impedance connection available. For the DFN package,

Pin 4 MUST be connected for electrical GND; the metal pad

under the package should also be connected to the GND

plane for thermal conductivity.

PHASE (SOIC Pin 8, DFN Pin 10)

Connect this pin to the source of the upper MOSFET, and

the drain of the lower MOSFET. It is used as the sink for the

UGATE driver, and to monitor the voltage drop across the

lower MOSFET for overcurrent protection. This pin is also

monitored by the adaptive shoot-through protection circuitry

to determine when the upper MOSFET has turned off.

UGATE (SOIC Pin 2, DFN Pin 2)

Connect this pin to the gate of upper MOSFET; it provides

the PWM-controlled gate drive. It is also monitored by the

adaptive shoot-through protection circuitry to determine

when the upper MOSFET has turned off.

BOOT (SOIC Pin 1, DFN Pin 1)

This pin provides ground referenced bias voltage to the upper

MOSFET driver . A bootstrap circuit is used to create a voltage

suitable to drive an N-channel MOSFET (equal to VCC minus

the on-chip BOOT diode voltage drop), with respect to PHASE.

COMP/SD (SOIC Pin 7, DFN Pin 9)

This is a multiplexed pin. During soft-start and normal converter

operation, this pin represents the output of the error amplifier .

Use COMP/SD, in combination with the FB pin, to compensate

the voltage-control feedback loop of the converter.

Pulling COMP/SD low (VDISABLE = 0.4V nomi nal) will

shut-down (disable) the controller, which causes the

oscillator to stop, the LGATE and UGATE outputs to be held

low, and the soft-start circuitry to re-arm. The external

pull-down device will initially need to overcome up to 5mA of

COMP/SD output current. However, once the IC is disabled,

the COMP output will also be disabled, so only a 20µA

current source will continue to draw current.

When the pull-down device is released, the COMP/SD pin

will start to rise, at a rate determined by the 20µA charging

up the capacitance on the COMP/SD pin. When the

COMP/SD pin rises above the VDISABLE trip point, the

ISL6545x will begin a new Initialization and soft-start cycle.

LGATE/OCSET (SOIC Pin 4, DFN Pin 5)

Connect this pin to the gate of the l ower MOSFET; it provides

the PWM-controlled gate drive (from VCC). This pin is also

monitored by the adaptive shoot-through pro tection circuitry to

determine when the l ower MOSFET has turned off.

During a short period of time following Power-On Reset

(POR) or shut-down release, this pin is also used to

determine the overcurrent threshold of the converter.

Connect a res ist or (ROCSET) from this pin to GND. See

“Overcurrent Protection (OCP)” on page 7 for equations. An

overcurrent trip cycles the soft-start function, after two

dummy soft-start time-outs. Some of the text describing the

LGATE function may leave off the OCSET pa rt of the name,

when it is not relevant to the discussion.

N/C (DFN only; Pin 3, Pin 7)

These two pins in the DFN package are No Connect.

Upper Gate Source Impedance RUG-SRCl VCC = 4.25V; I = 50mA 3.5 Ω

Upper Gate Sink Impedance RUG-SNKl VCC = 4.25V; I = 50mA 2.7 Ω

Lower Gate Source Impedance RLG-SRCl VCC = 4.25V; I = 50mA 2.75 Ω

Lower Gate Sink Impedance RLG-SNKl VCC = 4.25V; I = 50mA 2.1 Ω

PROTECTION/DISABLE

OCSET Current Source IOCSET ISL6545C; LGATE/OCSET = 0V 19.5 21.5 23.5 µA

ISL6545I; LGATE/OCSET = 0V 18.0 21.5 23.5 µA

Disable Threshold (COMP/SD pin) VDISABLE 0.375 0.400 0.425 V

NOTE:

7. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.

Electrical Specifications VCC = 12V, TJ = 0 to +85°C. Boldface limits apply over the operating temperature range,

-40°C to +85°C. (Continued)

PARAMETER SYMBOL TEST CONDITIONS MIN

(Note 7) TYP MAX

(Note 7) UNITS

ISL6545, ISL6545A

6FN6305.6

March 3, 2011

Functional Description

Initialization (POR and OCP Sampling)

Figure 1 shows a simplifi ed timing diagram. The

Power-On-Reset (POR) function continually moni to rs the

bias voltage at the VCC pin. Once the rising POR threshold

is exceeded (VPOR ~4V nominal), the POR function initiates

the Overcurrent Protection (OCP) sample and hold

operation (while COMP/SD is ~1V). When the sampling is

complete, VOUT begins the soft-start ramp.

If the COMP/SD pin is held low during power-up, that will just

delay the initialization until it is released, and the COMP/SD

voltage is above the VDISABLE trip point.

Figure 2 shows a typical power-up sequence in more detail.

The initialization starts at T0, when either VCC rises above

VPOR, or the COMP/SD pin is released (after POR). The

COMP/SD will be pulled up by an internal 20µA current

source, but the timing will not begin until the COMP/SD

exceeds the VDISABLE trip point (at t1). The external

capacitance of the disabling device, as well as the

compensation capacitors, will determine how quickly the

20µA current source will charge the COMP/SD pin. With

typical values, it should add a small delay compared to the

soft-start times. The COMP/SD will continue to ramp to ~1V.

From t1, there is a nominal 6.8ms delay, which allows the

VCC pin to exceed 6.5V (if rising up towards 12V), so that

the internal bias regulator can tu rn on cleanly. At the same

time, the LGATE/OCSET pin is initia lized, by disabling the

LGATE driver and drawing IOCSET (nominal 21.5µA)

through ROCSET. This sets up a voltage that will represent

the OCSET trip point. At t2, there is a variable time period for

the OCP sample and hold operation (0ms to 3.4ms nominal;

the longer time occurs with the higher overcurrent setting).

The sample and hold uses a digital counter and DAC to save

the voltage, so the stored value does not degrade, for as

long as the VCC is above VPOR. See “Overcurrent Protection

(OCP)” on page 7 for more details on the equations and

variables. Upon the completion of sample and hold at t3, the

soft-start operation is initiated, and the output voltage ramps

up between t4 and t5.

Soft-Start and Pre-Biased Outputs

Functionally, the soft-start internally ramps the reference on

the non-inverting terminal of the error amp from 0V to 0.6V in

a nominal 6.8ms The output voltage will thus follow the

ramp, from zero to final value, in the same 6.8ms (the actual

ramp seen on the VOUT will be less than the nominal time,

due to some initialization timi ng, between t3 and t4).

The ramp is created digitally, so there will be 64 small

discrete steps. There is no simple way to change this ramp

rate externally, and it is th e same for either frequency

version of the IC (300kHz or 600kHz).

After an initialization period (t3 to t4), the error amplifie r

(COMP/SD pin) is enabled, and begins to regulate the

converter’s output voltage during soft-start. T he oscillator’s

triangular waveform is compared to the ramping error

amplifier voltage. This generates PHASE pulses of

increasing width that charge the output capacitors. When the

internally generated soft-st art voltage exceeds the reference

voltage (0.6V), the soft-start is complete, and the output

should be in regulation at the expected voltage. This method

provides a rapid and controlled output voltage rise; there is

no large inrush current charging the output capacitors. The

entire start-up sequence from POR typically takes up to

17ms; up to 10.2ms for the delay an d OCP sample, and

6.8ms for the soft-start ramp.

Figure 3 shows the normal curve in blue; initialization begins

at t0, and the output ramps between t1 and t2. If the output is

pre-biased to a voltage less than the expected value, as

shown by the magenta curve, the ISL6545x will detect that

condition. Neither MOSFET will turn on until the soft-start

ramp voltage exceeds the output; VOUT starts seamlessly

ramping from there. If the output is pre-biased to a voltage

above the expected value, as in the red curve, neither

MOSFET will turn on until the end of the soft-start, at which

COMP/SD (1V/DIV)

VOUT (1V/D IV)

VCC (2V/DIV)

GND>

FIGURE 1. POR AND SOFT-START OPERATION

~4V POR

FIGURE 2. LGATE/OCSET AND SOFT- START OPERATION

0.4V

COMP/SD (0.25V/DIV)

LGATE/OCSET (0.25V/DIV)

LGATE

STARTS

SWITCHING

3.4ms3.4ms 0 - 3.4ms 6.8ms

VOUT

(0.5V/DIV)

GND>

t0 t1 t2 t3 t4 t5

ISL6545, ISL6545A

7FN6305.6

March 3, 2011

time it will pull the output voltage down to the final value. Any

resistive load connected to the output will help pull down the

voltage (at the RC rate of the R of the load and the C of the

output capacitance).

If the VIN to the upper MOSFET drain is from a different

supply that comes up after VCC, the soft-start would go

through its cycle, but with no output voltage ramp. When VIN

turns on, the output would follow the ramp of the VIN (at

close to 100% duty cycle, with COMP/SD pin >4V), from

zero up to the final expected voltage. If VIN is too fast, there

may be excessive inrush current charging the output

capacitors (only the beginning of the ramp, from zero to

VOUT matters here). If this is not acceptable, then consider

changing the sequencing of the power supplies, or sharing

the same supply, or adding sequencing logic to the

COMP/SD pin to delay the soft-start until the VIN supply is

ready (see “Input Voltage Considerations” on page 9).

If the IC is disabled after soft-start (by pulling COMP/SD pin

low), and then enabled (by releasing the COMP/SD pin),

then the full initializa tion (including OCP sample) will take

place. However, that there is no new OCP sampling during

overcurrent retries.

If the output is shorted to GND during soft-start, the OCP will

handle it, as described in the next section.

Overcurrent Protection (OCP)

The overcurrent function prot ects the converter from a

shorted output by using the lower MOSFET’s on-resistance,

rDS(ON), to monitor the current. A resistor (ROCSET)

programs the overcurrent trip level (see “T ypical Application”

on page 3). This method enhances the converter’s efficiency

and reduces cost by eliminating a current sensing resistor. If

overcurrent is detected, the output immediately shuts off, it

cycles the soft-start function in a hiccup mode (2 dummy

soft-start time-outs, then up to one real one) to provide fault

protection. If the shorted condition is not removed, this cycle

will continue indefinitely.

Following POR (and 6.8ms delay), the ISL6545x initiates the

Overcurrent Protection sample and hold operation. The

LGATE driver is disabled to allow an internal 21.5µA current

source to develop a voltag e across ROCSET. The ISL6545x

samples this voltage (which is referenced to the GND pin) at

the LGATE/OCSET pin, and holds it in a counter and DAC

combination. This sampled voltage is held internally as the

Overcurrent Set Point, for as long as power is applied , or

until a new sample is taken after coming out of a shut-down.

The actual monitoring of the lower MOSFET’s on-resistance

starts 200ns (nominal) after the edge of the internal PWM

logic signal (that creates the rising external LGATE signal).

This is done to allow the gate transition noise and ringing on

the PHASE pin to settle out before monitoring. The

monitoring ends when the internal PWM edge (and thus

LGATE) goes low. The OCP can be detected anywhere

within the above window.

If the regulator is running at high UGA TE duty cycles (around

75% for 600kHz or 87% for 300kHz operation), then the

LGATE pulse w idth may not be wide enough for the OCP to

properly sample the rDS(ON). For those cases, if the LGATE

is too narrow (or not there at all) for 3 consecutive pulses,

then the third pulse will be stretched and/or inserted to the

425ns minimum width. This allows for OCP monitoring every

third pulse under this condition. This can introduce a small

pulse-width error on the output voltage, which will be

corrected on the next pulse; and the output ripple voltage will

have an unusual 3-clock pattern, which may look like jitter.

This is not necessarily a problem; it is more of a compromise

to maintain OCP at the higher duty cycles. If the OCP is

disabled (by choosing a too-high value of ROCSET, or no

resistor at all), then the pulse stretching feature is also

disabled. Figure 4 illustrates the LGATE pulse width

stretching, as the width gets smaller.

FIGURE 3. SOFT-START WITH PRE-BIAS

GND> VOUT NORMAL

GND>GND>

VOUT PRE-BIASED

VOUT OVERCHARGED

t0 t1 t2

FIGURE 4. LGATE PULSE STRETCHING

> 425 ns

= 425 ns

< 425 ns

<< 425 ns

ISL6545, ISL6545A

8FN6305.6

March 3, 2011

The overcurrent function will trip at a peak inductor current

(IPEAK) determined by Equation 1:

where IOCSET is the internal OCSET current source (21.5µA

typical). The scale factor of 2 doubles the trip point of the

MOSFET voltage drop, compared to the setting on the

ROCSET resistor . The OC trip point varies in a system mainly

due to the MOSFET’s rDS(ON) variations (over process,

current and temperature). To avoid overcurrent tripping in

the normal operating load range, find the ROCSET resistor

from Equation 1 with:

1. The maximum rDS(ON) at the highest junction

temperature.

2. The minimum IOCSET from the specification table.

3. Determine IPEAK for

where ΔI is the output inductor ripple current.

For an equation for the ripple current see “Output Inductor

Selection” on page 12.

The range of allowable voltages detected

(2*IOCSET*ROCSET) is 0mV to 475mV; but the practical

range for typical MOSFETs is typically in the 20mV to 120mV

ballpark (500Ω to 3000Ω). If the voltage drop across

ROCSET is set too low, th at can cause almost co ntinuous

OCP tripping and retry. It would also be very sensitive to

system noise and inrush current spikes, so it should be

avoided. The maximum usable setting is around 0.2V across

ROCSET (0.4V across the MOSFET); values above that

might disable the protection. Any voltage drop across

ROCSET that is greater than 0.3V (0.6V MOSFET trip point)

will disable the OCP. The preferred method to disable OCP

is simply to remove the resistor; that will be detected that as

no OCP.

Note that conditi o ns du ri n g pow e r-u p or during a retry may

look different than normal opera ti o n . Du ri ng po w e r-up in a

12V system, the IC starts operation just above 4V; if the

supply ramp is slow, the soft-start ramp might be over well

before 12V is reached. So with lower gate drive voltages, the

rDS(ON) of the MOSFETs will be higher during power-up,

effectively lowering the OCP trip. In addition, the ripple

current will likely be different at lower input voltage.

Another factor is the digital nature of the soft-start ramp. On

each discrete voltage step, there is in effect a small load

transient, and a current spike to charge the output

capa citors. The hei ght of the curre nt spike is not controlled; it

is affected by the step size of the output, the value of the

output capacitors, as well as the IC error amp compensation.

So it is possible to trip the overcurrent with in-rush current, in

addition to the normal load and ripple considerations.

Figure 5 shows the output response duri ng a retry of an

output shorted to GND. At time t0, the output ha s been

turned off, due to sensing an overcurrent condition. There

are two internal soft-start delay cycles (t1 and t2) to allow the

MOSFETs to cool down, to keep the average power

dissipation in retry at an acceptable level. At time t2, the

output starts a normal soft-start cycle, and the output tries to

ramp. If the short is still applied, and the current reaches the

OCSET trip point any time during soft-start ramp period, the

output will shut off, and return to time t0 for another delay

cycle. The retry period is thus two dummy soft-start cycles

plus one variable one (which depends on how long it takes to

trip the sensor each time). Figure 5 shows an example

where the output gets about half-way up before shutting

down; therefore, the retry (or hiccup) time will be around

17ms. The minimum should be nominally 13.6ms and the

maximum 20.4ms. If the short condition is finally removed,

the output should ramp up normally on the next t2 cycle.

Starting up into a sh orted l oad lo oks the same as a retry int o

that same shorted load. In both cases, OCP is always

enabled during soft-start; once it trips, it will go into retry

(hiccup) mode. The retry cycle will always have two dummy

time-outs, plus whatever fraction of the real soft-start time

passes before the detection and shuto ff; at that point, the

logic immediately starts a new two dummy cycle time-out.

Output Voltage Selection

The output voltage can be programmed to any level between

the 0.6V internal reference, up to the VIN supply. The

ISL6545x can run at near 100% duty cycle at zero load, but

the rDS(ON) of the upper MOSFET will effectively limit it to

something less as the load current increases. In addition, the

OCP (if enabled) will also limit the maximum effective duty

cycle.

An external resistor divider is used to scale the output

voltage relative to the internal reference voltage, and feed it

back to the inverting input of the error amp. See “Typical

IPEAK 2I

OCSET

×xROCSET

rDS ON()

-----------------------------------------------------------= (EQ. 1)

IPEAK IOUT MAX()

ΔI()

----------

FIGURE 5. OVERCURRENT RETRY OPERATION

6.8ms6.8ms 0ms to 6.8ms 6.8ms

t0 t1 t2 t0

OUT

(0.5V/DIV)

GND>

INTERNAL SOFT-START RAMP

ISL6545, ISL6545A

9FN6305.6

March 3, 2011

Application” on page 3 for more detail; RS is the upper

resistor; ROFFSET (shortened to RO below) is the lower one.

The recommended value for RS is 1kΩ to 5kΩ (±1% for

accuracy) and then ROFFSET is chosen accord ing to the

equation below. Since RS is part of the compensation circuit

(see “Feedback Compensatio n” on page 10), it is often

easier to change ROFFSET to change the output voltage;

that way the compensation calculations do not need to be

repeated. If VOUT = 0.6V, then ROFFSET can be left open.

Output voltages less than 0.6V are not available as shown in

Equation 2.

Input Voltage Considerations

The Typical Application diagram on page 3 shows a

standard configuration where VCC is either 5V (±10% ) or

12V (±20%); in each case, the gate drivers use the VCC

voltage for LGATE and BOOT/UGATE. In addition, VCC is

allowed to work anywhere from 6.5V up to the 14.4V

maximum. The VCC range between 5.5V and 6.5V is NOT

allowed for long-term reliability reasons, but transitions

through it to voltages above 6.5V are acceptable.

There is an internal 5V regulator for bias; it turns on between

5.5V and 6.5V ; some of the delay af ter POR is there to allow

a typical power supply to ramp up past 6.5V before the

soft-start ramps begins. This prevents a disturbance on the

output, due to the internal regulator turning on or off. If the

transition is slow (not a step change), the disturbance should

be minimal. So while the recommendation is to not have the

output enabled during the transition through this region, it

may be acceptable. The user should monitor the output for

their application, to see if there is any problem.

The VIN to the upper MOSFET can share the same supply

as VCC, but can also run off a separate supply or other

sources, such as outputs of other regulators. If VCC powers

up first, and the VIN is not present by the time the

initialization is done, then the soft-start will not be able to

ramp the output, and the output will later follow part of the

VIN ramp when it is applied. If this is not desired, then

change the sequencing of the supplies, or use the

COMP/SD pin to disable VOUT until both supplies are ready.

Figure 6 shows a simple sequencer for this situation. If VCC

powers up first, Q1 will be off, and R3 pulling to VCC will turn

Q2 on, keeping the ISL6545x in shut-down. When VIN turns

on, the resistor divider R1 and R2 determines when Q1 turns

on, which will turn off Q2, and release the shut-down. If VIN

powers up first, Q1 will be on, turning Q2 off; so the

ISL6545x will start-up as soon as VCC comes up. The

VDISABLE trip point is 0.4V nominal, so a wide variety of

NFET’s or NPN’ s or even some logic IC’s can be used as Q1

or Q2; but Q2 must be low leakage when off (open-drain or

open-collector) so as not to interfere with the COMP output.

Q2 should also be placed near the COMP/SD pin.

The VIN range can be as low as ~1V (for VOUT as low as the

0.6V reference). It can be as high as 20V (for VOUT just

below VIN). There are some restrictions for running high VIN

voltage.

The first consideration for high VIN is the maximum BO OT

voltage of 36V. The VIN (as seen on PHASE) plus VCC (boot

voltage - minus the diode drop), plus any ringing (or other

transients) on the BOOT pin must be less than 36V. If VIN is

20V, that limits VCC plus ringing to 16V.

The second consideration for high VIN is the maximum

(BOOT - VCC) voltage; this must be less than 24V. Since

BOOT = VIN + VCC + ringing, that reduces to (VIN + ringing)

must be <24V. So based on typical circuits, a 20V maximum

VIN is a good starting assumption; the user should verify the

ringing in their particular application.

Another consideration for high VIN is duty cycle. Very low

duty cycles (such as 20V in to 1.0V out, for 5% duty cycle)

require component selection compatible with that choice

(such as low rDS(ON) lower MOSFET, and a good LC output

filter). At the other extreme (for example, 20V in to 12V out),

the upper MOSFET needs to be low rDS(ON). In addition, if

the duty cycle gets too high, it can affect the overcurrent

sample time. In all cases, the input and output capacitors

and both MOSFETs must be rated for the voltages present.

Switching Frequency

The switching frequency is either a fixed 300kHz or 600kHz,

depending on the part number chosen (ISL6545 is 300kHz;

ISL6545A is 600kHz). However , all of the other timing

mentioned (POR delay, OCP sample, soft-start, etc.) is

independent of the clock frequency, unless otherwise noted.

BOOT Refresh

In the event that the UGATE is on for an extended period of

time, the charge on the boot capacitor can start to sag,

raising the rDS(ON) of the upper MOSFET. T he ISL6545x

has a circuit that detects a long UGAT E on-time (nominal

100µs), and forces the LGA TE to go high for one clock cycle,

which will allow the boot capacitor some time to recharge.

Separately, the OCP circuit has an LGATE pulse stretcher

(to be sure the sample time is long enough), which can also

help refresh the boot. But if OCP is disabled (no current

VOUT 0.6V RSRO

+()

---------------------------

•=

RORS0.6V•

VOUT 0.6V–

----------------------------------

=(EQ. 2)

FIGURE 6. SEQUENCER CIRCUIT

VIN

VCC

to COMP/SD

ISL6545, ISL6545A

10 FN6305.6

March 3, 2011

sense resistor), the regular boot refresh circuit will still be

active.

Current Sinking

The ISL6545x incorporates a MOSFET shoot-throug h

protection method which allows a converter to sink current

as well as source current. Care should be exercised when

designing a converter with the ISL6545x when it is known

that the converter may sink current.

When the converter is sinking current, it is behaving as a

boost converter that is regulating its input voltage. This

means that the converter is boosting current into the VCC

rail, which supplies the bias voltage to the ISL6545x. If there

is nowhere for this current to go, such as to other distributed

loads on the VCC rail, through a voltage limiting prote c tion

device, or other methods, the capacitance on the VCC bus

will absorb the current. This situation will allow voltage level

of the VCC rail to increase. If the voltage level of the rail is

boosted to a level that exceeds the maximum voltage rating

of the ISL6545x, then the IC will experience an irreversible

failure and the converter will no longer be operational.

Ensuring that there is a path for the current to follow other

than the capacitance on the rail will prevent this failure

mode.

Application Guidelines

Layout Considerations

As in any high frequency switching converter, layout is very

important. Switching current from one power device to another

can generate voltage transients across the impedances of the

interconnecting bond wires and circuit traces. These

interconnecting impedances should be minimized by using

wide, short printed circuit traces. The critical components

should be located as close together as possible, using ground

plane construction or single point grounding.

Figure 7 shows the critical power components of the converter .

To minimize the voltage overshoot, the interconnecting wires

indicated by heavy lines should be part of a ground or power

plane in a printed circuit board. The components shown should

be located as close together as possible. Please note that the

capacitors CIN and CO may each represent numerous physical

ca pac i t o r s . For b e s t r e s u l ts, l o c ate the ISL6545x w i t h i n 1 inch

of the MOSFETs, Q1 and Q2. The circuit traces for the

MOSFET gate and source connections from the ISL6545x

must be sized to handle up to 1A peak current.

Figure 8 shows the circuit traces that require add itional

layout consideration. Use single point and ground plane

construction for the circuits shown. Minimize any leakage

current paths on the COMP/SD pin and locate the resistor,

ROSCET close to the COMP/SD pin because the internal

current source is only 20µA. Provide local VCC decoupling

between VCC and GND pi ns. Locate the capacitor, CBOOT

as close as practical to the BOOT and PHASE pins. All

components used for feedback compensation (not shown)

should be located as close to the IC as practical.

Feedback Compensation

This section highlights the design consideration for a

voltage-mode controller requiring external compensation. To

address a broad range of applications, a type-3 feedback

network is recommended, as shown in the top part of

Figure 9.

Figure 9 also highlights the voltage-mode control loop for a

synchronous-rectified buck converter, applicable to the

ISL6545x circuit. The output voltage (VOUT) is regulated to the

reference vol tag e, VREF. The error amplifier output (COMP pin

voltage) is compared with the oscillator (OSC) modified

sawtooth wave to provide a pulse-width modulated wave with

an amplitude of VIN at the PHASE node. The PWM wave is

smoothed by the output filter (L and C). The output filter

capacitor bank’s equivalent series resistance is represented by

the series resistor E.

LGATE/OCSET

UGATE

PHASE

VIN

VOUT

RETURN

ISL6545,

CIN

LOAD

FIGURE 7. PRINTED CIRCUIT BOARD POWER AND

GROUND PLANES OR ISLANDS

ISL6545A

FIGURE 8. PRINTED CIRCUIT BOARD SMALL SIGNAL

LAYOUT GUIDELINES

+VCC

ISL6545,

LGATE/OCSET

GND

VCC

BOOT

VOUT

LOAD

PHASE

+VIN

CBOOT

CVCC

ROCSET

ISL6545A

ISL6545, ISL6545A

11 FN6305.6

March 3, 2011

The modulator transfer function is the small-signal transfer

function of VOUT/VCOMP. This function is dominated by a DC

gain, given by dMAXVIN/VOSC, and shaped by the output

filter, with a double pole break freque ncy at FLC and a zero at

FCE. For the purpose of this analysis, L and D represent the

channel induct ance and its DCR, whi le C and E represent the

total output cap acit an ce and it s e quivalent series resist ance.

The compensation network consists of the error amplifier

(internal to the ISL6545x) and the external R1-R3, C1-C3

components. The goal of the compensation network is to

provide a closed loop transfer function with high 0dB crossing

frequency (F0; typic ally 0.1 to 0.3 of FSW) and adequate phase

margin (better than 45°). Phase margin is the difference

between the closed loop phase at F0dB and 180°. The

equations that follow relate the compensation network’s poles,

zeros and gain to the components (R1, R2, R3, C1, C2, and

C3) in Figure 9. Use the following guidelines for locating the

poles and zeros of the compensation network:

1. Select a value for R1 (1kΩ to 5kΩ, typically). Calculate

value for R2 for desired converter bandwidth (F0). If

setting the output voltage via an offset resistor connected

to the FB pin, Ro in Figure 9, the design procedure can

be followed as presented.

2. Calculate C1 such that FZ1 is placed at a fraction of the FLC,

at 0.1 to 0.75 of FLC (to adjust, change the 0.5 factor to

desired number). The higher the quality factor of the output

filter and/or the higher the ratio FCE/FLC, the lower the FZ1

frequency (to maximize phase boost at FLC).

3. Calculate C2 such that FP1 is placed at FCE.

4. Calculate R3 such that FZ2 is placed at FLC. Calculate C3

such that FP2 is placed below FSW (typically, 0.5 to 1.0

times FSW). FSW represents the switching frequency.

Change the numerical factor to reflect desired placement

of this pole. Placement of FP2 lower in frequency helps

reduce the gain of the compensation network at high

frequency, in turn reducing the HF ripple compo nent at

the COMP pin and minimizing resultant duty cycle jitter.

It is recommended a mathematical model is used to plot the

loop response. Check the loop gain against the error

amplifier’s open-loop gain. Verify phase ma rgin results and

adjust as necessary. Th e equations shown in Equations 8

and 9 describe the frequency response of the modulator

(GMOD), feedback compensation (GFB) and closed-loop

response (GCL):

COMPENSATION BREAK FREQUENCY EQUATIONS

FIGURE 9. VOL T AGE-MODE BUCK CONVERTER

COMPENSATION DESIGN

E/A

VREF

COMP C1

R3 C3

VIN

PWM

CIRCUIT

HALF-BRIDGE

DRIVE

OSCILLATOR

EXTERNAL CIRCUIT

ISL6545x

VOUT

VOSC

UGATE

LGATE

PHASE

FLC 1

2πLC⋅⋅

---------------------------

=FCE 1

2πCE⋅⋅

------------------------

=(EQ. 3)

R2 VOSC R1 F0

⋅⋅

dMAX VIN FLC

⋅⋅

---------------------------------------------

=(EQ. 4)

C1 1

2πR2 0.5 FLC

⋅⋅⋅

------------------------------------------------

=(EQ. 5)

C2 C1

2πR2 C1 FCE 1–⋅⋅⋅

---------------------------------------------------------

=(EQ. 6)

R3 R1

FSW

FLC

------------ 1–

----------------------

=C3 1

2πR3 0.7 FSW

⋅⋅⋅

-------------------------------------------------

(EQ. 7)

GMOD f() dMAX VIN

⋅

VOSC

------------------------------1sf() EC⋅⋅+

1sf() ED+()C⋅⋅s2f() LC⋅⋅++

----------------------------------------------------------------------------------------

⋅=

GFB f() 1sf() R2 C1⋅⋅+

sf() R1 C1 C2+()⋅⋅

------------------------------------------------------ ⋅=

1sf() R1 R3+()C3⋅⋅+

1sf() R3 C3⋅⋅+()1sf() R2 C1 C2⋅

C1 C2+

----------------------

⎝⎠

⎛⎞

⋅⋅+

⎝⎠

⎛⎞

⋅

-----------------------------------------------------------------------------------------------------------------------------

⋅

GCL f() GMOD f() GFB f()⋅=where s f(),2πfj⋅⋅=

(EQ. 8)

FZ1 1

2πR2 C1⋅⋅

--------------------------------

FZ2 1

2πR1 R3+()C3⋅⋅

---------------------------------------------------

FP1 1

2πR2 C1 C2⋅

C1 C2+

----------------------

⋅⋅

-----------------------------------------------

FP2 1

2πR3 C3⋅⋅

--------------------------------

(EQ. 9)

ISL6545, ISL6545A

12 FN6305.6

March 3, 2011

Figure 10 shows an asymptotic plot of the DC/DC converter’s

gain vs frequency. The actual Modulator Gain has a high gain

peak dependent on the quality factor (Q) of the output filter ,

which is not shown. Using the above guidelines should yield a

compensation gain similar to the curve plotted. The open loop

error amplifier gain bounds the compensation gain. Check the

compensation gain at FP2 against the capabilities of the error

amplifier . The closed loop gain, GCL, is constructed on the

log-log graph of Figure 10 by adding the modulator gain, GMOD

(in dB), to the feedback compensation gain, GFB (in dB). This is

equivalent to multiplying the modulator transfer function and the

compensation transfer function and then plotting the resulting

gain.

A stable control loop has a gain crossing with close to a

-20dB/decade slope and a phase margin greater than 45°.

Include worst case component variation s when determining

phase margin. The mathematical mo del presented makes a

number of approximations and is genera lly not accurate at

frequencies approachin g or exceedi ng half the switching

frequency. When designing compensation networks, select

target crossover frequencies in the range of 10% to 30% of

the switching frequency, FSW.

This is just one method to calculate compensation

components; there are variations of Equations 3 through 9.

The error amp is similar to that on other Intersil regul ators,

so existing tools can be used here as well. Special

consideration is needed if the size of a ceramic output

capacitance in parallel with bulk capacitors gets too large;

the calculation needs to model them both separately

(attempting to combine two different capacitors types into

one composite component model may not work properly; a

special tool may be needed; contact Intersil at

http://www.intersil.com/contacts/ for assistance.

Component Selection Guidelines

Output Capacitor Select ion

An output capacitor is required to filter the output and supply

the load transient current. The filtering requiremen ts are a

function of the switching frequency and the ripple current.

The load transient requirements are a function of the slew

rate (di/dt) and the magnitude of the transient load current.

These requirements are generally met with a mix of

capacitors and careful layout.

Modern components and loads are ca p able of pro ducing

transient load rates above 1A/ns. High frequency cap acitors

initially supply the transient and slow the current loa d ra te

seen by the bulk capacitors. The b ulk fi lter cap acitor value s

are generally determined by the ESR (Effective Series

Resistance) and volt ag e rati ng requirement s ra th er than

actual capacit a nce requireme nt s.

High frequency decoupling capacitors should be placed as

close to the power pins of the load as physically possible. Be

careful not to add inductance in the circuit board wiring that

could cancel the usefulness of these low inductance

components. Consult with the manufacturer of the load on

specific decoupling requirements.

Use only specialized low-ESR cap aci tors intend ed for

switching-regulator applications for the bulk cap acitors. Th e

bulk capacitor’s ESR will determine the output rippl e voltage

and the initial volt a ge drop af ter a high slew-rate tran sient. An

aluminum electrolytic cap acitor’s ESR value is related to the

case size with lowe r ESR avai lable in larger case sizes.

However, the Equivalent Series Inductance (ESL ) of these

capacitors increases with case size an d can red uce the

usefulness of the capaci tor to high slew-rate transient loading.

Unfortunately, ESL is not a specified parameter. Work with

your capacitor suppl ier and measure the cap acitor’s

impedance with frequency to select a suit a ble component. In

most cases, multiple electrolytic capacitors of small case size

perform better than a single large case capacitor .

Output Inductor Selection

The output inductor is selected to meet the output voltage

ripple requirements and minimize the converter’s response

time to the load transient. The inductor value determines the

converter’s ripple current and the ripple voltage is a function

of the ripple current. The ripple voltage and current are

approximated by the equations shown in Equation 10:

Increasing the value of inductance reduces the ripple current

and voltage. However, the large inductance values reduce

the converter’s response time to a load transient.

One of the parameters limiting the converter’s response to

a load transient is the time required to change the inductor

current. Given a sufficiently fast control loop design, the

ISL6545x will p r o v ide e i ther 0% o r 100% d u t y cy c l e in

response to a load transient. The response time is the time

required to slew the inductor current from an initial current

value to the transient current level. During this interval the

difference between the inductor current and the transient

current level must be supplied by the output capacitor.

FP1

FZ2

OPEN LOOP E/A GAIN

FZ1 FP2

FLC FCE

COMPENSATION GA IN

GAIN

FREQUENCY

MODULATOR GAIN

FIGURE 10. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN

CLOSED LOOP GAIN

20 dMAX V

⋅IN

VOSC

---------------------------------log

20 R2

--------

⎝⎠

⎛⎞

log

LOG

GMOD

GFB

GCL

ΔI=VIN - VOUT

Fsw x L VOUT

VIN ΔVOUT =ΔI x ESR

x(EQ. 10)

ISL6545, ISL6545A

13 FN6305.6

March 3, 2011

Minimizing the response time can minimize the output

capacitance required.

The response time to a transient is different for the

application of load and the removal of load. The equations in

Equation 11 give the approximate response time interva l for

application and removal of a transient load:

where: ITRAN is the transient load current step, tRISE is the

response time to the application of load, and tFALL is the

response time to the removal of load. Th e worst case

response time can be either at the application or removal of

load. Be sure to check both of these equations at the

minimum and maximum output levels fo r the worst case

response time.

Input Capacitor Selection

Use a mix of input bypa ss cap acitors to co ntro l the voltage

overshoot across the MOSFETs. Use small ceramic

capacitors for hi gh fre quency decoupli ng and b ulk cap acito rs

to supply the current nee ded each time Q1 turns on. Place the

small ceramic capacitors physically close to the MOSFETs

and between the drain of Q1 and the source of Q2.

The important parameters for the bulk input capacitor are the

voltage rating and the RMS current rating. For reliable

operation, select the bulk capacitor with voltage and current

ratings above the maximum input voltage and larges t RMS

current required by the circuit. The capacitor voltage rating

should be at least 1.25x greater than the maximum input

voltage and a voltage rating of 1.5x is a conservative

guideline. The RMS current rating requirement for the input

capacitor of a buck regulator is approximately 1/2 the DC

load current.

For a through-hole design, several electrolytic capacitors may

be needed. For surface mount designs, solid tantalum

capacitors can also be used, but caution must be exercised

with regard to the capacitor surge current rating. These

capacitors must be capable of handling the surge current at

power-up. Some capacitor series available from reputable

manufacturers are surge current tested.

MOSFET Selection/Considerations

The ISL6545x requires two N-Channel power MOSFETs.

These should be selected based upon rDS(ON), gate supply

requirements, and thermal management requirements.

In high-current applications, the MOSFET power dissipation,

package selection and heatsink are the dominant design

factors. The power dissipation includes two loss components;

conduction loss and switching loss. The conduction losses are

the largest component of power dissipation for both the upper

and the lower MOSFETs. These losses are distributed between

the two MOSFETs according to duty factor . The switching

losses seen when sourcing current will be different from the

switching losses seen when sinking current. When sourcing

current, the upper MOSFET realizes most of the switching

losses. The lower switch realizes most of the switching

losses when the converter is sinking current (see the

equations in Equation 12). These equations assume linear

vo l tag e - c u r r e nt tr ansi tions a nd do not adequately model

power loss due the reverse-recovery of the upper and lower

MOSFET’s body diode. The gate-charge losse s are

dissipated by the ISL6545x and d o not heat t he MOSFETs.

However, large gate-charge increases the switching interval,

tSW which increases the MOSFET switching losses. Ensure

that both MOSFETs are within their maximum junction

temperature at high ambient temperature by calculating the

temperature rise according to package thermal-resistance

specifications. A separate heatsink may be necessary

depending upon MOSFET power , pa ckage type, ambient

temperature and air flow .

When operating with a 12V power supply for VCC (or down to a

minimum supply voltage of 6.5V), a wide variety of

N-MOSFETs can be used. Check the absolute maximum VGS

rating for both MOSFETs; it needs to be above the highest VCC

voltage allowed in the system; that usually means a 20V VGS

rating (which typically correlates with a 30V VDS maximum

rating). Low threshold transistors (around 1V or below) are not

recommended, for the reasons explained in the next

paragraph.

For 5V only operation, given the reduced available gate bias

voltage (5V), logic-level transistors should be used for both

N-MOSFETs. Look for rDS(ON) ratings at 4.5V. Caution

should be exercised with devices exhibiting ve ry low

VGS(ON) characteristics. The shoot-through protection

present aboard the ISL6545x may be circumvented by these

MOSFETs if they have large parasitic impedences and/or

capacitances that would inhibit the gate of the MOSFET from

being discharged below its threshold level before the

complementary MOSFET is turned on. Also avoid MOSFETs

with excessive switching times; the circuitry is expecting

transitions to occur in under 50ns or so.

tRISE = L x ITRAN

VIN - VOUT tFALL = L x ITRAN

VOUT (EQ. 11)

PLOWER = Io2 x rDS(ON) x (1 - D)

Where: D is the duty cycle = VOUT / VIN,

tSW is the combined switch ON and OFF time, and

FSW is the switching frequency.

Losses while Sourcing Current

Losses while Sinking Current

PLOWER Io2rDS ON()

×1D–()×1

---Io⋅VIN

×tSW FS

××+=

PUPPER Io2rDS ON()

×D×1

---Io⋅VIN

×tSW FS

××+=

PUPPER = Io2 x rDS(ON) x D (EQ. 12)

ISL6545, ISL6545A

All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.

Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality

Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without

notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and

reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result

from its use. No lice nse is gran t ed by i mpli catio n or other wise u nder an y p a tent or patent rights of Inter sil or it s sub sidi aries.

For information regarding Intersil Corporation and its products, see www.intersil.com

FN6305.6

March 3, 2011

BOOTSTRAP Considerations

Figure 11 shows the upper gate drive (BOOT pin) supplied by

a bootstrap circuit from VCC. The boot capacitor, CBOOT,

develops a floatin g sup ply volt age referenced to the PHASE

pin. The supply is refreshed to a voltage of VCC less the boo t

diode drop (VD) each time the lower MOSFET, Q2, turns on.

Check that the voltage rating of th e ca p acitor is above the

maximum VCC voltage in th e system; a 16 V rating should be

sufficient for a 12 V system. A value of 0.1µF i s typica l for

many systems driving single MOSFETs.

If VCC is 12V, but VIN is lower (such as 5V), then another

option is to connect the BOOT pin to 12V, and remove the

BOOT cap (although, you may want to add a local cap from

BOOT to GND). This will make the UGATE VGS voltage

equal to (12V - 5V = 7V). That should be high enough to

drive most MOSFETs, and low enough to improve the

efficiency slightly. Do NOT leave the BOOT pin open, and try

to get the same effect by driving BOOT through VCC and the

internal diode; this path is not designed for the high current

pulses that will result.

For low VCC voltage applications where efficiency is very

important, an external BOOT diode (in parallel with the

internal one) may be considered. Th e external diode drop

has to be lower than the internal one; the resulting higher

VG-S of the upper FET will lower its rDS(ON). The modest

gain in efficiency should be balanced against the extra cost

and area of the external diode.

+VCC

ISL6545x

GND

LGATE/OCSET

UGATE

PHASE

BOOT

VCC

+VIN

VG-S ≈ VCC - VD

VG-S ≈ VCC

CBOOT Q1

FIGURE 11. UPPER GATE DRIVE BOOTSTRAP

VCC

+ VD -

ISL6545, ISL6545A

15 FN6305.6

March 3, 2011

ISL6545, ISL6545A

Package Outline Drawing

L10.3x3C

10 LEAD DUAL FLAT PACKAGE (DFN)

Rev 2, 09/09

located within the zone indicated. The pin #1 indentifier may be

Unless otherwise specified, tolerance : Decimal ± 0.05

Tiebar shown (if present) is a non-functional feature.

The configuration of the pin #1 identifier is optional, but must be

between 0.18mm and 0.30mm from the terminal tip.

Dimension b applies to the metallized terminal and is measured

Dimensions in ( ) for Reference Only.

Dimensioning and tolerancing conform to AMSE Y14.5m-1 994.

either a mold or mark feature.

Dimensions are in millimeters.1.

NOTES:

BOTTOM VIEW

DETAIL "X"

SIDE VIEW

TYPICAL RECOMMENDED LAND PATTERN

TOP VIEW

(4X) 0.10

INDEX AREA

PIN 1

PIN #1 INDEX AREA

SEATING PLANE

BASE PLANE

0.08

SEE DETAIL "X"

0.10 C

10 1

PACKAGE

0.90

0.20

0.50

2.38

3.00

(10x 0.25)

(8x 0.50)

2.38

1.64

(10 x 0.60)

3.00

0.05

0.20 REF

10 x 0.25

10x 0.40

1.64

OUTLINE

MAX

(4X) 0.10 CB

7. COMPLAINT TO JEDEC MO-229-WEED-3 except for E-PAD

dimensions.

16 FN6305.6

March 3, 2011

ISL6545, ISL6545A

Package Outline Drawing

M8.15

8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE

Rev 2, 11/10

DETAIL "A"

TOP VIEW

INDEX

AREA

123

-C-

SEATING PLANE

x 45°

NOTES:

1. Dimensioning and tolerancing per ANSI Y14.5M-1982.

2. Package length does not include mold flash, protrusions or gate burrs.

Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006

inch) per side.

3. Package width does not include interlead flash or protrusions. Interlead

flash and protrusions shall not exceed 0.25mm (0.010 inch) per side.

4. The chamfer on the body is optional. If it is not present, a visual index fea-

ture must be located within the crosshatched area.

5. Terminal numbers are shown for reference only.

6. The lead width as measur ed 0.36mm (0.014 inch) or greater above the seat-

ing plane, shall not exceed a maximum value of 0.61m m (0.024 inch).

7. Controlling dimension: MILLIMETER. Converted inch dimensions are not

necessarily exact.

8. This outline conforms to JEDEC publication MS-012-AA ISSUE C.

SIDE VIEW “A

SIDE VIEW “B”

1.27 (0.050)

6.20 (0.244)

5.80 (0.228)

4.00 (0.157)

3.80 (0.150)

0.50 (0.20)

0.25 (0.01)

5.00 (0.197)

4.80 (0.189) 1.75 (0.069)

1.35 (0.053)

0.25(0.010)

0.10(0.004)

0.51(0.020)

0.33(0.013)

8°

0°

0.25 (0.010)

0.19 (0.008)

1.27 (0.050)

0.40 (0.016)

1.27 (0.050)

0.200

TYPICAL RECOMMENDED LAND PATTERN

2.41 (0.095)

0.76 (0. 030)