AN_201701_PL52_007 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi -resonant PWM controller Author: Jared Huntington Scope and purpose The demo board described in this application note provides a test platform for the new 700 V CoolMOSTM P7 series of high voltage MOSFETs. The adapter uses the ICE2QS03G, a second generation current mode control quasi-resonant flyback controller and an IPA70R600P7S 700 V CoolMOSTM P7 series power MOSFET. This application note is intended for those that have experience with flyback converter designs and will not go in depth regarding the overall design process, but will cover specific design aspects for this controller and 700 V CoolMOSTM P7 in charger and adapter applications. It will also look at the overall benefits that the 700 V CoolMOSTM P7 presents for switch mode power supplies. For a detailed introduction on flyback converter design please read Design guide for QR Flyback converter [1]. Intended audience Power supply design engineers Table of contents 1 Description ................................................................................................................... 2 2 Quasi-resonant flyback overview .................................................................................... 3 3 ICE2QS03G functional overview ...................................................................................... 4 4 P7 700 V CoolMOS benefits for adapters ......................................................................... 5 5 5.1 5.2 Design considerations .................................................................................................... 7 700 V MOSFET and design changes ........................................................................................................ 7 UVLO circuit ........................................................................................................................................... 12 6 6.1 6.2 6.3 6.4 6.5 6.6 6.7 Demo board overview ................................................................................................... 14 Demo board pictures ............................................................................................................................ 14 Demo board specifications ................................................................................................................... 14 Demo board features ............................................................................................................................ 15 Schematic .............................................................................................................................................. 16 BOM with Infineon components in bold ............................................................................................... 17 PCB layout ............................................................................................................................................. 19 Transformer construction ..................................................................................................................... 20 7 7.1 7.2 Measurements .............................................................................................................. 21 High line and low line operation........................................................................................................... 21 Thermal performance under typical operating conditions ................................................................. 11 8 Conclusion ................................................................................................................... 23 9 References ................................................................................................................... 24 TM Application Note www.infineon.com/p7 Please read the Important Notice and Warnings at the end of this document Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Description 1 Description This 40 W adapter demo board is intended to be a form, fit, and function test platform for charger and adapter applications to show the operation of the 700 V CoolMOSTM P7 as well as the overall controller design. The demo board is designed around a quasi-resonant flyback topology for improved switching losses that allows higher power density designs and lower radiated and conducted emissions. A 40 W universal input isolated flyback demo board with a 19 V output based on the ICE2QS03G controller and the CoolMOSTM P7 MOSFET is described in this application note and some test results are presented. Figure 1 40 W flyback demo board Application Note 2 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Quasi-resonant flyback overview 2 Quasi-resonant flyback overview The quasi-resonant (QR) flyback offers improved efficiency and electro-magnetic interference (EMI) performance over the traditional fixed frequency flyback converter by reducing switching losses. This is accomplished by controlling the turn-on time of the primary MOSFET (Qpri in Figure 3). In a flyback operating in discontinuous conduction mode (DCM), the energy is first stored in the primary side when the primary MOSFET (Qpri) is turned on allowing the primary current to ramp up. The primary MOSFET (Qpri) turns-off and the energy stored in the transformer transfers into the secondary side capacitor. The energy that is left in the primary inductance (Lpri) after transferring the energy to the secondary then resonates with the combined output capacitance of the MOSFET(CDS_parasitic) consisting of the MOSFET output capacitance (COSS), stray drain source capacitance from the transformer and layout, and any additional added external drain source capacitance on this node. In a fixed frequency flyback the switch turn-on happens regardless of the MOSFET drain source voltage (VDS). If switching occurs at a higher VDS (Figure 2), this leads to more switching losses (EOSS losses). The QR flyback waits to turn on Qpri until the VDS voltage reaches the minimum possible voltage shown in Figure 2 and then turns-on the MOSFET. 2 _ = 0.5 Since the turn-on switching losses are a function of V2 (as shown above), this reduces the overall system switching losses. This has the added benefit of lowering the amount of switched energy which helps reduce switching noise from the converter, resulting in lower radiated and conducted emissions. The 700 V CoolMOSTM P7 technology generates improvements in the operation of QR flyback converters through having lower output capacitance (COSS) that helps to reduce the losses of the device during turn-on. The improvements that 700 V CoolMOSTM P7 offers will be further addressed in Section 4. MOSFET Turn ON MOSFET Turn ON Figure 2 Fixed frequency flyback primary MOSFET drain source waveform (left) vs. a QR flyback primary MOSFET drain source waveform (right). Figure 3 Simplified flyback schematic Application Note 3 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller functional overview ICE2QS03G 3 ICE2QS03G functional overview The ICE2QS03G PWM controller is a second-generation quasi-resonant flyback controller IC developed by Infineon Technologies. Typical applications include TV-sets, DVD-players, set-top boxes, netbook adapters, home audio, and printer applications. This controller implements switching at the lowest ringing voltage and also includes pulse skipping at light loads for maximum efficiency across a wide range of loads. Figure 4 ICE2QS03G pinout Table 1 ICE2QS03G pin description Pin Name Description 1 Zero Crossing (ZC) Detects the minimum trough (valley) voltage for turn-on for the primary switch turn-on time 2 Feedback (FB) Voltage feedback for output regulation 3 Current Sense (CS) Primary side current sense for short circuit protection and current mode control 4 Gate drive output (GATE) MOSFET gate driver pin 5 High Voltage (HV) Connects to the bus voltage for the initial startup through the high voltage startup cell 6 No Connect (NC) No connection 7 Power supply (VCC) Positive IC for the power supply 8 Ground (GND) Controller ground Application Note 4 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller 700 V CoolMOSTM P7 benefits for adapters TM 4 Figure 5 700 V CoolMOS P7 benefits for adapters The MOSFET drain source voltage in a flyback converter is the sum of the bus voltage (Vbus), reflected voltage (Vreflected), and snubber voltage (Vsnubber). The 700 V CoolMOSTM P7 provides several benefits for charger and adapter applications when compared to 600 V and 650 V MOSFETs. The additional breakdown voltage can be used to increase the efficiency of designs, increase the allowable AC input voltage, or increase the surge capabilities of designs. The P7 family of devices also has better performance when comparing switching losses to previous generations of MOSFETs. A 700 V breakdown voltage allows for a higher combination of bus voltage, reflected voltage, and snubber voltage than can be achieved with 600 V or 650 V devices. This allows for increasing the snubber and reflected voltage in order to lower the switching losses of a converter. It is also possible to use this extra voltage margin to allow for bus voltages extending beyond the typical 265 VAC high line. The device can also be used as a drop in replacement for 600 V and 650 V devices to give additional margin for abnormal conditions such as surge and output short circuit conditions in existing designs that need improved margins. This additional 50 - 100 V of drain source breakdown voltage gives designers more flexibility to improve the overall design. The P7 family of devices also has an improved switching performance that is better than existing Infineon and competitor devices. One switching loss mechanism is the EOSS of the MOSFET. The EOSS is the main loss contributor for the turn-on of the MOSFET in a QR flyback. The energy that is stored in the output capacitance of the MOSFET needs to be discharged every cycle before the MOSFET is turned on. As shown in Figure 6, the output capacitance energy storage of the 700 V CoolMOSTM P7 is better when compared to equivalent competitor devices. This improvement is most significant at higher AC input voltages. Additional details about the 700 V CoolMOSTM P7 device improvements such as the reduced gate charge (Qg), RDS(on) temperature dependency, QOSS, and transfer characteristics can be found in the CoolMOSTM 700V P7 Application note [3]. Application Note 5 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller 700 V CoolMOSTM P7 benefits for adapters 3,0 IPA60R600P6 IPA70R600P7 Competitor 2,5 EOSS (J) 2,0 1,5 1,0 0,5 0,0 0 Figure 6 100 200 VDS (V) 300 400 Eoss comparison of a P6 700 V 600 m MOSFET, a P7 700 V 600 m MOSFET, and a competitor's 700 V 600 m device. It can be seen that the amount of energy stored in the output capacitance during a typical QR high line turn on of 200 V is reduced by 0.5 J every switching cycle, which in a 100 kHz design corresponds to 50 mW. SPICE models of the P7 700 V MOSFETs are provided on the Infineon website. These models have been created with MOSFET characterization data covering different MOSFET parameters and provide a high level of accuracy. Below, Figure 7 shows the difference between the Infineon 40 W adapter's measured waveforms and the simulated waveforms. These models can be used to better understand the loss mechanisms that are responsible for power dissipation in the primary MOSFET of the Flyback converter and help to optimize designs. Figure 7 Simulated switching vs. measured switching at 230 VAC operation in the Infineon 40 W adapter. Application Note 6 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations 5 Design considerations 5.1 700 V MOSFET and design changes This section will compare the Infineon 35 W adapter design using an Infineon 600 V C6 MOSFET with the Infineon 700 V CoolMOSTM P7 based 40 W adapter to show the differences between the two designs. The flyback MOSFET was changed from the CoolMOSTM C6 family of devices to the P7 700 V IPA70R600P7S in order to have the best performance from the latest generation of Infineon devices. With the same RDS(on), the switching characteristics of the MOSFET are improved. In this particular design the peak drain source voltage was increased from 526 V to 560 V by slightly reducing the reflected voltage and increasing the snubber clamp voltage leading to a reduction in snubber energy dissipation. Reducing the switching losses helps to improve the high line and light load efficiency of the 40 W adapter. Even with increasing the drain source voltage, the calculated breakdown voltage margin increased from 12 percent to 20 percent. The output diode was also replaced with a lower cost, lower voltage drop diode in order to reduce the full load power losses. The reflected voltage was decreased from the Infineon 35 W adaptor in order to reduce the snubber losses. The RCD snubber resistor value was also increased to further reduce the snubber loss, which causes the maximum drain source peak voltage to increase. As shown in Table 2, the turns ratio is reduced by 20 V in order to reduce the reflected voltage, which helps to reduce the energy that is dissipated across the snubber network as shown in the equation below: 2 = ( + ) Table 2 Parameter Symbol 600 V design 700 V P7 design Transformer primary turns NP 66 turns 87 turns Transformer secondary turns NS 11 turns 17 turns Output voltage Voutput 19 V 19 V Transformer reflected voltage Vreflected 117 V 97 V The primary side resistor, capacitor, and diode (RCD) snubber network resistor power dissipation was reduced allowing the snubber voltage to reach a higher level and thus lowering the amount of energy that is dissipated in the snubber resistor. This comes into effect especially at lower power levels where the conduction loss is no longer the dominant source of power losses. The snubber clamping voltage can be calculated using the equation below: 1 2 = 2 ( +2 2 - ) Table 3 shows that by increasing the snubber resistor from 54 k to 99 k the snubber voltage increases from 40.1 V to 97 V. Table 3 Parameter Symbol 600 V design 700 V P7 design Leakage inductance Lleakage 25 H 25 H Peak primary current under load at high line Ipri 0.43 A 0.62 A Snubber resistor Rsnubber 54 k 99 k Switching period Ts 28.6 s 28.6 s Application Note 7 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations Parameter Symbol 600 V design 700 V P7 design Snubber voltage Vsnubber 40.1 V 97 V Table 4 sums up the different voltage components of the VDS waveform to arrive at the total calculated peak MOSFET VDS. As shown in Figure 5, the total drain source voltage of the MOSFET is the sum of the bus voltage, the reflected voltage, and the snubber voltage. Even with increasing the overall drain source voltage (VDS) by 34 V we still have an increase in margin from the MOSFET breakdown voltage. In this new design the margin has increased from 12 percent to 20 percent. This increases the overall margin from the MOSFET breakdown voltage while still increasing the peak drain source voltage. In reality, when measured under worst-case conditions this 20 perecent margin corresponds to 599 V or 15 percent margin. The difference between measured and calculated peak voltages comes from the fact that the snubber voltage equation above assumes an infinite snubber capacitor value. Because of this discrepancy, the drain source peak voltage at full load and high line should be empirically verified in the design. Table 4 Parameter Symbol 600 V design 700 V P7 design Primary bus voltage @265 VAC Vbus 373 V 373 V Reflected voltage Vreflected 117 V 97 V Snubber voltage Vsnubber 40.1 V 90 V Drain source voltage maximum VDS_max 526 V 560 V Margin from breakdown voltage VDS_margin 12 % 20 % In order to increase the power level from 35 W to 40 W, the output diode of the power supply needed less power dissipation at full load. A diode with a lower forward voltage drop was selected that improves the output power dissipation by 150 mW, even when operated at a higher output power levels and thus a higher output current. This reduces the temperature sufficiently to allow increasing the total output power. Table 5 Parameter Symbol 600 V design 700 V P7 design Output current Ioutput 1.84 A 2.11 A Diode forward voltage Vforward 0.55 V 0.40 V Diode conduction losses Pdio 1.01 W 0.84 A Application Note 8 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations With all of these changes made to the design, the overall efficiency improvement can be seen in Figure 8 and Figure 9 below. The light load benefits come from the P7 700 V switching loss improvements and RCD snubber changes. The P7 switching loss improvements can be seen by looking at the IPA60R600P6 to IPA70R600P7S delta efficiency curves shown in Figure 10 and Figure 11. The full load efficiency improvements come from changing to a better output diode with a lower forward voltage drop. By making these changes, the power level of the design was increased without switching to a lower RDS(on) device which would have cause an increase in the overall BOM cost. 92,0 91,0 90,0 Efficiency (%) 89,0 88,0 87,0 86,0 5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0 85,0 40W adapter, IPA70R600P7S 40W adapter, IPA60R600P6 40W adapter, competitor 35W adapter, IPD60R600P6 84,0 83,0 82,0 81,0 80,0 Output power (W) Figure 8 40 W adapter efficiency at 120 VAC using IPA70R600P7S efficiency vs. a 35 W adapter design using a P6 device. 92,0 91,0 90,0 89,0 Efficiency (%) 88,0 87,0 86,0 5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0 85,0 84,0 40W adapter, IPA70R600P7S 83,0 40W adapter, IPA60R600P6 40W adapter, competitor 82,0 35W adapter, IPA60R600P6 81,0 80,0 Output power (W) Application Note 9 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations Figure 9 40 W adapter efficiency at 230 VAC using IPA70R600P7S efficiency vs. a 35 W adapter design using a P6 device. Figure 10 and Figure 11 below show the benefits of changing from IPA60R600P7 or a competitor's device to a P7 device. The efficiency graphs below are done as efficiency deltas relative to the P7 IPA70R600P7S in order to make the efficiency differences clearer. The 100 VAC and full load efficiency difference ends up reducing the mold compound temperature by 2.9 C. 0,00 5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0 Efficiency (%) -0,50 -1,00 40W adapter, IPA70R600P7S -1,50 40W adapter, IPA60R600P6 40W adapter, competitor -2,00 Output power (W) Figure 10 Efficiency of the 40 W adapter at 120 VAC showing the P6 and competitor's devices referenced to the P7 MOSFET. 0,10 0,00 5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0 -0,10 Efficiency (%) -0,20 -0,30 -0,40 -0,50 -0,60 -0,70 40W adapter, IPA70R600P7S 40W adapter, IPA60R600P6 40W adapter, competitor -0,80 -0,90 -1,00 Figure 11 Output power (W) Efficiency of the 40 W adapter at 230 VAC showing the P6 and competitor's devices referenced to the P7 MOSFET. Application Note 10 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations 5.2 Thermal performance improvement The worst-case nominal thermal conditions for the system under steady state operation occur at 100 VAC and full output power (40 W). Table 10 shows the thermal improvement of P7 700 V when used in the 40 W adapter at 100 VAC and 40 W of output power due to the improvement in efficiency shown in the previous section. As shown below, the mold compound temperature of the P7 700 V device is 2.9 C lower. Table 11 shows the maximum temperature under the worst-case operating conditions for these components when using the IPA70R600P7S. These component temperatures are the limiting factor for the overall power density of the converter and this can help to increase power density or improve thermal margins in designs. Table 6 Flyback MOSFET thermal rise with 100 VAC, 40 W output Device Temperature rise Temperature referenced to P7 Competition 51.3 C +2.0 C IPA60R600P6 52.2 C +2.9 C IPA70R600P7S 49.3 C 0.0C Table 7 Maximum component thermal rise at 100 VAC, 40W output Ref. Des. Component description Maximum temperature rise Q1 Flyback MOSFET 61.7 C D3 Bridge rectifier 54.9 C L1 Common Mode Choke 59.3 C T1 Flyback transformer 67.8 C D2 Flyback output diode 53.6 C R22, R23, R28 Flyback snubber resistors 79.2 C T1 L1 D3 Figure 12 Q1 D2 100 VAC input, full load, top side. The line filter and bridge rectifier are hottest at this point due to higher AC input currents. Application Note 11 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations R22, R23, R28 Figure 13 100 VAC input, full load, bottom side. The snubber resistors are the hottest components. 5.3 UVLO circuit The under voltage lock out (UVLO) circuit provides a mechanism to shut down the power supply when the AC line input voltage is lower than the specified voltage range. The UVLO event is detected by sensing the voltage level at U2's (TL431) REF pin (VREF_typ = 2.5 V) through the voltage divider resistors (R12, R13, R14, and R17 in Figure 12) from the bulk capacitor C1. Q2 acts as a switch to enter or leave UVLO mode by controlling the FB pin voltage. Q3, together with R17, acts as voltage hysteresis for the UVLO circuit and U2 (TL431) act as a comparator. The system enters the UVLO mode by controlling the FB pin voltage of U1 to 0 V (when the voltage input level goes back to input voltage range), VREF increases to 2.5 V (then switches Q2 and Q3 off) and Vcc hits 18 V, the UVLO mode is released. The calculation for the UVLO circuit is shown below: VREF= 2.5 V R12 = 4.99 M R13 = 4.99 M R14 = 330 k R17 = 681 k _ = _ (12 + 13 + 14) 14 1417 [(14 + 17) + 12 + 13] = 1417 ( ) 14 + 17 _ = 77.8 _ = 114.3 The 'enter UVLO' threshold is set at 77.8 to allow for the BUS capacitance voltage to droop under 90 VAC at full load operation with some margin to avoid false triggering. Application Note 12 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Design considerations Figure 14 Power supply status vs. AC input voltage showing the hysteretic behavior of the UVLO circuit. Application Note 13 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6 Demo board overview 6.1 Demo board pictures Q1 IPA70R600P7S Figure 15 Top side of 40 W Infineon adapter with a TO-220 FullPAK populated 6.2 Demo board specifications Table 8 Section Parameter Specification Input ratings Input voltage 90 VAC - 265 VAC Input frequency 47 Hz - 63 Hz Input current at 100 VAC, 40 W 0.85 A maximum Power factor 0.53 @100 VAC 0.36 @265 VAC Peak efficiency 230 VAC, 40 W Peak efficiency 120 VAC, 40 W 91.3% 89.6% Surge 2 kV IEC61000-4-5 Nominal output voltage 19.0 V Tolerance 2% Output current 2.10 A Output power 40 W Line regulation 0.5% Load regulation 0.5% Output ripple <200 mVPP Quiescent power draw 55 mW @100 VAC 111 mW @265 VAC Switching frequency 25 kHz- 60 kHz Mechanical Dimensions Length: 10.0 cm (3.94 in.) Width: 3.7 cm (1.46 in.) Height: 2.6 cm (1.02 in.) Environmental Ambient operating temperature -25C to 50C Output ratings Application Note 14 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.3 Demo board features Fold back point protection - For a QR flyback converter, the maximum possible output power is increased when a constant current limit value is used across the entire mains input voltage range. This is usually not desired as this will increase the cost of the transformer and output diode in the case of output over power conditions. The internal fold back protection is implemented to adjust the VCS voltage limit according to the bus voltage. Here, the input line voltage is sensed using the current flowing out of the ZC pin, during the MOSFET on-time. As the result, the maximum current limit adjusts with the AC line voltage. VCC over voltage and under voltage protection - During normal operation, the Vcc voltage is continuously monitored. When the Vcc voltage increases to VVCC OVP or Vcc voltage falls below the under voltage lock out level VVCC off, the IC will enter into auto restart mode. Over load/open loop protection - In the case of an open control loop, the feedback voltage is pulled up with an internal block. After a fixed blanking time, the IC enters into auto restart mode. In case of a secondary short-circuit or overload, the regulation voltage VFB will also be pulled up, the same protection is applied and the IC will auto restart. Adjustable output overvoltage protection - During the off-time of the power switch, the voltage at the zerocrossing pin, ZC, is monitored for output overvoltage detection. If the voltage is higher than the preset threshold 3.7 V for a preset period of 100 s, the IC is latched off. Auto restart for over temperature protection - The IC has a built-in over temperature protection function. When the controller's temperature reaches 140 C, the IC will shut down the switch and enters into auto restart. This can protect the power MOSFET from overheating. Short winding protection - The source current of the MOSFET is sensed via external resistors, R15 and R16. If the voltage at the current sensing pin is higher than the preset threshold VCSSW of 1.68 V during the on-time of the power switch, the IC is latched off. This constitutes a short winding protection. To avoid an accidental latch off, a spike blanking time of 190 ns is integrated in the output of internal comparator. Application Note 15 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.4 Figure 16 Schematic 40 W adapter schematic Application Note 16 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.5 BOM with Infineon components in bold Table 9 Reference Description Part number Manufacturer C1 Electrolytic capacitor, 82 uF, 20%, 400 V EKXG401ELL820MM25S United Chemi-Con C2 Electrolytic capacitor, 470 uF, 20%, 25 V EKZE250ELL471MJ16S United Chemi-Con C3 Electrolytic capacitor, 100 uF, 20%, 25 V EEU-FR1E101 Panasonic C4 Capacitor ceramic, 22 nF, X7R, 50 V, CAP0805W VJ0805Y223KNAAO Vishay C5, C20 Capacitor ceramic, 100 nF, X7R, 50 V, CAP0805W C2012X7R2A104K125AA TDK C6 Electrolytic capacitor, 47uF, 20%, 25V, 5 mm UPM1E470MED Nichicon C7 Foil capacitor, 330 nF X2, 20%, 310 VAC, C_Foil 15 mm - V2 R463I33305002K Kemet C10 Capacitor ceramic, 1nF, NP0, 50 V, CAP0805W CGA4C2C0G1H102J060AA TDK C11 Capacitor Y2, 2.2 nF, Y2, 300 V, CAPDISC 7.5 mm AY2222M35Y5US63L7 Vishay C13 Capacitor ceramic, 4.7 nF, NPO, 630 V, CAP1206W C1206C472JBGACTU Kemet C15 Capacitor ceramic, 220 nF, X7R, 25 V, CAP0805W C2012X7R1H224K125AA TDK C16 Capacitor ceramic, 100pF, NP0, 100 V, CAP0805W CGA4C2C0G2A101J060AA TDK C17, C21, C22 Capacitor ceramic, 2.2 uF, X7R, 25 V, CAP1206W C3216X7R1E225K160AA TDK C18, C19 220pF/250 VAC, 220pF, 250 Vac, C075045X100 VY2221K29Y5SS63V0 Vishay C24 Capacitor ceramic, 100 pF, NPO, 630 V, CAP1206W CGA5C4C0G2J101J060AA TDK CON1 ST-04A, IEC C6 AC Connector, ST-A04 6160.0003 Schurter D1 Diode, US1K-E3/61T, 600V, SMA US1K-E3/61T Vishay D2 Diode, NTST30100SG, 100V, TO220_standing NTST30100SG OnSemi D3 2KBP06M, 2KBP06M, 600V, KBPM 2KBP06M-E4/51 Vishay D4 Diode, BAS21-03W, 200V, SOD323 BAS21-03W Infineon D5 Diode, 22V Zener, SOD323 BZX384-C22 NXP F1 T2, 2 A, 250 Vac, Fuse small 40012000440 Littelfuse H1 Heatsink, TO-220 Heatsink 577202B00000G Aavid thermalloy H2 Hardware, Screw, M3, 8 mm M38 PRSTMCZ100- DURATOOL Application Note 17 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview Reference Description Part number Manufacturer H3 Hardware, Nut, A2, M3 M3- HFA2-S100- DURATOOL H4 Hardware, insulator, Insert, 0.15 mm, 19 x 13 mm SPK10-0.006-00-54 Bergquist H5 Hardware, insulator, washer, TO220 insulating washer 7721-7PPSG AAVID THERMALLOY H6 Cable assembly 172-4202 Memory Protection Devices, Inc. IC1 QR PWM controller ICE2QS03G Infineon IC12 VOL617A-2, VOL617A-2, LSOP 4pin VOL617A-2X001T Vishay L1 Choke, 1.0 uH, 20%, INDUCTOR 4 u7 4,2 A 7447462010 Wurth L2 Inductance, 10 mH, Inductor common mode small 744821110 Wurth Q1 NMOS, IPA70R600P7S, 700 V, TO220FP IPA70R600P7S Infineon Q2, Q3 NMOS, 2N7002, 60 V, SOT23 2N7002 Infineon R1 Resistor, 0R, 1%, RES0805R CRCW08050000Z0EA Vishay R2 Resistor, 39k2, 1%, RES0805R ERJ6ENF3922V Panasonic R3 Resistor, 4k99, 1%, RES0805R CRCW08054K99FKEA Vishay R4 Resistor, 33k2, 1%, RES0805R CRCW080533K2FKEA Vishay R5 Resistor, 100k, 1%, RES0805R CRCW0805100KFKEA Vishay R6, R8, R11 Resistor, 10k, 1%, RES0805R CRCW080510K0FKEA Vishay R7, R15 Resistor, 1R, 1%, RES1206W CRCW12061R00FKEA Vishay R10 Resistor, 2k, 1%, RES0805R CRCW08052K00FKEA Vishay R12, R13 Resistor, 4.99M, 1%, RES1206W CRCW12064M99FKEB Vishay R14 Resistor, 330k, 1%, RES0805R CRCW0805330KFKEA Vishay R16 Resistor, 1R5, 1%, RES1206W CRCW12061R50JNEAIF Vishay R17 Resistor, 681k, 1%, RES0805R CRCW0805681KFKEA Vishay R18 Resistor, 51k1, 1%, RES0805R ERJ6ENF5112V Panasonic R19, R24 Resistor, 200k, 1%, RES0805R CRCW0805200KFKEA Vishay R22, R23, R28 Resistor, 33k, 1%, RES1206W CRCW120633K0FKEA Vishay R25 Resistor, 10R, 1%, RES1206W CRCW120610R0FKEA Vishay R27 Resistor, 27R, 1%, RES1206W CRCW120627R0FKEA Vishay T1 Transformer, RM10 ICE160487(spec_700V_v1) I.C.E. Transformers U2, U3 Reference IC, TL431 TL431ACDBZT TI VR1 Varistor, 8.6J, 275Vac B72205S0271K101 EPCOS Application Note 18 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.6 PCB layout The PCB was designed using Altium Designer 16. Schematic and board files are available on request. Figure 17 Board layout top Figure 18 Board layout bottom Application Note 19 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.7 Transformer construction The transformer for the 40 W adapter was built by I.C.E. Transformers: http://www.icetransformers.com/ Table 10 Transformer specification Manufacturer I.C.E. Transformers Core size RM10 Core material 3C95 Bobbin 8 pin RM10 vertical Primary inductance 1500 H measured from pin 6 to pin 4 @10kHz Leakage inductance < 25 H measured from pin 6 to pin 4 @10kHz pins S-,S+,1, and 2 shorted *100% of components are Hi-Pot tested to 4.2 kV primary to secondary for 1 minute Figure 19 1. 2. 3. 4. 5. 6. 7. Transformer windings stackup S- in red tube, S+ in black tube S- length 30 mm, solder length 5 mm S+ length 30 mm, solder length 5 mm Cut pin 3, pin 5, core clip PCB mount pins, and secondary pins. Add a flux band of 8mm copper foil with 2 layers of tape and 3mm of cuffing on each side. Add around the core with the tape side facing out. Using 0.35 mm solder to pin 2. Vacuum varnish the entire assembly. Cut the core clamp pins off of the transformer. Table 11 Transformer windings stackup Name Start Stop Turns Wire gauge Layer Winding P1 6 5 58 1 x 0.35 mm Primary Evenly spaced S1 S- S+ 17 2 x 0.5 mm triple insulated Secondary Evenly spaced P3 2 1 14 1 x 0.15 mm, with margin tape Auxiliary Evenly spaced P2 5 4 29 1 x 0.35 mm Primary Evenly spaced 2 tape T1 Application Note 20 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Measurements 7 Measurements 7.1 High line and low line operation Start of burst mode pulses Figure 20 Last burst mode pulse High line (265 VAC), no load. The ICE2QS03G is operating in burst mode to minimize idle power consumption. The burst mode pulse train shown above occurs every 33.8 ms. QR valley switching does not occur during burst mode due to the ICE2QS03G changing operating modes at light load. VDS maximum is only 553 V in burst mode operation. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Figure 21 Low line (100 VAC), Full load (40 W). This is the peak current that the primary MOSFET Q1 will encounter during steady-state operation. The measured peak current is 1.3 A (780 mV / 0.6 ) giving margin from the power supply maximum current limit of 1.6 A. This is necessary for brown out conditions (90 VAC) and design tolerance. CH1 (Yellow): CH2 (Cyan): CH3 (Magenta): Application Note Q1 VDS Q1 IDS Q1 VGS 21 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Measurements Figure 22 High line (265 VAC), Full load (40 W). This shows the worst case drain source voltage of 599 V. This still gives 14.4 percent margin from the MOSFET breakdown voltage under worst case conditions. The measured peak voltage is higher than calculated due to the snubber equation not considering the RCD snubber capacitance. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Application Note 22 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller Conclusion 8 Conclusion The P7 series of CoolMOSTM MOSFETs offer the best solution for flyback applications. The improvement in switching loss performance over the Infineon CoolMOSTM P6 and competitor devices in this particular design leads to 120 VAC and light load (5 W) efficiency improvements of 2.0 percent and 0.3 percent at 230 VAC. There is also an improvement at 35 W of 0.1 percent at both high line and low line. The improved efficiency of the P7 700 V MOSFET leads to a 2.9 C thermal improvement at 100 VAC and full load operation. The improved efficiency of the P7 700 V MOSFETs is then used to increase the 35 W adapter design to 40 W with the same RDS(on) value. The 700 V breakdown voltage allows for additional safety margin when compared to 650 V for improved surge robustness. The drain source voltage margin is increased by 8 percent while still increasing the drain source voltage compared to a 600 V MOSFET. These changes can be implemented in other charger and adapter designs in order to take advantage of the benefits of the P7 700 V MOSFET. This new benchmark in 700 V MOSFETs enables higher efficiency, higher power density, and more robust designs. Application Note 23 Revision 1.0 2017-01-20 40 W adapter demo board Using the new 700 V CoolMOSTM P7 and ICE2QS03G quasi-resonant PWM controller References 9 References [1] Design Guide for QR Flyback Converter [2] IPA70R600P7S data sheet, 700 V CoolMOSTM P7 Power Transistor [3] 700 V CoolMOS P7TM Application Note [4] ICE2QS03G data sheet, Infineon Technologies AG [5] 2N7002 data sheet, Infineon Technologies AG [6] BAS21-03W data sheet, Infineon Technologies AG [7] ICE2QS03G design guide. [ANPS0027] [8] Converter Design Using the Quasi-Resonant PWM Controller ICE2QS03, Infineon Technologies AG, 2006. [ANPS0003] Revision history Major changes since the last revision Page or reference Application Note Description of change 24 Revision 1.0 2017-01-20 Trademarks of Infineon Technologies AG AURIXTM, C166TM, CanPAKTM, CIPOSTM, CoolGaNTM, CoolMOSTM, CoolSETTM, CoolSiCTM, CORECONTROLTM, CROSSAVETM, DAVETM, DI-POLTM, DrBladeTM, EasyPIMTM, EconoBRIDGETM, EconoDUALTM, EconoPACKTM, EconoPIMTM, EiceDRIVERTM, eupecTM, FCOSTM, HITFETTM, HybridPACKTM, InfineonTM, ISOFACETM, IsoPACKTM, i-WaferTM, MIPAQTM, ModSTACKTM, my-dTM, NovalithICTM, OmniTuneTM, OPTIGATM, OptiMOSTM, ORIGATM, POWERCODETM, PRIMARIONTM, PrimePACKTM, PrimeSTACKTM, PROFETTM, PRO-SILTM, RASICTM, REAL3TM, ReverSaveTM, SatRICTM, SIEGETTM, SIPMOSTM, SmartLEWISTM, SOLID FLASHTM, SPOCTM, TEMPFETTM, thinQ!TM, TRENCHSTOPTM, TriCoreTM. 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