Application Note Please read the Important Notice and Warnings at the end of this document Revision 1.0
www.infineon.com/p7 2017-01-20
AN_201701_PL52_007
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Author: Jared Huntington
Scope and purpose
The demo board described in this application note provides a test platform for the new 700 V CoolMOS™ P7
series of high voltage MOSFETs. The adapter uses the ICE2QS03G, a second generation current mode control
quasi-resonant flyback controller and an IPA70R600P7S 700 V CoolMOS™ P7 series power MOSFET. This
application note is intended for those that have experience with flyback converter designs and will not go in
depth regarding the overall design process, but will cover specific design aspects for this controller and 700 V
CoolMOS™ P7 in charger and adapter applications. It will also look at the overall benefits that the 700 V
CoolMOS™ P7 presents for switch mode power supplies. For a detailed introduction on flyback converter
design please read Design guide for QR Flyback converter [1].
Intended audience
Power supply design engineers
Table of contents
1 Description ................................................................................................................... 2
2 Quasi-resonant flyback overview .................................................................................... 3
3 ICE2QS03G functional overview ...................................................................................... 4
4 P7 700 V CoolMOS™ benefits for adapters ......................................................................... 5
5 Design considerations .................................................................................................... 7
5.1 700 V MOSFET and design changes ........................................................................................................ 7
5.2 UVLO circuit ........................................................................................................................................... 12
6 Demo board overview ................................................................................................... 14
6.1 Demo board pictures ............................................................................................................................ 14
6.2 Demo board specifications ................................................................................................................... 14
6.3 Demo board features ............................................................................................................................ 15
6.4 Schematic .............................................................................................................................................. 16
6.5 BOM with Infineon components in bold ............................................................................................... 17
6.6 PCB layout ............................................................................................................................................. 19
6.7 Transformer construction ..................................................................................................................... 20
7 Measurements .............................................................................................................. 21
7.1 High line and low line operation ........................................................................................................... 21
7.2 Thermal performance under typical operating conditions ................................................................. 11
8 Conclusion ................................................................................................................... 23
9 References ................................................................................................................... 24
Application Note 2 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Description
1 Description
This 40 W adapter demo board is intended to be a form, fit, and function test platform for charger and adapter
applications to show the operation of the 700 V CoolMOS™ P7 as well as the overall controller design. The
demo board is designed around a quasi-resonant flyback topology for improved switching losses that allows
higher power density designs and lower radiated and conducted emissions. A 40 W universal input isolated
flyback demo board with a 19 V output based on the ICE2QS03G controller and the CoolMOS™ P7 MOSFET is
described in this application note and some test results are presented.
Figure 1 40 W flyback demo board
Application Note 3 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Quasi-resonant flyback overview
2 Quasi-resonant flyback overview
The quasi-resonant (QR) flyback offers improved efficiency and electro-magnetic interference (EMI) performance
over the traditional fixed frequency flyback converter by reducing switching losses. This is accomplished by
controlling the turn-on time of the primary MOSFET (Qpri in Figure 3). In a flyback operating in discontinuous
conduction mode (DCM), the energy is first stored in the primary side when the primary MOSFET (Qpri) is turned
on allowing the primary current to ramp up. The primary MOSFET (Qpri) turns-off and the energy stored in the
transformer transfers into the secondary side capacitor. The energy that is left in the primary inductance (Lpri)
after transferring the energy to the secondary then resonates with the combined output capacitance of the
MOSFET(CDS_parasitic) consisting of the MOSFET output capacitance (COSS), stray drain source capacitance from the
transformer and layout, and any additional added external drain source capacitance on this node. In a fixed
frequency flyback the switch turn-on happens regardless of the MOSFET drain source voltage (VDS). If switching
occurs at a higher VDS (Figure 2), this leads to more switching losses (EOSS losses). The QR flyback waits to turn on
Qpri until the VDS voltage reaches the minimum possible voltage shown in Figure 2 and then turns-on the MOSFET.
 
Since the turn-on switching losses are a function of V2 (as shown above), this reduces the overall system
switching losses. This has the added benefit of lowering the amount of switched energy which helps reduce
switching noise from the converter, resulting in lower radiated and conducted emissions.
The 700 V CoolMOS™ P7 technology generates improvements in the operation of QR flyback converters through
having lower output capacitance (COSS) that helps to reduce the losses of the device during turn-on. The
improvements that 700 V CoolMOS™ P7 offers will be further addressed in Section 4.
Figure 2 Fixed frequency flyback primary MOSFET drain source waveform (left) vs. a QR flyback
primary MOSFET drain source waveform (right).
Figure 3 Simplified flyback schematic
MOSFET Turn ON
MOSFET Turn ON
Application Note 4 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
ICE2QS03G functional overview
3 ICE2QS03G functional overview
The ICE2QS03G PWM controller is a second-generation quasi-resonant flyback controller IC developed by
Infineon Technologies. Typical applications include TV-sets, DVD-players, set-top boxes, netbook adapters,
home audio, and printer applications. This controller implements switching at the lowest ringing voltage and
also includes pulse skipping at light loads for maximum efficiency across a wide range of loads.
Figure 4 ICE2QS03G pinout
Table 1 ICE2QS03G pin description
Name
Description
Zero Crossing (ZC)
Detects the minimum trough (valley) voltage for turn-on for the primary
switch turn-on time
Feedback (FB)
Voltage feedback for output regulation
Current Sense (CS)
Primary side current sense for short circuit protection and current
mode control
Gate drive output (GATE)
MOSFET gate driver pin
High Voltage (HV)
Connects to the bus voltage for the initial startup through the high
voltage startup cell
No Connect (NC)
No connection
Power supply (VCC)
Positive IC for the power supply
Ground (GND)
Controller ground
Application Note 5 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
700 V CoolMOS™ P7 benefits for adapters
4 700 V CoolMOS™ P7 benefits for adapters
Figure 5 The MOSFET drain source voltage in a flyback converter is the sum of the bus voltage (Vbus),
reflected voltage (Vreflected), and snubber voltage (Vsnubber).
The 700 V CoolMOS™ P7 provides several benefits for charger and adapter applications when compared to 600 V
and 650 V MOSFETs. The additional breakdown voltage can be used to increase the efficiency of designs,
increase the allowable AC input voltage, or increase the surge capabilities of designs. The P7 family of devices
also has better performance when comparing switching losses to previous generations of MOSFETs.
A 700 V breakdown voltage allows for a higher combination of bus voltage, reflected voltage, and snubber
voltage than can be achieved with 600 V or 650 V devices. This allows for increasing the snubber and reflected
voltage in order to lower the switching losses of a converter. It is also possible to use this extra voltage margin to
allow for bus voltages extending beyond the typical 265 VAC high line. The device can also be used as a drop in
replacement for 600 V and 650 V devices to give additional margin for abnormal conditions such as surge and
output short circuit conditions in existing designs that need improved margins. This additional 50 – 100 V of
drain source breakdown voltage gives designers more flexibility to improve the overall design.
The P7 family of devices also has an improved switching performance that is better than existing Infineon and
competitor devices. One switching loss mechanism is the EOSS of the MOSFET. The EOSS is the main loss
contributor for the turn-on of the MOSFET in a QR flyback. The energy that is stored in the output capacitance of
the MOSFET needs to be discharged every cycle before the MOSFET is turned on. As shown in Figure 6, the
output capacitance energy storage of the 700 V CoolMOS™ P7 is better when compared to equivalent competitor
devices. This improvement is most significant at higher AC input voltages.
Additional details about the 700 V CoolMOS™ P7 device improvements such as the reduced gate charge (Qg),
RDS(on) temperature dependency, QOSS, and transfer characteristics can be found in the CoolMOS™ 700V P7
Application note [3].
Application Note 6 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
700 V CoolMOS™ P7 benefits for adapters
Figure 6 Eoss comparison of a P6 700 V 600 mΩ MOSFET, a P7 700 V 600 mΩ MOSFET, and a competitor’s
700 V 600 mΩ device. It can be seen that the amount of energy stored in the output
capacitance during a typical QR high line turn on of 200 V is reduced by 0.5 µJ every switching
cycle, which in a 100 kHz design corresponds to 50 mW.
SPICE models of the P7 700 V MOSFETs are provided on the Infineon website. These models have been created
with MOSFET characterization data covering different MOSFET parameters and provide a high level of accuracy.
Below, Figure 7 shows the difference between the Infineon 40 W adapter’s measured waveforms and the
simulated waveforms. These models can be used to better understand the loss mechanisms that are responsible
for power dissipation in the primary MOSFET of the Flyback converter and help to optimize designs.
Figure 7 Simulated switching vs. measured switching at 230 VAC operation in the Infineon 40 W adapter.
0,0
0,5
1,0
1,5
2,0
2,5
3,0
0 100 200 300 400
EOSS (µJ)
V
DS
(V)
IPA60R600P6
IPA70R600P7
Competitor
Application Note 7 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
5 Design considerations
5.1 700 V MOSFET and design changes
This section will compare the Infineon 35 W adapter design using an Infineon 600 V C6 MOSFET with the Infineon
700 V CoolMOS™ P7 based 40 W adapter to show the differences between the two designs.
The flyback MOSFET was changed from the CoolMOS™ C6 family of devices to the P7 700 V IPA70R600P7S in
order to have the best performance from the latest generation of Infineon devices. With the same RDS(on), the
switching characteristics of the MOSFET are improved. In this particular design the peak drain source voltage
was increased from 526 V to 560 V by slightly reducing the reflected voltage and increasing the snubber clamp
voltage leading to a reduction in snubber energy dissipation. Reducing the switching losses helps to improve the
high line and light load efficiency of the 40 W adapter. Even with increasing the drain source voltage, the
calculated breakdown voltage margin increased from 12 percent to 20 percent. The output diode was also
replaced with a lower cost, lower voltage drop diode in order to reduce the full load power losses.
The reflected voltage was decreased from the Infineon 35 W adaptor in order to reduce the snubber losses. The
RCD snubber resistor value was also increased to further reduce the snubber loss, which causes the maximum
drain source peak voltage to increase. As shown in Table 2, the turns ratio is reduced by 20 V in order to reduce
the reflected voltage, which helps to reduce the energy that is dissipated across the snubber network as shown
in the equation below:
 

Table 2
Parameter
Symbol
600 V design
700 V P7 design
Transformer primary turns
NP
66 turns
87 turns
Transformer secondary turns
NS
11 turns
17 turns
Output voltage
Voutput
19 V
19 V
Transformer reflected voltage
Vreflected
117 V
97 V
The primary side resistor, capacitor, and diode (RCD) snubber network resistor power dissipation was reduced
allowing the snubber voltage to reach a higher level and thus lowering the amount of energy that is dissipated in
the snubber resistor. This comes into effect especially at lower power levels where the conduction loss is no
longer the dominant source of power losses. The snubber clamping voltage can be calculated using the
equation below:

󰇧


󰇨
Table 3 shows that by increasing the snubber resistor from 54 kΩ to 99 kΩ the snubber voltage increases from
40.1 V to 97 V.
Table 3
Parameter
Symbol
600 V design
700 V P7 design
Leakage inductance
Lleakage
25 µH
25 µH
Peak primary current under load at high line
Ipri
0.43 A
0.62 A
Snubber resistor
Rsnubber
54 kΩ
99 kΩ
Switching period
Ts
28.6 µs
28.6 µs
Application Note 8 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
Parameter
Symbol
600 V design
700 V P7 design
Snubber voltage
Vsnubber
40.1 V
97 V
Table 4 sums up the different voltage components of the VDS waveform to arrive at the total calculated peak
MOSFET VDS. As shown in Figure 5, the total drain source voltage of the MOSFET is the sum of the bus voltage, the
reflected voltage, and the snubber voltage. Even with increasing the overall drain source voltage (VDS) by 34 V we
still have an increase in margin from the MOSFET breakdown voltage. In this new design the margin has
increased from 12 percent to 20 percent. This increases the overall margin from the MOSFET breakdown voltage
while still increasing the peak drain source voltage. In reality, when measured under worst-case conditions this
20 perecent margin corresponds to 599 V or 15 percent margin. The difference between measured and
calculated peak voltages comes from the fact that the snubber voltage equation above assumes an infinite
snubber capacitor value. Because of this discrepancy, the drain source peak voltage at full load and high line
should be empirically verified in the design.
Table 4
Parameter
Symbol
600 V design
700 V P7 design
Primary bus voltage @265 VAC
Vbus
373 V
373 V
Reflected voltage
Vreflected
117 V
97 V
Snubber voltage
Vsnubber
40.1 V
90 V
Drain source voltage maximum
VDS_max
526 V
560 V
Margin from breakdown voltage
VDS_margin
12 %
20 %
In order to increase the power level from 35 W to 40 W, the output diode of the power supply needed less power
dissipation at full load. A diode with a lower forward voltage drop was selected that improves the output power
dissipation by 150 mW, even when operated at a higher output power levels and thus a higher output current.
This reduces the temperature sufficiently to allow increasing the total output power.
Table 5
Parameter
Symbol
600 V design
700 V P7 design
Output current
Ioutput
1.84 A
2.11 A
Diode forward voltage
Vforward
0.55 V
0.40 V
Diode conduction losses
Pdio
1.01 W
0.84 A
Application Note 9 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
With all of these changes made to the design, the overall efficiency improvement can be seen in Figure 8 and
Figure 9 below. The light load benefits come from the P7 700 V switching loss improvements and RCD snubber
changes. The P7 switching loss improvements can be seen by looking at the IPA60R600P6 to IPA70R600P7S delta
efficiency curves shown in Figure 10 and Figure 11. The full load efficiency improvements come from changing to
a better output diode with a lower forward voltage drop. By making these changes, the power level of the design
was increased without switching to a lower RDS(on) device which would have cause an increase in the overall BOM
cost.
Figure 8 40 W adapter efficiency at 120 VAC using IPA70R600P7S efficiency vs. a 35 W adapter design
using a P6 device.
80,0
81,0
82,0
83,0
84,0
85,0
86,0
87,0
88,0
89,0
90,0
91,0
92,0
5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0
Efficiency (%)
Output power (W)
40W adapter, IPA70R600P7S
40W adapter, IPA60R600P6
40W adapter, competitor
35W adapter, IPD60R600P6
80,0
81,0
82,0
83,0
84,0
85,0
86,0
87,0
88,0
89,0
90,0
91,0
92,0
5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0
Efficiency (%)
Output power (W)
40W adapter, IPA70R600P7S
40W adapter, IPA60R600P6
40W adapter, competitor
35W adapter, IPA60R600P6
Application Note 10 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
Figure 9 40 W adapter efficiency at 230 VAC using IPA70R600P7S efficiency vs. a 35 W adapter design
using a P6 device.
Figure 10 and Figure 11 below show the benefits of changing from IPA60R600P7 or a competitor’s device to a P7
device. The efficiency graphs below are done as efficiency deltas relative to the P7 IPA70R600P7S in order to
make the efficiency differences clearer. The 100 VAC and full load efficiency difference ends up reducing the mold
compound temperature by 2.9 °C.
Figure 10 Efficiency of the 40 W adapter at 120 VAC showing the P6 and competitor’s devices referenced
to the P7 MOSFET.
Figure 11 Efficiency of the 40 W adapter at 230 VAC showing the P6 and competitor’s devices referenced
to the P7 MOSFET.
-2,00
-1,50
-1,00
-0,50
0,00 5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0
∆ Efficiency (%)
Output power (W)
40W adapter, IPA70R600P7S
40W adapter, IPA60R600P6
40W adapter, competitor
-1,00
-0,90
-0,80
-0,70
-0,60
-0,50
-0,40
-0,30
-0,20
-0,10
0,00
0,10
5,0 10,0 15,0 20,0 25,0 30,0 35,0 40,0
∆ Efficiency (%)
Output power (W)
40W adapter, IPA70R600P7S
40W adapter, IPA60R600P6
40W adapter, competitor
Application Note 11 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
5.2 Thermal performance improvement
The worst-case nominal thermal conditions for the system under steady state operation occur at 100 VAC and full
output power (40 W). Table 10 shows the thermal improvement of P7 700 V when used in the 40 W adapter at 100
VAC and 40 W of output power due to the improvement in efficiency shown in the previous section. As shown
below, the mold compound temperature of the P7 700 V device is 2.9 °C lower. Table 11 shows the maximum
temperature under the worst-case operating conditions for these components when using the IPA70R600P7S.
These component temperatures are the limiting factor for the overall power density of the converter and this
can help to increase power density or improve thermal margins in designs.
Table 6 Flyback MOSFET thermal rise with 100 VAC, 40 W output
Device
Temperature rise
Temperature Δ referenced to P7
Competition
51.3 °C
+2.0 °C
IPA60R600P6
52.2 °C
+2.9 °C
IPA70R600P7S
49.3 °C
0.0°C
Table 7 Maximum component thermal rise at 100 VAC, 40W output
Ref. Des.
Component description
Maximum temperature rise
Q1
Flyback MOSFET
61.7 °C
D3
Bridge rectifier
54.9 °C
L1
Common Mode Choke
59.3 °C
T1
Flyback transformer
67.8 °C
D2
Flyback output diode
53.6 °C
R22, R23, R28
Flyback snubber resistors
79.2 °C
Figure 12 100 VAC input, full load, top side. The line filter and bridge rectifier are hottest at this point due
to higher AC input currents.
Q1
L1
D3
T1
D2
Application Note 12 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
Figure 13 100 VAC input, full load, bottom side. The snubber resistors are the hottest components.
5.3 UVLO circuit
The under voltage lock out (UVLO) circuit provides a mechanism to shut down the power supply when the AC
line input voltage is lower than the specified voltage range. The UVLO event is detected by sensing the voltage
level at U2’s (TL431) REF pin (VREF_typ = 2.5 V) through the voltage divider resistors (R12, R13, R14, and R17 in
Figure 12) from the bulk capacitor C1. Q2 acts as a switch to enter or leave UVLO mode by controlling the FB pin
voltage. Q3, together with R17, acts as voltage hysteresis for the UVLO circuit and U2 (TL431) act as a
comparator. The system enters the UVLO mode by controlling the FB pin voltage of U1 to 0 V (when the voltage
input level goes back to input voltage range), VREF increases to 2.5 V (then switches Q2 and Q3 off) and Vcc hits 18
V, the UVLO mode is released. The calculation for the UVLO circuit is shown below:
VREF= 2.5 V
R12 = 4.99 MΩ R13 = 4.99 MΩ R14 = 330 kΩ R17 = 681 kΩ
 󰇛󰇜

 󰇣󰇡
󰇢󰇤
󰇡
󰇢
 
 
The 'enter UVLO' threshold is set at 77.8 to allow for the BUS capacitance voltage to droop under 90 VAC at
full load operation with some margin to avoid false triggering.
R22, R23, R28
Application Note 13 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Design considerations
Figure 14 Power supply status vs. AC input voltage showing the hysteretic behavior of the UVLO circuit.
Application Note 14 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6 Demo board overview
6.1 Demo board pictures
Figure 15 Top side of 40 W Infineon adapter with a TO-220 FullPAK populated
6.2 Demo board specifications
Table 8
Section
Parameter
Specification
Input ratings
Input voltage
90 VAC – 265 VAC
Input frequency
47 Hz – 63 Hz
Input current at 100 VAC, 40 W
0.85 A maximum
Power factor
0.53 @100 VAC
0.36 @265 VAC
Peak efficiency 230 VAC, 40 W
Peak efficiency 120 VAC, 40 W
91.3%
89.6%
Surge
2 kV IEC61000-4-5
Output ratings
Nominal output voltage
19.0 V
Tolerance
2%
Output current
2.10 A
Output power
40 W
Line regulation
0.5%
Load regulation
0.5%
Output ripple
<200 mVPP
Quiescent power draw
55 mW @100 VAC
111 mW @265 VAC
Switching frequency
25 kHz– 60 kHz
Mechanical
Dimensions
Length: 10.0 cm (3.94 in.)
Width: 3.7 cm (1.46 in.)
Height: 2.6 cm (1.02 in.)
Environmental
Ambient operating temperature
-25°C to 50°C
Q1 IPA70R600P7S
Application Note 15 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6.3 Demo board features
Fold back point protection - For a QR flyback converter, the maximum possible output power is increased
when a constant current limit value is used across the entire mains input voltage range. This is usually not
desired as this will increase the cost of the transformer and output diode in the case of output over power
conditions. The internal fold back protection is implemented to adjust the VCS voltage limit according to the
bus voltage. Here, the input line voltage is sensed using the current flowing out of the ZC pin, during the
MOSFET on-time. As the result, the maximum current limit adjusts with the AC line voltage.
VCC over voltage and under voltage protection - During normal operation, the Vcc voltage is continuously
monitored. When the Vcc voltage increases to VVCC OVP or Vcc voltage falls below the under voltage lock out
level VVCC off, the IC will enter into auto restart mode.
Over load/open loop protection - In the case of an open control loop, the feedback voltage is pulled up with
an internal block. After a fixed blanking time, the IC enters into auto restart mode. In case of a secondary
short-circuit or overload, the regulation voltage VFB will also be pulled up, the same protection is applied and
the IC will auto restart.
Adjustable output overvoltage protection - During the off-time of the power switch, the voltage at the zero-
crossing pin, ZC, is monitored for output overvoltage detection. If the voltage is higher than the preset
threshold 3.7 V for a preset period of 100 μs, the IC is latched off.
Auto restart for over temperature protection - The IC has a built-in over temperature protection function.
When the controller’s temperature reaches 140 °C, the IC will shut down the switch and enters into auto
restart. This can protect the power MOSFET from overheating.
Short winding protection - The source current of the MOSFET is sensed via external resistors, R15 and R16. If
the voltage at the current sensing pin is higher than the preset threshold VCSSW of 1.68 V during the on-time
of the power switch, the IC is latched off. This constitutes a short winding protection. To avoid an accidental
latch off, a spike blanking time of 190 ns is integrated in the output of internal comparator.
Application Note 16 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6.4 Schematic
Figure 16 40 W adapter schematic
Application Note 17 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6.5 BOM with Infineon components in bold
Table 9
Reference
Description
Part number
Manufacturer
C1
Electrolytic capacitor, 82 uF, 20%,
400 V
EKXG401ELL820MM25S
United Chemi-Con
C2
Electrolytic capacitor, 470 uF, 20%,
25 V
EKZE250ELL471MJ16S
United Chemi-Con
C3
Electrolytic capacitor, 100 uF, 20%,
25 V
EEU-FR1E101
Panasonic
C4
Capacitor ceramic, 22 nF, X7R, 50 V,
CAP0805W
VJ0805Y223KNAAO
Vishay
C5, C20
Capacitor ceramic, 100 nF, X7R, 50 V,
CAP0805W
C2012X7R2A104K125AA
TDK
C6
Electrolytic capacitor, 47uF, 20%,
25V, 5 mm
UPM1E470MED
Nichicon
C7
Foil capacitor, 330 nF X2, 20%, 310
VAC, C_Foil 15 mm - V2
R463I33305002K
Kemet
C10
Capacitor ceramic, 1nF, NP0, 50 V,
CAP0805W
CGA4C2C0G1H102J060AA
TDK
C11
Capacitor Y2, 2.2 nF, Y2, 300 V, CAP-
DISC 7.5 mm
AY2222M35Y5US63L7
Vishay
C13
Capacitor ceramic, 4.7 nF, NPO, 630
V, CAP1206W
C1206C472JBGACTU
Kemet
C15
Capacitor ceramic, 220 nF, X7R, 25 V,
CAP0805W
C2012X7R1H224K125AA
TDK
C16
Capacitor ceramic, 100pF, NP0, 100 V,
CAP0805W
CGA4C2C0G2A101J060AA
TDK
C17, C21, C22
Capacitor ceramic, 2.2 uF, X7R, 25 V,
CAP1206W
C3216X7R1E225K160AA
TDK
C18, C19
220pF/250 VAC, 220pF, 250 Vac, C075-
045X100
VY2221K29Y5SS63V0
Vishay
C24
Capacitor ceramic, 100 pF, NPO, 630
V, CAP1206W
CGA5C4C0G2J101J060AA
TDK
CON1
ST-04A, IEC C6 AC Connector, ST-A04
6160.0003
Schurter
D1
Diode, US1K-E3/61T, 600V, SMA
US1K-E3/61T
Vishay
D2
Diode, NTST30100SG, 100V,
TO220_standing
NTST30100SG
OnSemi
D3
2KBP06M, 2KBP06M, 600V, KBPM
2KBP06M-E4/51
Vishay
D4
Diode, BAS21-03W, 200V, SOD323
BAS21-03W
Infineon
D5
Diode, 22V Zener, SOD323
BZX384-C22
NXP
F1
T2, 2 A, 250 Vac, Fuse small
40012000440
Littelfuse
H1
Heatsink, TO-220 Heatsink
577202B00000G
Aavid thermalloy
H2
Hardware, Screw, M3, 8 mm
M38 PRSTMCZ100-
DURATOOL
Application Note 18 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
Reference
Description
Part number
Manufacturer
H3
Hardware, Nut, A2, M3
M3- HFA2-S100-
DURATOOL
H4
Hardware, insulator, Insert, 0.15 mm,
19 x 13 mm
SPK10-0.006-00-54
Bergquist
H5
Hardware, insulator, washer, TO220
insulating washer
7721-7PPSG
AAVID THERMALLOY
H6
Cable assembly
172-4202
Memory Protection
Devices, Inc.
IC1
QR PWM controller
ICE2QS03G
Infineon
IC12
VOL617A-2, VOL617A-2, LSOP 4pin
VOL617A-2X001T
Vishay
L1
Choke, 1.0 uH, 20%, INDUCTOR 4 u7
4,2 A
7447462010
Wurth
L2
Inductance, 10 mH, Inductor
common mode small
744821110
Wurth
Q1
NMOS, IPA70R600P7S, 700 V,
TO220FP
IPA70R600P7S
Infineon
Q2, Q3
NMOS, 2N7002, 60 V, SOT23
2N7002
Infineon
R1
Resistor, 0R, 1%, RES0805R
CRCW08050000Z0EA
Vishay
R2
Resistor, 39k2, 1%, RES0805R
ERJ6ENF3922V
Panasonic
R3
Resistor, 4k99, 1%, RES0805R
CRCW08054K99FKEA
Vishay
R4
Resistor, 33k2, 1%, RES0805R
CRCW080533K2FKEA
Vishay
R5
Resistor, 100k, 1%, RES0805R
CRCW0805100KFKEA
Vishay
R6, R8, R11
Resistor, 10k, 1%, RES0805R
CRCW080510K0FKEA
Vishay
R7, R15
Resistor, 1R, 1%, RES1206W
CRCW12061R00FKEA
Vishay
R10
Resistor, 2k, 1%, RES0805R
CRCW08052K00FKEA
Vishay
R12, R13
Resistor, 4.99M, 1%, RES1206W
CRCW12064M99FKEB
Vishay
R14
Resistor, 330k, 1%, RES0805R
CRCW0805330KFKEA
Vishay
R16
Resistor, 1R5, 1%, RES1206W
CRCW12061R50JNEAIF
Vishay
R17
Resistor, 681k, 1%, RES0805R
CRCW0805681KFKEA
Vishay
R18
Resistor, 51k1, 1%, RES0805R
ERJ6ENF5112V
Panasonic
R19, R24
Resistor, 200k, 1%, RES0805R
CRCW0805200KFKEA
Vishay
R22, R23, R28
Resistor, 33k, 1%, RES1206W
CRCW120633K0FKEA
Vishay
R25
Resistor, 10R, 1%, RES1206W
CRCW120610R0FKEA
Vishay
R27
Resistor, 27R, 1%, RES1206W
CRCW120627R0FKEA
Vishay
T1
Transformer, RM10
ICE160487(spec_700V_v1)
I.C.E. Transformers
U2, U3
Reference IC, TL431
TL431ACDBZT
TI
VR1
Varistor, 8.6J, 275Vac
B72205S0271K101
EPCOS
Application Note 19 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6.6 PCB layout
The PCB was designed using Altium Designer 16. Schematic and board files are available on request.
Figure 17 Board layout top
Figure 18 Board layout bottom
Application Note 20 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Demo board overview
6.7 Transformer construction
The transformer for the 40 W adapter was built by I.C.E. Transformers: http://www.icetransformers.com/
Table 10 Transformer specification
Manufacturer
I.C.E. Transformers
Core size
RM10
Core material
3C95
Bobbin
8 pin RM10 vertical
Primary inductance
1500 µH measured from pin 6 to pin 4 @10kHz
Leakage inductance
< 25 µH measured from pin 6 to pin 4 @10kHz pins S-,S+,1, and 2 shorted
*100% of components are Hi-Pot tested to 4.2 kV primary to secondary for 1 minute
Figure 19 Transformer windings stackup
1. S- in red tube, S+ in black tube
2. S- length
30 mm
, solder length 5 mm
3. S+ length
30 mm
, solder length 5 mm
4. Cut pin 3, pin 5, core clip PCB mount pins, and secondary pins.
5. Add a flux band of 8mm copper foil with 2 layers of tape and 3mm of cuffing on each side. Add around the
core with the tape side facing out. Using ɸ0.35 mm solder to pin 2.
6. Vacuum varnish the entire assembly.
7. Cut the core clamp pins off of the transformer.
Table 11 Transformer windings stackup
Name
Start
Stop
Turns
Wire gauge
Layer
Winding
P1
6
5
58
1 x ɸ0.35 mm
Primary
Evenly spaced
S1
S-
S+
17
2 x ɸ 0.5 mm triple insulated
Secondary
Evenly spaced
P3
2
1
14
1 x ɸ0.15 mm, with margin tape
Auxiliary
Evenly spaced
P2
5
4
29
1 x ɸ0.35 mm
Primary
Evenly spaced
T1
2
tape
Application Note 21 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Measurements
7 Measurements
7.1 High line and low line operation
Figure 20 High line (265 VAC), no load. The ICE2QS03G is operating in burst mode to minimize idle power
consumption. The burst mode pulse train shown above occurs every 33.8 ms. QR valley
switching does not occur during burst mode due to the ICE2QS03G changing operating modes
at light load. VDS maximum is only 553 V in burst mode operation.
CH1 (Yellow): Q1 VDS
CH2 (Cyan): Q1 IDS
CH3 (Magenta): Q1 VGS
Figure 21 Low line (100 VAC), Full load (40 W). This is the peak current that the primary MOSFET Q1 will
encounter during steady-state operation. The measured peak current is 1.3 A (780 mV / 0.6 Ω)
giving margin from the power supply maximum current limit of 1.6 A. This is necessary for
brown out conditions (90 VAC) and design tolerance.
CH1 (Yellow): Q1 VDS
CH2 (Cyan): Q1 IDS
CH3 (Magenta): Q1 VGS
Start of burst mode pulses
Last burst mode pulse
Application Note 22 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Measurements
Figure 22 High line (265 VAC), Full load (40 W). This shows the worst case drain source voltage of 599 V.
This still gives 14.4 percent margin from the MOSFET breakdown voltage under worst case
conditions. The measured peak voltage is higher than calculated due to the snubber equation
not considering the RCD snubber capacitance.
CH1 (Yellow): Q1 VDS
CH2 (Cyan): Q1 IDS
CH3 (Magenta): Q1 VGS
Application Note 23 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
Conclusion
8 Conclusion
The P7 series of CoolMOS™ MOSFETs offer the best solution for flyback applications. The improvement in
switching loss performance over the Infineon CoolMOS™ P6 and competitor devices in this particular design
leads to 120 VAC and light load (5 W) efficiency improvements of 2.0 percent and 0.3 percent at 230 VAC. There is
also an improvement at 35 W of 0.1 percent at both high line and low line. The improved efficiency of the P7 700
V MOSFET leads to a 2.9 °C thermal improvement at 100 VAC and full load operation. The improved efficiency of
the P7 700 V MOSFETs is then used to increase the 35 W adapter design to 40 W with the same RDS(on) value. The
700 V breakdown voltage allows for additional safety margin when compared to 650 V for improved surge
robustness. The drain source voltage margin is increased by 8 percent while still increasing the drain source
voltage compared to a 600 V MOSFET. These changes can be implemented in other charger and adapter designs
in order to take advantage of the benefits of the P7 700 V MOSFET. This new benchmark in 700 V MOSFETs
enables higher efficiency, higher power density, and more robust designs.
Application Note 24 Revision 1.0
2017-01-20
40 W adapter demo board
Using the new 700 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM
controller
References
9 References
[1] Design Guide for QR Flyback Converter
[2] IPA70R600P7S data sheet, 700 V CoolMOS™ P7 Power Transistor
[3] 700 V CoolMOS P7™ Application Note
[4] ICE2QS03G data sheet, Infineon Technologies AG
[5] 2N7002 data sheet, Infineon Technologies AG
[6] BAS21-03W data sheet, Infineon Technologies AG
[7] ICE2QS03G design guide. [ANPS0027]
[8] Converter Design Using the Quasi-Resonant PWM Controller ICE2QS03, Infineon Technologies AG, 2006.
[ANPS0003]
Revision history
Major changes since the last revision
Page or reference
Description of change
Published by
Infineon Technologies AG
81726 München, Germany
© 2017 Infineon Technologies AG.
All Rights Reserved.
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Edition 2017-01-20
ifx1
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