a Dual, Current Feedback Low Power Op Amp AD812 PIN CONFIGURATION 8-Lead Plastic Mini-DIP and SOIC FEATURES Two Video Amplifiers in One 8-Lead SOIC Package Optimized for Driving Cables in Video Systems Excellent Video Specifications (RL = 150 ): Gain Flatness 0.1 dB to 40 MHz 0.02% Differential Gain Error 0.02 Differential Phase Error Low Power Operates on Single +3 V Supply 5.5 mA/Amplifier Max Power Supply Current High Speed 145 MHz Unity Gain Bandwidth (3 dB) 1600 V/s Slew Rate Easy to Use 50 mA Output Current Output Swing to 1 V of Rails (150 Load) PRODUCT DESCRIPTION The AD812 is a low power, single supply, dual video amplifier. Each of the amplifiers have 50 mA of output current and are optimized for driving one back-terminated video load (150 ) each. Each amplifier is a current feedback amplifier and features gain flatness of 0.1 dB to 40 MHz while offering differential gain and phase error of 0.02% and 0.02. This makes the AD812 ideal for professional video electronics such as cameras and video switchers. -IN1 2 8 V+ 7 OUT2 + 6 -IN2 +IN1 3 + V- 4 AD812 5 +IN2 The AD812 offers low power of 4.0 mA per amplifier max (VS = +5 V) and can run on a single +3 V power supply. The outputs of each amplifier swing to within one volt of either supply rail to easily accommodate video signals of 1 V p-p. Also, at gains of +2 the AD812 can swing 3 V p-p on a single +5 V power supply. All this is offered in a small 8-lead plastic DIP or 8-lead SOIC package. These features make this dual amplifier ideal for portable and battery powered applications where size and power is critical. The outstanding bandwidth of 145 MHz along with 1600 V/s of slew rate make the AD812 useful in many general purpose high speed applications where a single +5 V or dual power supplies up to 15 V are available. The AD812 is available in the industrial temperature range of -40C to +85C. 0.4 0.06 G = +2 RL = 150V 0.3 0.04 0.2 0.1 DIFFERENTIAL PHASE - Degrees NORMALIZED GAIN - dB DIFFERENTIAL GAIN 0 -0.1 -0.2 VS = 615V -0.3 65V -0.4 5V -0.5 3V -0.6 100k 1M 10M FREQUENCY - Hz 100M Figure 1. Fine-Scale Gain Flatness vs. Frequency, Gain = +2, RL = 150 0.08 0.02 0.06 DIFFERENTIAL GAIN - % APPLICATIONS Video Line Driver Professional Cameras Video Switchers Special Effects OUT1 1 DIFFERENTIAL PHASE 0.04 0.02 0 5 6 7 8 9 10 11 12 SUPPLY VOLTAGE - 6Volts 13 14 15 Figure 2. Differential Gain and Phase vs. Supply Voltage, Gain = +2, RL = 150 REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 (c) Analog Devices, Inc., 1998 AD812-SPECIFICATIONS Dual Supply (@ T = +25C, R = 150 , unless otherwise noted) A L Model DYNAMIC PERFORMANCE -3 dB Bandwidth Bandwidth for 0.1 dB Flatness Conditions VS Min G = +2, No Peaking 5 V 15 V 15 V 5 V 15 V 5 V 15 V 5 V 15 V 50 75 100 20 25 275 1400 Gain = +1 G = +2 Slew Rate1 G = +2, RL = 1 k 20 V Step G = -1, RL = 1 k Settling Time to 0.1% G = -1, RL = 1 k VO = 3 V Step VO = 10 V Step NOISE/HARMONIC PERFORMANCE Total Harmonic Distortion Input Voltage Noise Input Current Noise Differential Gain Error fC = 1 MHz, RL = 1 k f = 10 kHz f = 10 kHz, +In f = 10 kHz, -In NTSC, G = +2, RL = 150 Differential Phase Error DC PERFORMANCE Input Offset Voltage TMIN -TMAX Offset Drift -Input Bias Current TMIN -T MAX +Input Bias Current Open-Loop Voltage Gain Open-Loop Transresistance INPUT CHARACTERISTICS Input Resistance Input Capacitance Input Common Mode Voltage Range Common-Mode Rejection Ratio Input Offset Voltage -Input Current +Input Current Input Offset Voltage -Input Current +Input Current TMIN -T MAX VO = 2.5 V, RL = 150 TMIN -T MAX VO = 10 V, RL = 1 k TMIN -T MAX VO = 2.5 V, RL = 150 TMIN -T MAX VO = 10 V, RL = 1 k TMIN -T MAX 5 V 15 V 50 40 ns ns 15 V 5 V, 15 V 5 V, 15 V 5 V, 15 V 5 V 15 V 5 V 15 V -90 3.5 1.5 18 0.05 0.02 0.07 0.02 dBc nV/Hz pA/Hz pA/Hz % % Degrees Degrees 5 V, 15 V 2 5 V, 15 V 5 V, 15 V 15 7 5 V, 15 V 0.3 5 V 15 V 5 V 15 V 5 V VCM = 12 V 15 V -2- Units MHz MHz MHz MHz MHz V/s V/s V/s V/s 68 69 76 75 350 270 450 370 0.1 0.06 0.15 0.06 5 12 25 38 1.5 2.0 76 82 550 800 15 65 1.7 4.0 13.5 5 V 15 V VCM = 2.5 V Max 65 100 145 30 40 425 1600 250 600 15 V +Input -Input +Input AD812A Typ 51 55 58 2 0.07 60 1.5 0.05 mV mV V/C A A A A dB dB dB dB k k k k M pF V V 3.0 0.15 3.3 0.15 dB A/V A/V dB A/V A/V REV. B AD812 Model OUTPUT CHARACTERISTICS Output Voltage Swing Conditions VS Min RL = 150 , TMIN -TMAX RL = 1 k, TMIN -TMAX 5 V 15 V 5 V 15 V 15 V 3.5 13.6 30 40 Output Resistance MATCHING CHARACTERISTICS Dynamic Crosstalk Gain Flatness Match DC Input offset Voltage -Input Bias Current POWER SUPPLY Operating Range Quiescent Current Units V V mA mA mA 15 V 15 G = +2, f = 5 MHz G = +2, f = 40 MHz 5 V, 15 V 15 V -75 0.1 dB dB TMIN -TMAX TMIN -TMAX 5 V, 15 V 5 V, 15 V 0.5 2 3.6 25 mV A Per Amplifier 5 V 15 V 15 V 3.5 4.5 18 4.0 5.5 6.0 V mA mA mA 0.6 0.05 dB A/V A/V G = +2, RF = 715 VIN = 2 V Open-Loop TMIN -TMAX Power Supply Rejection Ratio Input Offset Voltage -Input Current +Input Current Max 3.8 14.0 40 50 100 Output Current Short Circuit Current AD812A Typ VS = 1.5 V to 15 V 1.2 70 80 0.3 0.005 NOTES 1 Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain. Specifications subject to change without notice. Single Supply (@ TA = +25C, RL = 150 , unless otherwise noted) Model DYNAMIC PERFORMANCE -3 dB Bandwidth Bandwidth for 0.1 dB Flatness Slew Rate1 NOISE/HARMONIC PERFORMANCE Input Voltage Noise Input Current Noise Differential Gain Error2 Differential Phase Error 2 REV. B AD812A Typ Conditions VS Min G = +2, No Peaking +5 V +3 V 35 30 50 40 MHz MHz G = +2 +5 V +3 V +5 V +3 V 13 10 20 18 125 60 MHz MHz V/s V/s 3.5 1.5 18 0.07 0.15 0.06 0.15 nV/Hz pA/Hz pA/Hz % % Degrees Degrees G = +2, RL = 1 k f = 10 kHz f = 10 kHz, +In f = 10 kHz, -In NTSC, G = +2, RL = 150 G = +1 G = +2 G = +1 -3- +5 V, +3 V +5 V, +3 V +5 V, +3 V +5 V +3 V +5 V +3 V Max Units AD812-SPECIFICATIONS Single Supply (Continued) Model Conditions VS DC PERFORMANCE Input Offset Voltage Min AD812A Typ +5 V, +3 V 1.5 +5 V, +3 V +5 V, +3 V 7 2 +5 V, +3 V 0.2 TMIN -TMAX Offset Drift -Input Bias Current TMIN -TMAX +Input Bias Current Open-Loop Voltage Gain Open-Loop Transresistance INPUT CHARACTERISTICS Input Resistance Input Capacitance Input Common Mode Voltage Range Common-Mode Rejection Ratio Input Offset Voltage -Input Current +Input Current Input Offset Voltage -Input Current +Input Current OUTPUT CHARACTERISTICS Output Voltage Swing p-p TMIN -TMAX VO = +2.5 V p-p VO = +0.7 V p-p VO = +2.5 V p-p VO = +0.7 V p-p +5 V +3 V +5 V +3 V +Input -Input +Input MATCHING CHARACTERISTICS Dynamic Crosstalk Gain Flatness Match DC Input offset Voltage -Input Bias Current POWER SUPPLY Operating Range Quiescent Current VCM = 1.25 V to 3.75 V 1.0 1.0 +5 V 52 +3 V RL = 1 k, TMIN -TMAX RL = 150 , TMIN -TMAX +5 V +5 V +3 V +5 V +3 V +5 V G = +2, RF = 715 VIN = 1 V Units 4.5 7.0 mV mV V/C A A A A dB dB k k 20 30 1.5 2.0 73 70 400 300 15 90 2 +5 V +3 V VCM = 1 V to 2 V 3.0 2.8 1.0 20 15 4.0 2.0 55 3 0.1 52 3.5 0.1 5.5 0.2 M pF V V dB A/V A/V dB A/V A/V 3.2 3.1 1.3 30 25 40 V p-p V p-p V p-p mA mA mA dB dB G = +2, f = 5 MHz G = +2, f = 20 MHz +5 V, +3 V +5 V, +3 V -72 0.1 TMIN -TMAX TMIN -TMAX +5 V, +3 V +5 V, +3 V 0.5 2 3.5 25 mV A Per Amplifier +5 V +3 V +5 V 3.2 3.0 36 4.0 3.5 4.5 V mA mA mA 0.6 0.05 dB A/V A/V 2.4 TMIN -TMAX Power Supply Rejection Ratio Input Offset Voltage -Input Current +Input Current 250 +5 V +5 V Output Current Short Circuit Current 67 Max VS = +3 V to +30 V 70 TRANSISTOR COUNT 80 0.3 0.005 56 NOTES 1 Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain. 2 Single supply differential gain and phase are measured with the ac coupled circuit of Figure 53. Specifications subject to change without notice. -4- REV. B AD812 ABSOLUTE MAXIMUM RATINGS 1 MAXIMUM POWER DISSIPATION Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipation2 Plastic (N) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts Small Outline (R) . . . . . . . . . . . . . . . . . . . . . . . . . . 0.9 Watts Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . VS Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . 1.2 V Output Short Circuit Duration . . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves Storage Temperature Range N, R . . . . . . . . . -65C to +125C Operating Temperature Range . . . . . . . . . . . . -40C to +85C Lead Temperature Range (Soldering, 10 sec) . . . . . . . +300C The maximum power that can be safely dissipated by the AD812 is limited by the associated rise in junction temperature. The maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature of the plastic, about 150C. Exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175C for an extended period can result in device failure. While the AD812 is internally short circuit protected, this may not be sufficient to guarantee that the maximum junction temperature (150 degrees) is not exceeded under all conditions. To ensure proper operation, it is important to observe the derating curves. NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air: 8-lead plastic package: JA = 90C/Watt; 8-lead SOIC package: JA = 150C/Watt. It must also be noted that in high (noninverting) gain configurations (with low values of gain resistor), a high level of input overdrive can result in a large input error current, which may result in a significant power dissipation in the input stage. This power must be included when computing the junction temperature rise due to total internal power. ORDERING GUIDE AD812AN -40C to +85C AD812AR -40C to +85C AD812AR-REEL AD812AR-REEL7 Package Description Package Option 2.0 MAXIMUM POWER DISSIPATION - Watts Temperature Range Model 8-Lead Plastic DIP N-8 8-Lead Plastic SOIC SO-8 13" Reel 7" Reel METALIZATION PHOTO Dimensions shown in inches and (mm). 0.0783 (1.99) V+ 8 OUT2 7 -IN2 6 TJ = +1508C 8-LEAD MINI-DIP PACKAGE 1.5 1.0 8-LEAD SOIC PACKAGE 0.5 0 -50 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 AMBIENT TEMPERATURE - 8C 5 +IN2 Figure 3. Plot of Maximum Power Dissipation vs. Temperature 0.0539 (1.37) 4 V- 1 OUT1 2 -IN1 3 +IN1 4 V- CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD812 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. B -5- WARNING! ESD SENSITIVE DEVICE AD812-Typical Performance Characteristics 16 14 TOTAL SUPPLY CURRENT - mA COMMON-MODE VOLTAGE RANGE - 6Volts 20 15 10 5 0 0 5 10 15 SUPPLY VOLTAGE - 6Volts VS = 615V 12 10 VS = 65V 8 6 4 -60 20 -40 -20 0 20 40 60 80 100 120 140 JUNCTION TEMPERATURE - 8C Figure 4. Input Common-Mode Voltage Range vs. Supply Voltage Figure 7. Total Supply Current vs. Junction Temperature 10 20 TA = +25 C TOTAL SUPPLY CURRENT - mA OUTPUT VOLTAGE - V p-p NO LOAD 15 10 RL = 150V 5 9 8 7 6 5 0 0 5 10 15 SUPPLY VOLTAGE - 6Volts 0 2 4 20 Figure 5. Output Voltage Swing vs. Supply Voltage 10 12 6 8 SUPPLY VOLTAGE - 6Volts 14 16 Figure 8. Total Supply Current vs. Supply Voltage 30 25 615V SUPPLY 20 INPUT BIAS CURRENT - mA OUTPUT VOLTAGE - Volts p-p 25 20 15 10 65V SUPPLY 15 -IB, VS = 65V 10 5 0 +IB, VS = 65V, 615V -5 -10 -IB, VS = 615V -15 5 -20 0 10 100 1k LOAD RESISTANCE - V -25 -60 10k Figure 6. Output Voltage Swing vs. Load Resistance -40 -20 0 20 40 60 80 100 JUNCTION TEMPERATURE - 8C 120 140 Figure 9. Input Bias Current vs. Junction Temperature -6- REV. B AD812 70 4 2 60 OUTPUT CURRENT - mA INPUT OFFSET VOLTAGE - mV VS = 65V 0 -2 -4 VS = 615V -6 -8 -10 50 40 30 -12 -14 -16 -60 20 -40 -20 0 20 40 60 80 100 JUNCTION TEMPERATURE - 8C 120 140 Figure 10. Input Offset Voltage vs. Junction Temperature 10 15 SUPPLY VOLTAGE - 6Volts 20 1k CLOSED-LOOP OUTPUT RESISTANCE - V SHORT CIRCUIT CURRENT - mA 5 Figure 13. Linear Output Current vs. Supply Voltage 160 140 120 100 SOURCE 80 60 -40 -20 0 20 40 60 80 100 JUNCTION TEMPERATURE - 8C G = +2 100 VS = 615V SINK 40 -60 0 120 10 1 65VS 0.1 615VS 0.01 10k 140 Figure 11. Short Circuit Current vs. Junction Temperature 100k 1M FREQUENCY - Hz 10M 100M Figure 14. Closed-Loop Output Resistance vs. Frequency 80 30 70 25 OUTPUT VOLTAGE - V p-p OUTPUT CURRENT - mA VS = 615V 60 50 VS = 65V 40 VS = 615V -40 -20 0 20 40 60 80 100 120 10 0 100k 140 JUNCTION TEMPERATURE - 8C Figure 12. Linear Output Current vs. Junction Temperature REV. B RL = 1kV 15 VS = 65V 5 30 20 -60 20 1M 10M FREQUENCY - Hz 100M Figure 15. Large Signal Frequency Response -7- AD812 100 0 120 VOLTAGE NOISE NONINVERTING INPUT CURRENT NOISE 100 1k FREQUENCY - Hz TRANSIMPEDANCE - dB 10 10 CURRENT NOISE - pA/ Hz VOLTAGE NOISE - nV/ Hz -90 INVERTING INPUT CURRENT NOISE 1 10 1 100k 10k -135 100 GAIN 80 100k 1M FREQUENCY - Hz 10M 100M Figure 19. Open-Loop Transimpedance vs. Frequency (Relative to 1 ) -30 681V VIN VOUT 681V 70 G = +2 VS = 2V p-p VS = 615V ; RL = 1kV 681V HARMONIC DISTORTION - dBc 80 COMMON-MODE REJECTION - dB VS = 615V VS = 3V 40 10k 90 681V 60 50 VS = 615V 40 VS = 3V 30 -50 VS = 65V ; RL = 150V -70 VS = 65V VS = 615V 2ND HARMONIC -90 3RD HARMONIC -110 20 2ND 100k 1M FREQUENCY - Hz 10M 3RD -130 100M 1k Figure 17. Common-Mode Rejection vs. Frequency 10k 100k 1M FREQUENCY - Hz 10M 100M Figure 20. Harmonic Distortion vs. Frequency 80 10 70 8 615V OUTPUT SWING FROM 6V TO 0 POWER SUPPLY REJECTION - dB -180 VS = 3V 60 Figure 16. Input Current and Voltage Noise vs. Frequency 10 10k -45 VS = 615V PHASE PHASE - Degrees 100 60 50 61.5V 40 30 20 GAIN = -1 VS = 615V 6 4 2 0 1% 0.1% 0.025% -2 -4 -6 10 -8 0 10k 100k 1M FREQUENCY - Hz 10M -10 20 100M Figure 18. Power Supply Rejection vs. Frequency 30 40 SETTLING TIME - ns 50 60 Figure 21. Output Swing and Error vs. Settling Time -8- REV. B AD812 1400 1400 G = +1 VS = 615V RL = 500V G = +1 1200 1000 1000 SLEW RATE - V/ms SLEW RATE - V/ms 1200 G = +2 800 G = +10 600 400 G = +2 800 G = +10 600 400 G = -1 G = -1 200 200 0 0 0 1 2 3 4 5 6 7 8 10 9 0 3.0 1.5 Figure 22. Slew Rate vs. Output Step Size 9.0 10.5 12.0 13.5 15.0 20ns 100 90 VIN 90 VIN 10 VOUT 10 VOUT 0% 0% 500mV 2V Figure 26. Small Signal Pulse Response, Gain = +1, (RF = 750 , RL = 150 , VS = 5 V) Figure 23. Large Signal Pulse Response, Gain = +1, (RF = 750 , RL = 150 , VS = 5 V) 65V 1 GAIN -90 -180 5V -270 0 3V -1 160 VS = 615V -2 65V -3 G = +1 RL = 150V 180 -3dB BANDWIDTH - MHz VS = 615V 0 200 PHASE SHIFT - Degrees G = +1 RL = 150V PHASE CLOSED-LOOP GAIN - dB 7.5 500mV 100 5V RF = 750V 140 RF = 866V 120 PEAKING 1dB 100 PEAKING 80 0.2dB 60 40 -4 3V 20 -5 0 -6 1 10 100 FREQUENCY - MHz 0 1000 2 4 6 8 10 12 14 16 18 20 SUPPLY VOLTAGE - 6Volts Figure 24. Closed-Loop Gain and Phase vs. Frequency, G = +1 REV. B 6.0 Figure 25. Maximum Slew Rate vs. Supply Voltage 50ns 2V 4.5 SUPPLY VOLTAGE - 6Volts OUTPUT STEP SIZE - Vp-p Figure 27. -3 dB Bandwidth vs. Supply Voltage, G = +1 -9- AD812 100 100 90 VIN 90 VIN 10 VOUT 10 VOUT 0% 0% 500mV 5V Figure 28. Large Signal Pulse Response, Gain = +10, (RF = 357 , RL = 500 , VS = 15 V) -90 65V 5V -180 1 3V GAIN -270 0 VS = 615V -1 PHASE 5V -2 3V -3 -4 65V -5 -6 1 10 100 FREQUENCY - MHz 1000 Figure 29. Closed-Loop Gain and Phase vs. Frequency, Gain = +10, RL = 150 100 G = +10 RL = 1kV VS = 615V 0 65V 3V -90 5V 1 -180 GAIN 0 -270 -1 -360 5V -2 PHASE SHIFT - Degrees 0 CLOSED-LOOP GAIN (NORMALIZED) - dB G = +10 RL = 150V VS = 615V Figure 31. Small Signal Pulse Response, Gain = +10, (RF = 357 , RL = 150 , VS = 5 V) PHASE SHIFT - Degrees CLOSED-LOOP GAIN (NORMALIZED) - dB PHASE VS = 615V 3V -3 65V -4 -5 -6 1 10 100 FREQUENCY - MHz 1000 Figure 32. Closed-Loop Gain and Phase vs. Frequency, Gain = +10, RL = 1 k 110 G = +10 RL= 150V 90 G = +10 RL = 1kV 100 80 90 -3dB BANDWIDTH - MHz -3dB BANDWIDTH - MHz 20ns 50mV 50ns 500mV 70 PEAKING 1dB RF = 357V 60 RF = 154V 50 RF = 649V 40 30 20 RF = 357V 80 70 RF = 154V 60 RF = 649V 50 40 30 10 20 0 0 2 4 6 8 10 12 14 SUPPLY VOLTAGE - 6Volts 16 18 10 20 0 Figure 30. -3 dB Bandwidth vs. Supply Voltage, Gain = +10, RL = 150 2 4 6 8 10 12 14 SUPPLY VOLTAGE - 6Volts 16 18 20 Figure 33. -3 dB Bandwidth vs. Supply Voltage, Gain = +10, RL = 1 k -10- REV. B AD812 50ns 2V 500mV 100 90 VIN 90 VIN 10 VOUT 10 VOUT 0% 0% 500mV 2V 0 -90 3V 1 -180 GAIN 0 -270 -1 VS = 615V 65V -4 5V -5 3V -6 1 10 100 FREQUENCY - MHz 1000 Figure 35. Closed-Loop Gain and Phase vs. Frequency, Gain = -1, RL = 150 -90 -180 5V 1 GAIN 3V -270 0 -1 VS = 615V -2 65V -3 5V -4 3V -5 -6 1 10 100 FREQUENCY - MHz 1000 100 G = -1 RL = 150V 120 G = -10 RL = 1kV 90 RF = 681V 110 80 PEAKING # 1.0dB 100 -3dB BANDWIDTH - MHz -3dB BANDWIDTH - MHz 0 Figure 38. Closed-Loop Gain and Phase vs. Frequency, Gain = -10, RL = 1 k 130 RF = 715V 90 80 PEAKING # 0.2dB 70 60 50 RF = 357V 70 60 RF = 154V RF = 649V 50 40 30 20 40 10 0 2 4 6 8 10 12 14 16 18 0 20 0 SUPPLY VOLTAGE - 6Volts Figure 36. -3 dB Bandwidth vs. Supply Voltage, Gain = -1, RL = 150 REV. B G = -10 RL = 1kV 65V -2 -3 VS = 615V PHASE PHASE SHIFT - Degrees CLOSED-LOOP GAIN (NORMALIZED) - dB G = -1 RL = 150V 65V 5V CLOSED-LOOP GAIN (NORMALIZED) - dB VS = 615V PHASE Figure 37. Small Signal Pulse Response, Gain = -1, (RF = 750 , RL = 150 , VS = 5 V) PHASE SHIFT - Degrees Figure 34. Large Signal Pulse Response, Gain = -1, (RF = 750 , RL = 150 , VS = 5 V) 30 20ns 100 2 4 6 8 10 12 14 SUPPLY VOLTAGE - 6Volts 16 18 20 Figure 39. -3 dB Bandwidth vs. Supply Voltage, Gain = -10, RL = 1 k -11- AD812 General Considerations To estimate the -3 dB bandwidth for closed-loop gains or feedback resistors not listed in the above table, the following two pole model for the AD812 many be used: The AD812 is a wide bandwidth, dual video amplifier which offers a high level of performance on less than 5.5 mA per amplifier of quiescent supply current. It is designed to offer outstanding performance at closed-loop inverting or noninverting gains of one or greater. ACL = Built on a low cost, complementary bipolar process, and achieving bandwidth in excess of 100 MHz, differential gain and phase errors of better than 0.1% and 0.1 (into 150 ), and output current greater than 40 mA, the AD812 is an exceptionally efficient video amplifier. Using a conventional current feedback architecture, its high performance is achieved through careful attention to design details. where: Choice of Feedback and Gain Resistors Because it is a current feedback amplifier, the closed-loop bandwidth of the AD812 depends on the value of the feedback resistor. The bandwidth also depends on the supply voltage. In addition, attenuation of the open-loop response when driving load resistors less than about 250 will affect the bandwidth. Table I contains data showing typical bandwidths at different supply voltages for some useful closed-loop gains when driving a load of 150 . (Bandwidths will be about 20% greater for load resistances above a few hundred ohms.) ( ) Table II. Two-Pole Model Parameters at Various Supply Voltages VS rIN () CT (pF) f2 (MHz) 15 5 +5 +3 85 90 105 115 2.5 3.8 4.8 5.5 150 125 105 95 VS (V) Gain RF () BW (MHz) 15 +1 +2 +10 -1 -10 866 715 357 715 357 145 100 65 100 60 where: +1 +2 +10 -1 -10 750 681 154 715 154 90 65 45 70 45 and: +1 +2 +10 -1 -10 750 681 154 715 154 60 50 35 50 35 +1 +2 +10 -1 -10 750 681 154 715 154 50 40 30 40 25 +3 ) As discussed in many amplifier and electronics textbooks (such as Roberge's Operational Amplifiers: Theory and Practice), the -3 dB bandwidth for the 2-pole model can be obtained as: Table I. -3 dB Bandwidth vs. Closed-Loop Gain and Feedback Resistor (RL = 150 ) +5 ( RF + GrIN CT + S RF + GrIN CT + 1 S2 2f 2 ACL = closed-loop gain G = 1 + RF /RG rIN = input resistance of the inverting input CT = "transcapacitance," which forms the open-loop dominant pole with the tranresistance RF = feedback resistor RG = gain resistor f2 = frequency of second (nondominant) pole S = 2 j f Appropriate values for the model parameters at different supply voltages are listed in Table II. Reasonable approximations for these values at supply voltages not found in the table can be obtained by a simple linear interpolation between those tabulated values which "bracket" the desired condition. The choice of feedback resistor is not critical unless it is important to maintain the widest, flattest frequency response. The resistors recommended in the table are those (metal film values) that will result in the widest 0.1 dB bandwidth. In those applications where the best control of the bandwidth is desired, 1% metal film resistors are adequate. Wider bandwidths can be attained by reducing the magnitude of the feedback resistor (at the expense of increased peaking), while peaking can be reduced by increasing the magnitude of the feedback resistor. 5 G f3 = fN [1 - 2d2 + (2 - 4d2 + 4d4 )1/2]1/2 1/ 2 f2 fN = R + Gr C IN T F ( ) d = (1/2) [f2 (RF + GrIN ) CT]1/2 This model will predict -3 dB bandwidth within about 10 to 15% of the correct value when the load is 150 . However, it is not an accurate enough to predict either the phase behavior or the frequency response peaking of the AD812. Printed Circuit Board Layout Guidelines As with all wideband amplifiers, printed circuit board parasitics can affect the overall closed-loop performance. Most important for controlling the 0.1 dB bandwidth are stray capacitances at the output and inverting input nodes. Increasing the space between signal lines and ground plane will minimize the coupling. Also, signal lines connecting the feedback and gain resistors should be kept short enough that their associated inductance does not cause high frequency gain errors. -12- REV. B AD812 The input and output signal return paths must also be kept from overlapping. Since ground connections are not of perfectly zero impedance, current in one ground return path can produce a voltage drop in another ground return path if they are allowed to overlap. Power Supply Bypassing Adequate power supply bypassing can be very important when optimizing the performance of high speed circuits. Inductance in the supply leads can (for example) contribute to resonant circuits that produce peaking in the amplifier's response. In addition, if large current transients must be delivered to a load, then large (greater than 1 F) bypass capacitors are required to produce the best settling time and lowest distortion. Although 0.1 F capacitors may be adequate in some applications, more elaborate bypassing is required in other cases. Electric field coupling external to (and across) the package can be reduced by arranging for a narrow strip of ground plane to be run between the pins (parallel to the pin rows). Doing this on both sides of the board can reduce the high frequency crosstalk by about 5 dB or 6 dB. When multiple bypass capacitors are connected in parallel, it is important to be sure that the capacitors themselves do not form resonant circuits. A small (say 5 ) resistor may be required in series with one of the capacitors to minimize this possibility. Driving Capacitive Loads As discussed below, power supply bypassing can have a significant impact on crosstalk performance. Achieving Low Crosstalk Measured crosstalk from the output of amplifier 2 to the input of amplifier 1 of the AD812 is shown in Figure 40. The crosstalk from the output of amplifier 1 to the input of amplifier 2 is a few dB better than this due to the additional distance between critical signal nodes. When used with the appropriate output series resistor, any load capacitance can be driven without peaking or oscillation. In most cases, less than 50 is all that is needed to achieve an extremely flat frequency response. As illustrated in Figure 44, the AD812 can be very attractive for driving largely capacitive loads. In this case, the AD812's high output short circuit current allows for a 150 V/s slew rate when driving a 510 pF capacitor. RF +VS A carefully laid-out PC board should be able to achieve the level of crosstalk shown in the figure. The most significant contributors to difficulty in achieving low crosstalk are inadequate power supply bypassing, overlapped input and/or output signal paths, and capacitive coupling between critical nodes. 0.1mF 1.0mF RG 8 RS AD812 The bypass capacitors must be connected to the ground plane at a point close to and between the ground reference points for the two loads. (The bypass of the negative power supply is particularly important in this regard.) There are two amplifiers in the package, and low impedance signal return paths must be provided for each load. (Using a parallel combination of 1 F, 0.1 F, and 0.01 F bypass capacitors will help to achieve optimal crosstalk.) VIN 4 VO 1.0mF CL RL RT 0.1mF -VS Figure 41. Circuit for Driving a Capacitive Load -10 VS = 65V G = +2 RF = 750V RL = 1kV CL = 10pF -20 RL = 150V -30 12 CLOSED-LOOP GAIN - dB CROSSTALK - dB -40 -50 -60 -70 -80 -90 1M 10M RS = 30V 3 RS = 50V 0 -3 1 100M FREQUENCY - Hz 10 100 FREQUENCY - MHz 1000 Figure 42. Response to a Small Load Capacitor at 5 V Figure 40. Crosstalk vs. Frequency REV. B RS = 0 6 -6 -100 -110 100k 9 -13- AD812 VS = 615V G = +2 RF = 750V RL = 1kV 100 12 CLOSED-LOOP GAIN - dB 50ns 1V 90 VIN 10 VOUT 9 6 CL = 150pF, RS = 30V 3 0 0% -3 CL = 510pF, RS = 15V -6 2V -9 1 10 100 FREQUENCY - MHz 1000 Figure 45. 6 dB Overload Recovery; G = 10, RL = 500 , VS = 5 V Figure 43. Response to Large Load Capacitor, VS = 15 V 5V In the case of high gains with very high levels of input overdrive, a longer recovery time may occur. For example, if the input common-mode voltage range is exceeded in a gain of +10, the recovery time will be on the order of 100 ns. This is primarily due to current overloading of the input stage. 100ns VIN 100 90 As noted in the warning under "Maximum Power Dissipation," a high level of input overdrive in a high noninverting gain circuit can result in a large current flow in the input stage. For differential input voltages of less than about 1.25 V, this will be internally limited to less than 20 mA (decreasing with supply voltage). For input overdrives which result in higher differential input voltages, power dissipation in the input stage must be considered. It is recommended that external diode clamps be used in cases where the differential input voltage is expected to exceed 1.25 V. VOUT 10 0% 5V Figure 44. Pulse Response of Circuit of Figure 41 with CL = 510 pF, RL = 1 k, RF = RG = 715 , RS = 15 High Performance Video Line Driver Overload Recovery There are three important overload conditions to consider. They are due to input common mode voltage overdrive, input current overdrive, and output voltage overdrive. When the amplifier is configured for low closed-loop gains, and its input common-mode voltage range is exceeded, the recovery time will be very fast, typically under 10 ns. When configured for a higher gain, and overloaded at the output, the recovery time will also be short. For example, in a gain of +10, with 6 dB of input overdrive, the recovery time of the AD812 is about 10 ns. At a gain of +2, the AD812 makes an excellent driver for a backterminated 75 video line. Low differential gain and phase errors and wide 0.1 dB bandwidth can be realized over a wide range of power supply voltage. Outstanding gain and group delay matching are also attainable over the full operating supply voltage range. RG RF +VS 0.1mF 75V 75V CABLE 8 75V CABLE VOUT AD812 VIN 75V 4 75V 0.1mF -VS Figure 46. Gain of +2 Video Line Driver (RF = RG from Table I) -14- REV. B G = +2 RL = 150V 0 -90 3V VS = 615V 5V 1 65V GAIN CLOSED-LOOP GAIN - dB -180 -270 0 5V -1 3V -2 0.4 0.2 VS = 615V -3 65V -4 G = +2 RL = 150V 0.3 NORMALIZED GAIN - dB 90 PHASE PHASE SHIFT - Degrees AD812 0.1 0 -0.1 -0.2 VS = 615V -0.3 65V -0.4 5V -5 -0.5 3V -0.6 100k -6 1 10 100 FREQUENCY -MHz 1000 Figure 47. Closed-Loop Gain and Phase vs. Frequency for the Line Driver 1.0 RF = 590V G = +2 RL = 150V 110 RL = 150V VS = 3V 0.6 RF = 750V PEAKING # 1dB G = +2 0.8 RF = 715V 100 RF = 681V 90 0.4 GAIN MATCH - dB -3dB BANDWIDTH - MHz 100M Figure 50. Fine-Scale Gain Flatness vs. Frequency, Gain = +2, RL = 150 120 80 NO PEAKING 70 60 50 0.2 0 RF = 715V -0.4 40 -0.6 30 -0.8 0 2 4 6 8 10 12 14 SUPPLY VOLTAGE - 6Volts 16 18 -1.0 20 1 Figure 48. -3 dB Bandwidth vs. Supply Voltage, Gain = +2, RL = 150 0.02 0.06 1000 DELAY 8 3V GROUP DELAY - ns 0.04 DIFFERENTIAL GAIN 10 100 FREQUENCY - MHz Figure 51. Closed-Loop Gain Matching vs. Frequency, Gain = +2, RL = 150 DIFFERENTIAL GAIN - % 0.06 0.08 VS = 615V -0.2 20 DIFFERENTIAL PHASE - Degrees 1M 10M FREQUENCY - Hz DIFFERENTIAL PHASE 0.04 6 5V 4 65V 615V 2 0 DELAY MATCHING 0.4 0.2 VS = 3V TO 615V 0 0.02 -0.2 -0.4 100k 0 5 6 7 8 9 10 11 12 13 14 15 SUPPLY VOLTAGE - 6Volts 10M FREQUENCY - Hz 100M Figure 52. Group Delay and Group Delay Matching vs. Frequency, G = +2, RL = 150 Figure 49. Differential Gain and Phase vs. Supply Voltage, Gain = +2, RL = 150 REV. B 1M -15- 90 The AD812 will operate with total supply voltages from 36 V down to 2.4 V. With proper biasing (see Figure 53), it can be an outstanding single supply video amplifier. Since the input and output voltage ranges extend to within 1 volt of the supply rails, it will handle a 1.3 V p-p signal on a single 3.3 V supply, or a 3 V p-p signal on a single 5 V supply. The small signal, 0.1 dB bandwidths will exceed 10 MHz in either case, and the large signal bandwidths will exceed 6 MHz. PHASE CLOSED-LOOP GAIN - dB 0 -180 -0.5 -270 -1.0 -1.5 -2.0 -2.5 -3.0 -3.5 1 10 100 FREQUENCY - MHz 1000 Figure 54. Closed-Loop Gain and Phase vs. Frequency, Circuit of Figure 53 649V C3 30mF -90 GAIN The capacitively coupled cable driver in Figure 53 will achieve outstanding differential gain and phase errors of 0.07% and 0.06 degrees respectively on a single 5 V supply. Resistor R2, in this circuit, is selected to optimize the differential gain and phase by operating the amplifier in its most linear region. To optimize the circuit for a 3 V supply, a value of 8 k is recommended for R2. 649V 0 VS = 5V 0.5 C1859b-0-9/98 Operation Using a Single Supply PHASE SHIFT - Degrees AD812 R3 1kV +VS C2 1mF 1V R1 9kV COUT 8 C1 2mF 47mF 75V 75V CABLE 100 VIN 90 VOUT AD812 VIN 50ns 75V 4 R2 11.8kV VOUT Figure 53. Biasing for Single Supply Operation 10 0% 500mV Figure 55. Pulse Response of the Circuit of Figure 53 with VS = 5 V OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 8-Lead Plastic SOIC (SO-8) 0.39 (9.91) 8 0.1968 (5.00) 0.1890 (4.80) 5 0.25 (6.35) 1 4 PIN 1 0.165 60.01 (4.19 60.25) 0.125 (3.18) MIN 0.060 (1.52) 0.015 (0.38) SEATING 0.018 60.003 0.10 0.033 (0.84) PLANE (0.46 +0.08) (2.54) NOM BSC 0.1574 (4.00) 0.1497 (3.80) 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) PIN 1 0.0098 (0.25) 0.0040 (0.10) 0.015 (0.381) 0.008 (0.204) 8 5 1 4 0.2440 (6.20) 0.2284 (5.80) 0.0688 (1.75) 0.0532 (1.35) 0.0500 0.0192 (0.49) SEATING (1.27) 0.0098 (0.25) PLANE BSC 0.0138 (0.35) 0.0075 (0.19) -16- 0.0196 (0.50) 3 458 0.0099 (0.25) 88 08 0.0500 (1.27) 0.0160 (0.41) REV. B PRINTED IN U.S.A. 8-Lead Plastic DIP (N-8) Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Analog Devices Inc.: AD812AR-EBZ