AD812
a
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
Dual, Current Feedback
Low Power Op Amp
PIN CONFIGURATION
8-Lead Plastic
Mini-DIP and SOIC
OUT1
–IN1
+IN1
V+
OUT2
–IN2
+IN2
V–
AD812
+
+
4
3
2
1
5
6
7
8
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 World Wide Web Site: http://www.analog.com
Fax: 781/326-8703 © Analog Devices, Inc., 1998
REV. B
FEATURES
Two Video Amplifiers in One 8-Lead SOIC Package
Optimized for Driving Cables in Video Systems
Excellent Video Specifications (RL = 150 ⍀):
Gain Flatness 0.1 dB to 40 MHz
0.02% Differential Gain Error
0.02ⴗ Differential Phase Error
Low Power
Operates on Single +3 V Supply
5.5 mA/Amplifier Max Power Supply Current
High Speed
145 MHz Unity Gain Bandwidth (3 dB)
1600 V/␮s Slew Rate
Easy to Use
50 mA Output Current
Output Swing to 1 V of Rails (150 ⍀ Load)
APPLICATIONS
Video Line Driver
Professional Cameras
Video Switchers
Special Effects
PRODUCT DESCRIPTION
The AD812 is a low power, single supply, dual video amplifier.
Each of the amplifiers have 50 mA of output current and are
optimized for driving one back-terminated video load (150 Ω)
each. Each amplifier is a current feedback amplifier and fea-
tures gain flatness of 0.1 dB to 40 MHz while offering differen-
tial gain and phase error of 0.02% and 0.02°. This makes the
AD812 ideal for professional video electronics such as cameras
and video switchers.
–0.1
–0.6 1M 100M10M
–0.2
–0.3
–0.4
–0.5
0
0.1
NORMALIZED GAIN – dB
100k FREQUENCY – Hz
0.2
0.3
0.4
G = +2
R
L
= 150V
5V
3V
V
S
= 615V
65V
Figure 1. Fine-Scale Gain Flatness vs. Frequency, Gain
= +2, R
L
= 150
Ω
The AD812 offers low power of 4.0 mA per amplifier max (V
S
=
+5 V) and can run on a single +3 V power supply. The outputs
of each amplifier swing to within one volt of either supply rail to
easily accommodate video signals of 1 V p-p. Also, at gains of
+2 the AD812 can swing 3 V p-p on a single +5 V power sup-
ply. All this is offered in a small 8-lead plastic DIP or 8-lead
SOIC package. These features make this dual amplifier ideal
for portable and battery powered applications where size and
power is critical.
The outstanding bandwidth of 145 MHz along with 1600 V/µs
of slew rate make the AD812 useful in many general purpose
high speed applications where a single +5 V or dual power sup-
plies up to ±15 V are available. The AD812 is available in the
industrial temperature range of –40°C to +85°C.
15
0.06
0.02
6
0.04
5
0.08
14121110 13987 SUPPLY VOLTAGE – 6Volts
DIFFERENTIAL PHASE – Degrees
0.06
0.02
0.04
DIFFERENTIAL GAIN – %
0
DIFFERENTIAL GAIN
DIFFERENTIAL PHASE
Figure 2. Differential Gain and Phase vs. Supply Voltage,
Gain = +2, R
L
= 150
Ω
Model AD812A
Conditions V
S
Min Typ Max Units
DYNAMIC PERFORMANCE
–3 dB Bandwidth G = +2, No Peaking ±5 V 50 65 MHz
±15 V 75 100 MHz
Gain = +1 ±15 V 100 145 MHz
Bandwidth for 0.1 dB Flatness G = +2 ±5 V 20 30 MHz
±15 V 25 40 MHz
Slew Rate
1
G = +2, R
L
= 1 kΩ±5 V 275 425 V/µs
20 V Step ±15 V 1400 1600 V/µs
G = –1, R
L
= 1 kΩ±5 V 250 V/µs
±15 V 600 V/µs
Settling Time to 0.1% G = –1, R
L
= 1 kΩ
V
O
= 3 V Step ±5 V 50 ns
V
O
= 10 V Step ±15 V 40 ns
NOISE/HARMONIC PERFORMANCE
Total Harmonic Distortion f
C
= 1 MHz, R
L
= 1 kΩ±15 V –90 dBc
Input Voltage Noise f = 10
kHz ±5 V, ±15 V 3.5 nV/√Hz
Input Current Noise f = 10 kHz, +In ±5 V, ±15 V 1.5 pA/√Hz
f = 10 kHz, –In ±5 V, ±15 V 18 pA/√Hz
Differential Gain Error NTSC, G = +2, R
L
= 150 Ω±5 V 0.05 0.1 %
±15 V 0.02 0.06 %
Differential Phase Error ±5 V 0.07 0.15 Degrees
±15 V 0.02 0.06 Degrees
DC PERFORMANCE
Input Offset Voltage ±5 V, ±15 V 2 5 mV
T
MIN
–T
MAX
12 mV
Offset Drift ±5 V, ±15 V 15 µV/°C
–Input Bias Current ±5 V, ±15 V 7 25 µA
T
MIN
–T
MAX
38 µA
+Input Bias Current ±5 V, ±15 V 0.3 1.5 µA
T
MIN
–T
MAX
2.0 µA
Open-Loop Voltage Gain V
O
= ±2.5 V, R
L
= 150 Ω±5 V 68 76 dB
T
MIN
–T
MAX
69 dB
V
O
= ±10 V, R
L
= 1 kΩ±15 V 76 82 dB
T
MIN
–T
MAX
75 dB
Open-Loop Transresistance V
O
= ±2.5 V, R
L
= 150 Ω±5 V 350 550 kΩ
T
MIN
–T
MAX
270 kΩ
V
O
= ±10 V, R
L
= 1 kΩ±15 V 450 800 kΩ
T
MIN
–T
MAX
370 kΩ
INPUT CHARACTERISTICS
Input Resistance +Input ±15 V 15 MΩ
–Input 65 Ω
Input Capacitance +Input 1.7 pF
Input Common Mode ±5 V 4.0 ±V
Voltage Range ±15 V 13.5 ±V
Common-Mode Rejection Ratio
Input Offset Voltage V
CM
= ±2.5 V ±5 V 51 58 dB
–Input Current 23.0µA/V
+Input Current 0.07 0.15 µA/V
Input Offset Voltage V
CM
= ±12 V ±15 V 55 60 dB
–Input Current 1.5 3.3 µA/V
+Input Current 0.05 0.15 µA/V
(@ TA = +25ⴗC, RL = 150 ⍀, unless otherwise noted)
Dual Supply
AD812–SPECIFICATIONS
–2–
REV. B
Model AD812A
Conditions V
S
Min Typ Max Units
OUTPUT CHARACTERISTICS
Output Voltage Swing R
L
= 150 Ω, T
MIN
–T
MAX
±5 V 3.5 3.8 ±V
R
L
= 1 kΩ, T
MIN
–T
MAX
±15 V 13.6 14.0 ±V
Output Current ±5 V 30 40 mA
±15 V 40 50 mA
Short Circuit Current G = +2, R
F
= 715 Ω±15 V 100 mA
V
IN
= 2 V
Output Resistance Open-Loop ±15 V 15 Ω
MATCHING CHARACTERISTICS
Dynamic
Crosstalk G = +2, f = 5 MHz ±5 V, ±15 V –75 dB
Gain Flatness Match G = +2, f = 40 MHz ±15 V 0.1 dB
DC
Input offset Voltage T
MIN
–T
MAX
±5 V, ±15 V 0.5 3.6 mV
–Input Bias Current T
MIN
–T
MAX
±5 V, ±15 V 2 25 µA
POWER SUPPLY
Operating Range ±1.2 ±18 V
Quiescent Current Per Amplifier ±5 V 3.5 4.0 mA
±15 V 4.5 5.5 mA
T
MIN
–T
MAX
±15 V 6.0 mA
Power Supply Rejection Ratio
Input Offset Voltage V
S
= ±1.5 V to ±15 V 70 80 dB
–Input Current 0.3 0.6 µA/V
+Input Current 0.005 0.05 µA/V
NOTES
1
Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain.
Specifications subject to change without notice.
Single Supply
Model AD812A
Conditions V
S
Min Typ Max Units
DYNAMIC PERFORMANCE
–3 dB Bandwidth G = +2, No Peaking +5 V 35 50 MHz
+3 V 30 40 MHz
Bandwidth for 0.1 dB
Flatness G = +2 +5 V 13 20 MHz
+3 V 10 18 MHz
Slew Rate
1
G = +2, R
L
= 1 kΩ+5 V 125 V/µs
+3 V 60 V/µs
NOISE/HARMONIC PERFORMANCE
Input Voltage Noise f = 10
kHz +5 V, +3 V 3.5 nV/√Hz
Input Current Noise f = 10 kHz, +In +5 V, +3 V 1.5 pA/√Hz
f = 10 kHz, –In +5 V, +3 V 18 pA/√Hz
Differential Gain Error
2
NTSC, G = +2, R
L
= 150 Ω+5 V 0.07 %
G = +1 +3 V 0.15 %
Differential Phase Error
2
G = +2 +5 V 0.06 Degrees
G = +1 +3 V 0.15 Degrees
AD812
REV. B
–3–
(@ TA = +25ⴗC, RL = 150 ⍀, unless otherwise noted)
AD812A
Model Conditions V
S
Min Typ Max Units
DC PERFORMANCE
Input Offset Voltage +5 V, +3 V 1.5 4.5 mV
T
MIN
–T
MAX
7.0 mV
Offset Drift +5 V, +3 V 7 µV/°C
–Input Bias Current +5 V, +3 V 2 20 µA
T
MIN
–T
MAX
30 µA
+Input Bias Current +5 V, +3 V 0.2 1.5 µA
T
MIN
–T
MAX
2.0 µA
Open-Loop Voltage Gain V
O
= +2.5 V p-p +5 V 67 73 dB
V
O
= +0.7 V p-p +3 V 70 dB
Open-Loop Transresistance V
O
= +2.5 V p-p +5 V 250 400 kΩ
V
O
= +0.7 V p-p +3 V 300 kΩ
INPUT CHARACTERISTICS
Input Resistance +Input +5 V 15 MΩ
–Input +5 V 90 Ω
Input Capacitance +Input 2 pF
Input Common Mode +5 V 1.0 4.0 V
Voltage Range +3 V 1.0 2.0 V
Common-Mode Rejection Ratio
Input Offset Voltage V
CM
= 1.25 V to 3.75 V +5 V 52 55 dB
–Input Current 35.5µA/V
+Input Current 0.1 0.2 µA/V
Input Offset Voltage V
CM
= 1 V to 2 V +3 V 52 dB
–Input Current 3.5 µA/V
+Input Current 0.1 µA/V
OUTPUT CHARACTERISTICS
Output Voltage Swing p-p R
L
= 1 kΩ, T
MIN
–T
MAX
+5 V 3.0 3.2 V p-p
R
L
= 150 Ω, T
MIN
–T
MAX
+5 V 2.8 3.1 V p-p
+3 V 1.0 1.3 V p-p
Output Current +5 V 20 30 mA
+3 V 15 25 mA
Short Circuit Current G = +2, R
F
= 715 Ω+5 V 40 mA
V
IN
= 1 V
MATCHING CHARACTERISTICS
Dynamic
Crosstalk G = +2, f = 5 MHz +5 V, +3 V –72 dB
Gain Flatness Match G = +2, f = 20 MHz +5 V, +3 V 0.1 dB
DC
Input offset Voltage T
MIN
–T
MAX
+5 V, +3 V 0.5 3.5 mV
–Input Bias Current T
MIN
–T
MAX
+5 V, +3 V 2 25 µA
POWER SUPPLY
Operating Range 2.4 36 V
Quiescent Current Per Amplifier +5 V 3.2 4.0 mA
+3 V 3.0 3.5 mA
T
MIN
–T
MAX
+5 V 4.5 mA
Power Supply Rejection Ratio
Input Offset Voltage V
S
= +3 V to +30 V 70 80 dB
–Input Current 0.3 0.6 µA/V
+Input Current 0.005 0.05 µA/V
TRANSISTOR COUNT 56
NOTES
1
Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain.
2
Single supply differential gain and phase are measured with the ac coupled circuit of Figure 53.
Specifications subject to change without notice.
AD812–SPECIFICATIONS
Single Supply (Continued)
REV. B
–4–
AD812
–5–
REV. B
MAXIMUM POWER DISSIPATION
The maximum power that can be safely dissipated by the
AD812 is limited by the associated rise in junction temperature.
The maximum safe junction temperature for the plastic encap-
sulated parts is determined by the glass transition temperature
of the plastic, about 150°C. Exceeding this limit temporarily
may cause a shift in parametric performance due to a change in
the stresses exerted on the die by the package. Exceeding a
junction temperature of 175°C for an extended period can result
in device failure.
While the AD812 is internally short circuit protected, this may
not be sufficient to guarantee that the maximum junction tem-
perature (150 degrees) is not exceeded under all conditions. To
ensure proper operation, it is important to observe the derating
curves.
It must also be noted that in high (noninverting) gain configura-
tions (with low values of gain resistor), a high level of input
overdrive can result in a large input error current, which may
result in a significant power dissipation in the input stage. This
power must be included when computing the junction tempera-
ture rise due to total internal power.
MAXIMUM POWER DISSIPATION – Watts
AMBIENT TEMPERATURE – 8C
2.0
1.5
0
–50 90–40 –30 –20 –10 0 10 20 30 50 60 70 8040
1.0
0.5
8-LEAD SOIC PACKAGE
8-LEAD MINI-DIP PACKAGE
T
J
= +1508C
Figure 3. Plot of Maximum Power Dissipation vs.
Temperature
ABSOLUTE MAXIMUM RATINGS
1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .±18 V
Internal Power Dissipation
2
Plastic (N) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts
Small Outline (R) . . . . . . . . . . . . . . . . . . . . . . . . . . 0.9 Watts
Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . ±V
S
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . ±1.2 V
Output Short Circuit Duration
. . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves
Storage Temperature Range N, R . . . . . . . . . –65°C to +125°C
Operating Temperature Range . . . . . . . . . . . . –40°C to +85°C
Lead Temperature Range (Soldering, 10 sec) . . . . . . . +300°C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for device in free air: 8-lead plastic package: θ
JA
= 90°C/Watt;
8-lead SOIC package: θ
JA
= 150°C/Watt.
ORDERING GUIDE
Temperature Package Package
Model Range Description Option
AD812AN –40°C to +85°C 8-Lead Plastic DIP N-8
AD812AR –40°C to +85°C 8-Lead Plastic SOIC SO-8
AD812AR-REEL 13" Reel
AD812AR-REEL7 7" Reel
METALIZATION PHOTO
Dimensions shown in inches and (mm).
V+
8OUT2
7–IN2
6
2
–IN1 3
+IN1 4
V–
1
OUT1
5 +IN2
4 V–
0.0783
(1.99)
0.0539
(1.37)
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD812 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
20
0020
15
5
5
10
10 15
SUPPLY VOLTAGE – 6Volts
COMMON-MODE VOLTAGE RANGE – 6Volts
Figure 4. Input Common-Mode Voltage Range vs. Supply
Voltage
20
0020
15
5
5
10
10 15
SUPPLY VOLTAGE – 6Volts
OUTPUT VOLTAGE – V p-p
R
L
= 150V
NO LOAD
Figure 5. Output Voltage Swing vs. Supply Voltage
30
15
010 100 10k1k
10
5
20
25
LOAD RESISTANCE – V
OUTPUT VOLTAGE – Volts p-p
615V SUPPLY
65V SUPPLY
Figure 6. Output Voltage Swing vs. Load Resistance
16
4
10
6
8
14
12
140–40–60 120806040 100200–20
JUNCTION TEMPERATURE – 8C
TOTAL SUPPLY CURRENT – mA
VS = 615V
VS = 65V
Figure 7. Total Supply Current vs. Junction Temperature
10
5
8
6
7
9
16
20 141210
864 SUPPLY VOLTAGE – 6Volts
TOTAL SUPPLY CURRENT – mA
T
A
= +25 C
Figure 8. Total Supply Current vs. Supply Voltage
25
–25
–10
–20
–15
5
–5
0
10
15
20
–60 140–40 120100806040200–20
INPUT BIAS CURRENT – mA
JUNCTION TEMPERATURE – 8C
–I
B
, V
S
= 615V
+I
B
, V
S
= 65V, 615V
–I
B
, V
S
= 65V
Figure 9. Input Bias Current vs. Junction Temperature
AD812–Typical Performance Characteristics
REV. B
–6–
AD812
–7–
REV. B
4
–16
–10
–14
–12
–4
–8
–6
–2
0
2
140–40–60 120100806040200–20
INPUT OFFSET VOLTAGE – mV
JUNCTION TEMPERATURE – 8C
V
S
= 615V
V
S
= 65V
Figure 10. Input Offset Voltage vs. Junction Temperature
160
40
100
60
80
140
120
140–40–60 120806040 100200–20 JUNCTION TEMPERATURE – 8C
SHORT CIRCUIT CURRENT – mA
SINK V
S
= 615V
SOURCE
Figure 11. Short Circuit Current vs. Junction Temperature
80
20
50
30
40
70
60
140–40–60 120806040 100200–20 JUNCTION TEMPERATURE – 8C
OUTPUT CURRENT – mA
VS = 615V
VS = 65V
Figure 12. Linear Output Current vs. Junction Temperature
70
20
50
30
40
60
20
50 1510
SUPPLY VOLTAGE – 6Volts
OUTPUT CURRENT – mA
Figure 13. Linear Output Current vs. Supply Voltage
100k 100M10M1M10k
0.01
1k
10
1
0.1
100
FREQUENCY – Hz
CLOSED-LOOP OUTPUT RESISTANCE – V
615V
S
65V
S
G = +2
Figure 14. Closed-Loop Output Resistance vs. Frequency
30
15
0100k 1M 100M10M
10
5
20
25
FREQUENCY – Hz
OUTPUT VOLTAGE – V p-p
V
S
= 615V
V
S
= 65V
R
L
= 1kV
Figure 15. Large Signal Frequency Response
AD812
REV. B
–8–
100
10
110 100 100k10k1k
FREQUENCY – Hz
VOLTAGE NOISE – nV/ Hz
100
10
1
CURRENT NOISE – pA/ Hz
INVERTING INPUT
CURRENT NOISE
VOLTAGE NOISE
NONINVERTING INPUT
CURRENT NOISE
Figure 16. Input Current and Voltage Noise vs. Frequency
10k 100k 100M10M1M
FREQUENCY – Hz
90
60
50
70
80
20
10
30
40
COMMON-MODE REJECTION – dB
681V
681V
V
OUT
V
IN
681V
681V
V
S
= 615V
V
S
= 3V
Figure 17. Common-Mode Rejection vs. Frequency
FREQUENCY – Hz
POWER SUPPLY REJECTION – dB
80
40
010k 100k 100M10M1M
20
60
50
30
10
70
615V
61.5V
Figure 18. Power Supply Rejection vs. Frequency
10k 100k 100M10M1M
FREQUENCY – Hz
100
40
120
60
80
TRANSIMPEDANCE – dB
0
–45
–90
–135
–180
PHASE – Degrees
PHASE
GAIN
V
S
= 3V V
S
= 615V
V
S
= 3V
V
S
= 615V
Figure 19. Open-Loop Transimpedance vs. Frequency
(Relative to 1
Ω
)
–30
FREQUENCY – Hz
HARMONIC DISTORTION – dBc
1k
–130 10k 100k 1M 10M 100M
–70
–50
–110
–90
G = +2
V
S
= 2V p-p
V
S
= 615V ; R
L
= 1kV
V
S
= 65V ; R
L
= 150V
V
S
= 65V
2
ND
HARMONIC
3
RD
HARMONIC
2
ND
3
RD
V
S
= 615V
Figure 20. Harmonic Distortion vs. Frequency
SETTLING TIME – ns
OUTPUT SWING FROM 6V TO 0
10
–10 60
–4
–8
–6
20
2
–2
0
4
6
8
4030 50
0.1%
GAIN = –1
VS = 615V
1% 0.025%
Figure 21. Output Swing and Error vs. Settling Time
AD812
–9–
REV. B
1400
010
600
200
1
400
0
1200
800
1000
98765432 OUTPUT STEP SIZE – Vp-p
SLEW RATE – V/ms
V
S
= 615V
R
L
= 500V
G = +1
G = +2
G = +10
G = –1
Figure 22. Slew Rate vs. Output Step Size
10
100
0%
2V 50ns
2V
V
IN
V
OUT
90
Figure 23. Large Signal Pulse Response, Gain = +1,
(R
F
= 750
Ω
, R
L
= 150
Ω
, V
S
=
±
5 V)
–1
–6 10 100
–2
–3
–4
–5
0
1
CLOSED-LOOP GAIN – dB
1FREQUENCY – MHz 1000
0
–90
–180
–270
PHASE SHIFT – Degrees
65V
5V
G = +1
R
L
= 150V
5V
3V
PHASE
GAIN
V
S
= 615V
65V
V
S
= 615V
3V
Figure 24. Closed-Loop Gain and Phase vs. Frequency,
G = +1
1400
015.0
600
200
1.5
400
0
1200
800
1000
13.512.010.59.07.56.04.53.0 SUPPLY VOLTAGE – 6Volts
SLEW RATE – V/ms
G = +2
G = +10
G = –1
G = +1
Figure 25. Maximum Slew Rate vs. Supply Voltage
10
90
100
0%
500mV 20ns
V
IN
V
OUT
500mV
Figure 26. Small Signal Pulse Response, Gain = +1,
(R
F
= 750
Ω
, R
L
= 150
Ω
, V
S
=
±
5 V)
200
020
60
20
2
40
0
120
80
100
140
160
180
1816141210864 SUPPLY VOLTAGE – 6Volts
–3dB BANDWIDTH – MHz
G = +1
R
L
= 150V
PEAKING 1dB
PEAKING 0.2dB
R
F
= 750V
R
F
= 866V
Figure 27. –3 dB Bandwidth vs. Supply Voltage, G = +1
AD812
REV. B
–10–
10
90
100
0%
500mV 50ns
5V
V
IN
V
OUT
Figure 28. Large Signal Pulse Response, Gain = +10,
(R
F
= 357
Ω
, R
L
= 500
Ω
, V
S
=
±
15 V)
1 10 1000100
FREQUENCY – MHz
–1
–6
1
0
–2
–3
–4
–5
CLOSED-LOOP GAIN (NORMALIZED) – dB
0
–90
–180
–270
PHASE SHIFT – Degrees
3V
5V
5V 65V
3V
65V
PHASE
GAIN
V
S
= 615V G = +10
R
L
= 150V
V
S
= 615V
Figure 29. Closed-Loop Gain and Phase vs. Frequency,
Gain = +10, R
L
= 150
Ω
216141210864 SUPPLY VOLTAGE – 6Volts
–3dB BANDWIDTH – MHz
30
20
60
40
50
70
80
90
100
10
001820
G = +10
R
L
= 150V
PEAKING 1dB
R
F
= 154V
R
F
= 357V
R
F
= 649V
Figure 30. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, R
L
= 150
Ω
10
90
100
0%
20ns
500mV
50mV
V
IN
V
OUT
Figure 31. Small Signal Pulse Response, Gain = +10,
(R
F
= 357
Ω
, R
L
= 150
Ω
, V
S
=
±
5 V)
1 10 1000100
FREQUENCY – MHz
–1
–6
1
0
–2
–3
–4
–5
CLOSED-LOOP GAIN (NORMALIZED) – dB
3V
5V
0
–90
–180
–270
PHASE SHIFT – Degrees
–360
PHASE
GAIN
3V 5V
V
S
= 615V G = +10
R
L
= 1kV
65V
V
S
= 615V
65V
Figure 32. Closed-Loop Gain and Phase vs. Frequency,
Gain = +10, R
L
= 1 k
Ω
216141210864 SUPPLY VOLTAGE – 6Volts
–3dB BANDWIDTH – MHz
30
20
60
40
50
70
80
90
01820
100
110
10
G = +10
R
L
= 1kV
R
F
= 154V
R
F
= 357V
R
F
= 649V
Figure 33. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, R
L
= 1 k
Ω
AD812
–11–
REV. B
10
90
100
0%
50ns
2V
V
IN
V
OUT
2V
Figure 34. Large Signal Pulse Response, Gain = –1,
(R
F
= 750
Ω
, R
L
= 150
Ω
, V
S
=
±
5 V)
–1
–6 1 10 1000100
–2
–3
–4
–5
0
1
–270
–180
–90
0
FREQUENCY – MHz
CLOSED-LOOP GAIN (NORMALIZED) – dB
G = –1
R
L
= 150V
PHASE SHIFT – Degrees
PHASE
GAIN
V
S
= 615V
65V5V
3V
V
S
= 615V
65V
5V
3V
Figure 35. Closed-Loop Gain and Phase vs. Frequency,
Gain = –1, R
L
= 150
Ω
130
30 20
60
40
2
50
0
90
70
80
100
110
120
1816141210864
–3dB BANDWIDTH – MHz
SUPPLY VOLTAGE – 6Volts
G = –1
R
L
= 150V
PEAKING # 1.0dB
R
F
= 681V
R
F
= 715V
PEAKING # 0.2dB
Figure 36. –3 dB Bandwidth vs. Supply Voltage,
Gain = –1, R
L
= 150
Ω
10
90
100
0%
500mV 20ns
500mV
V
IN
V
OUT
Figure 37. Small Signal Pulse Response, Gain = –1,
(R
F
= 750
Ω
, R
L
= 150
Ω
, V
S
=
±
5 V)
–1
–6 1 10 1000100
–2
–3
–4
–5
0
1
–270
–180
–90
0
FREQUENCY – MHz
CLOSED-LOOP GAIN (NORMALIZED) – dB
G = –10
R
L
= 1kV
65V
5V
3V
PHASE V
S
= 615V
GAIN 5V 3V
65V
V
S
= 615V
PHASE SHIFT – Degrees
Figure 38. Closed-Loop Gain and Phase vs. Frequency,
Gain = –10, R
L
= 1 k
Ω
100
020
30
10
2
20
0
60
40
50
70
80
90
1816141210864
–3dB BANDWIDTH – MHz
SUPPLY VOLTAGE – 6Volts
G = –10
R
L
= 1kV
R
F
= 357V
R
F
= 649V
R
F
= 154V
Figure 39. –3 dB Bandwidth vs. Supply Voltage,
Gain = –10, R
L
= 1 k
Ω
AD812
REV. B
–12–
General Considerations
The AD812 is a wide bandwidth, dual video amplifier which
offers a high level of performance on less than 5.5 mA per am-
plifier of quiescent supply current. It is designed to offer out-
standing performance at closed-loop inverting or noninverting
gains of one or greater.
Built on a low cost, complementary bipolar process, and achiev-
ing bandwidth in excess of 100 MHz, differential gain and phase
errors of better than 0.1% and 0.1° (into 150 Ω), and output
current greater than 40 mA, the AD812 is an exceptionally
efficient video amplifier. Using a conventional current feedback
architecture, its high performance is achieved through careful
attention to design details.
Choice of Feedback and Gain Resistors
Because it is a current feedback amplifier, the closed-loop band-
width of the AD812 depends on the value of the feedback resis-
tor. The bandwidth also depends on the supply voltage. In
addition, attenuation of the open-loop response when driving
load resistors less than about 250 Ω will affect the bandwidth.
Table I contains data showing typical bandwidths at different
supply voltages for some useful closed-loop gains when driving a
load of 150 Ω. (Bandwidths will be about 20% greater for load
resistances above a few hundred ohms.)
The choice of feedback resistor is not critical unless it is impor-
tant to maintain the widest, flattest frequency response. The
resistors recommended in the table are those (metal film values)
that will result in the widest 0.1 dB bandwidth. In those appli-
cations where the best control of the bandwidth is desired, 1%
metal film resistors are adequate. Wider bandwidths can be
attained by reducing the magnitude of the feedback resistor (at
the expense of increased peaking), while peaking can be reduced
by increasing the magnitude of the feedback resistor.
Table I. –3 dB Bandwidth vs. Closed-Loop Gain and
Feedback Resistor (R
L
= 150 Ω)
V
S
(V) Gain R
F
(⍀) BW (MHz)
±15 +1 866 145
+2 715 100
+10 357 65
–1 715 100
–10 357 60
±5 +1 750 90
+2 681 65
+10 154 45
–1 715 70
–10 154 45
+5 +1 750 60
+2 681 50
+10 154 35
–1 715 50
–10 154 35
+3 +1 750 50
+2 681 40
+10 154 30
–1 715 40
–10 154 25
To estimate the –3 dB bandwidth for closed-loop gains or feed-
back resistors not listed in the above table, the following two
pole model for the AD812 many be used:
AG
SRGrC
fS R Gr C
CL
FINT
FINT
=+
()
++
()
+
2
2
21
π
where: A
CL
= closed-loop gain
G= 1 + R
F
/R
G
r
IN
= input resistance of the inverting input
C
T
= “transcapacitance,” which forms the open-loop
dominant pole with the tranresistance
R
F
= feedback resistor
R
G
= gain resistor
f
2
= frequency of second (nondominant) pole
S= 2
π
j
f
Appropriate values for the model parameters at different supply
voltages are listed in Table II. Reasonable approximations for
these values at supply voltages not found in the table can be
obtained by a simple linear interpolation between those tabu-
lated values which “bracket” the desired condition.
Table II. Two-Pole Model Parameters at Various
Supply Voltages
V
S
r
IN
(⍀)C
T
(pF) f
2
(MHz)
±15 85 2.5 150
±5 90 3.8 125
+5 105 4.8 105
+3 115 5.5 95
As discussed in many amplifier and electronics textbooks (such
as Roberge’s Operational Amplifiers: Theory and Practice), the
–3 dB bandwidth for the 2-pole model can be obtained as:
f
3
= f
N
[1 – 2d
2
+ (2 – 4d
2
+ 4d
4
)
1/2
]
1/2
where:
ff
RGrC
N
FINT
=+
()
2
12/
and:
d = (1/2) [f
2
(R
F
+ Gr
IN
) C
T
]
1/2
This model will predict –3 dB bandwidth within about 10 to
15% of the correct value when the load is 150 Ω. However, it is
not an accurate enough to predict either the phase behavior or
the frequency response peaking of the AD812.
Printed Circuit Board Layout Guidelines
As with all wideband amplifiers, printed circuit board parasitics
can affect the overall closed-loop performance. Most important
for controlling the 0.1 dB bandwidth are stray capacitances at
the output and inverting input nodes. Increasing the space between
signal lines and ground plane will minimize the coupling. Also,
signal lines connecting the feedback and gain resistors should be
kept short enough that their associated inductance does not
cause high frequency gain errors.
AD812
–13–
REV. B
Power Supply Bypassing
Adequate power supply bypassing can be very important when
optimizing the performance of high speed circuits. Inductance
in the supply leads can (for example) contribute to resonant
circuits that produce peaking in the amplifier’s response. In
addition, if large current transients must be delivered to a load,
then large (greater than 1 µF) bypass capacitors are required to
produce the best settling time and lowest distortion. Although
0.1 µF capacitors may be adequate in some applications, more
elaborate bypassing is required in other cases.
When multiple bypass capacitors are connected in parallel, it is
important to be sure that the capacitors themselves do not form
resonant circuits. A small (say 5 Ω) resistor may be required in
series with one of the capacitors to minimize this possibility.
As discussed below, power supply bypassing can have a signifi-
cant impact on crosstalk performance.
Achieving Low Crosstalk
Measured crosstalk from the output of amplifier 2 to the input
of amplifier 1 of the AD812 is shown in Figure 40. The crosstalk
from the output of amplifier 1 to the input of amplifier 2 is a few
dB better than this due to the additional distance between criti-
cal signal nodes.
A carefully laid-out PC board should be able to achieve the level
of crosstalk shown in the figure. The most significant contribu-
tors to difficulty in achieving low crosstalk are inadequate power
supply bypassing, overlapped input and/or output signal paths,
and capacitive coupling between critical nodes.
The bypass capacitors must be connected to the ground plane at
a point close to and between the ground reference points for the
two loads. (The bypass of the negative power supply is particu-
larly important in this regard.) There are two amplifiers in the
package, and low impedance signal return paths must be pro-
vided for each load. (Using a parallel combination of 1 µF,
0.1 µF, and 0.01 µF bypass capacitors will help to achieve opti-
mal crosstalk.)
–10
–60
–110 1M 100M10M
–70
–80
–90
–100
–50
–40
–30
–20
CROSSTALK – dB
100k FREQUENCY – Hz
R
L
= 150V
Figure 40. Crosstalk vs. Frequency
The input and output signal return paths must also be kept from
overlapping. Since ground connections are not of perfectly zero
impedance, current in one ground return path can produce a
voltage drop in another ground return path if they are allowed
to overlap.
Electric field coupling external to (and across) the package can
be reduced by arranging for a narrow strip of ground plane to be
run between the pins (parallel to the pin rows). Doing this on
both sides of the board can reduce the high frequency crosstalk
by about 5 dB or 6 dB.
Driving Capacitive Loads
When used with the appropriate output series resistor, any load
capacitance can be driven without peaking or oscillation. In
most cases, less than 50 Ω is all that is needed to achieve an
extremely flat frequency response. As illustrated in Figure 44,
the AD812 can be very attractive for driving largely capacitive
loads. In this case, the AD812’s high output short circuit
current allows for a 150 V/µs slew rate when driving a 510 pF
capacitor.
AD812
8
4
R
G
R
F
V
IN
R
T
V
O
R
L
C
L
R
S
+V
S
0.1mF
1.0mF
0.1mF
1.0mF
–V
S
Figure 41. Circuit for Driving a Capacitive Load
110 1000100
FREQUENCY – MHz
6
9
3
0
–3
CLOSED-LOOP GAIN – dB
12
–6
V
S
= 65V
G = +2
R
F
= 750V
R
L
= 1kV
C
L
= 10pF
R
S
= 0
R
S
= 30V
R
S
= 50V
Figure 42. Response to a Small Load Capacitor at
±
5 V
AD812
REV. B
–14–
110 1000100
FREQUENCY – MHz
6
9
3
0
–3
CLOSED-LOOP GAIN – dB
12
–6
–9
V
S
= 615V
G = +2
R
F
= 750V
R
L
= 1kV
C
L
= 510pF, R
S
= 15V
C
L
= 150pF, R
S
= 30V
Figure 43. Response to Large Load Capacitor, V
S
=
±
15 V
10
100
0%
100ns
5V
5V
V
IN
V
OUT
90
Figure 44. Pulse Response of Circuit of Figure 41 with
C
L
= 510 pF, R
L
= 1 k
Ω
, R
F
= R
G
= 715
Ω
, R
S
= 15
Ω
Overload Recovery
There are three important overload conditions to consider.
They are due to input common mode voltage overdrive, input
current overdrive, and output voltage overdrive. When the
amplifier is configured for low closed-loop gains, and its input
common-mode voltage range is exceeded, the recovery time will
be very fast, typically under 10 ns. When configured for a higher
gain, and overloaded at the output, the recovery time will also
be short. For example, in a gain of +10, with 6 dB of input
overdrive, the recovery time of the AD812 is about 10 ns.
10
90
100
0%
2V
1V 50ns
V
IN
V
OUT
Figure 45. 6 dB Overload Recovery; G = 10, R
L
= 500
Ω
,
V
S
=
±
5 V
In the case of high gains with very high levels of input overdrive,
a longer recovery time may occur. For example, if the input
common-mode voltage range is exceeded in a gain of +10, the
recovery time will be on the order of 100 ns. This is primarily
due to current overloading of the input stage.
As noted in the warning under “Maximum Power Dissipation,”
a high level of input overdrive in a high noninverting gain circuit
can result in a large current flow in the input stage. For differ-
ential input voltages of less than about 1.25 V, this will be inter-
nally limited to less than 20 mA (decreasing with supply voltage).
For input overdrives which result in higher differential input
voltages, power dissipation in the input stage must be consid-
ered. It is recommended that external diode clamps be used in
cases where the differential input voltage is expected to exceed
1.25 V.
High Performance Video Line Driver
At a gain of +2, the AD812 makes an excellent driver for a back-
terminated 75 Ω video line. Low differential gain and phase
errors and wide 0.1 dB bandwidth can be realized over a wide
range of power supply voltage. Outstanding gain and group
delay matching are also attainable over the full operating supply
voltage range.
AD812
8
4
R
G
R
F
V
IN
75V
V
OUT
75V
+V
S
0.1mF
0.1mF
–V
S
75V
CABLE
75V
CABLE
75V
Figure 46. Gain of +2 Video Line Driver (R
F
= R
G
from
Table I)
AD812
–15–
REV. B
1 10 1000100
FREQUENCY –MHz
–1
–6
1
0
–2
–3
–4
–5
CLOSED-LOOP GAIN – dB
90
0
–90
–180
–270
PHASE SHIFT – Degrees
3V 5V
PHASE
GAIN
G = +2
R
L
= 150V
V
S
= 615V
65V
V
S
= 615V
65V
5V
3V
Figure 47. Closed-Loop Gain and Phase vs. Frequency for
the Line Driver
120
20 20
50
30
2
40
0
80
60
70
90
100
110
1816141210864 SUPPLY VOLTAGE – 6Volts
–3dB BANDWIDTH – MHz
G = +2
R
L
= 150V
R
F
= 590V
R
F
= 715V
R
F
= 750V
PEAKING # 1dB
NO PEAKING
Figure 48. –3 dB Bandwidth vs. Supply Voltage,
Gain = +2, R
L
= 150
Ω
15
0.06
0.02
6
0.04
5
0.08
14121110 13987 SUPPLY VOLTAGE – 6Volts
DIFFERENTIAL PHASE – Degrees
0.06
0.02
0.04
DIFFERENTIAL GAIN – %
0
DIFFERENTIAL GAIN
DIFFERENTIAL PHASE
Figure 49. Differential Gain and Phase vs. Supply Voltage,
Gain = +2, R
L
= 150
Ω
–0.1
–0.6 1M 100M10M
–0.2
–0.3
–0.4
–0.5
0
0.1
NORMALIZED GAIN – dB
100k FREQUENCY – Hz
0.2
0.3
0.4
G = +2
R
L
= 150V
5V
3V
V
S
= 615V
65V
Figure 50. Fine-Scale Gain Flatness vs. Frequency,
Gain = +2, R
L
= 150
Ω
1.0
0
–1.0 10 100
–0.2
–0.4
–0.6
–0.8
0.2
0.4
0.6
0.8
GAIN MATCH – dB
1FREQUENCY – MHz 1000
V
S
= 3V
R
F
= 681V
G = +2
R
L
= 150V
V
S
= 615V
R
F
= 715V
Figure 51. Closed-Loop Gain Matching vs. Frequency,
Gain = +2, R
L
= 150
Ω
0
1M 10M
–0.2
–0.4
0.2
0.4
GROUP DELAY – ns
100k FREQUENCY – Hz 100M
4
2
6
8DELAY
DELAY MATCHING
0
3V
5V
65V
615V
V
S
= 3V TO 615V
Figure 52. Group Delay and Group Delay Matching vs.
Frequency, G = +2, R
L
= 150
Ω
AD812
REV. B
–16–
Operation Using a Single Supply
The AD812 will operate with total supply voltages from 36 V
down to 2.4 V. With proper biasing (see Figure 53), it can be an
outstanding single supply video amplifier. Since the input and
output voltage ranges extend to within 1 volt of the supply rails,
it will handle a 1.3 V p-p signal on a single 3.3 V supply, or a
3 V p-p signal on a single 5 V supply. The small signal, 0.1 dB
bandwidths will exceed 10 MHz in either case, and the large
signal bandwidths will exceed 6 MHz.
The capacitively coupled cable driver in Figure 53 will achieve
outstanding differential gain and phase errors of 0.07% and 0.06
degrees respectively on a single 5 V supply. Resistor R2, in this
circuit, is selected to optimize the differential gain and phase by
operating the amplifier in its most linear region. To optimize the
circuit for a 3 V supply, a value of 8 kΩ is recommended for R2.
AD812
8
4
V
IN
R2
11.8kV
V
OUT
75V
75V
+V
S
R3
1kV
75V
CABLE
R1
9kV
C1
2mF
C3
30mF
649V649V
C
OUT
47mF
C2
1mF
Figure 53. Biasing for Single Supply Operation
8-Lead Plastic DIP
(N-8)
8
14
5
PIN 1
SEATING
PLANE
0.060 (1.52)
0.015 (0.38)
0.165 60.01
(4.19 60.25)
0.10
(2.54)
BSC
0.325 (8.25)
0.300 (7.62)
0.015 (0.381)
0.008 (0.204)
0.195 (4.95)
0.115 (2.93)
0.39 (9.91)
0.25
(6.35)
0.125 (3.18)
MIN
0.018 60.003
(0.46 +0.08) 0.033 (0.84)
NOM
8-Lead Plastic SOIC
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
85
41
0.2440 (6.20)
0.2284 (5.80)
PIN 1
0.1574 (4.00)
0.1497 (3.80)
0.0688 (1.75)
0.0532 (1.35)
SEATING
PLANE
0.0098 (0.25)
0.0040 (0.10)
0.0192 (0.49)
0.0138 (0.35)
0.0500
(1.27)
BSC 0.0098 (0.25)
0.0075 (0.19) 0.0500 (1.27)
0.0160 (0.41)
88
08
0.0196 (0.50)
0.0099 (0.25) 3 458
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
PRINTED IN U.S.A. C1859b–0–9/98
–0.5
–3.0
1 10 1000100
–1.0
–1.5
–2.0
–2.5
0
0.5
–270
–180
–90
0
FREQUENCY – MHz
CLOSED-LOOP GAIN – dB
PHASE SHIFT – Degrees
–3.5
V
S
= 5V
90
GAIN
PHASE
Figure 54. Closed-Loop Gain and Phase vs. Frequency,
Circuit of Figure 53
10
90
100
0%
500mV
1V 50ns
V
IN
V
OUT
Figure 55. Pulse Response of the Circuit of Figure 53 with
V
S
= 5 V
Mouser Electronics
Authorized Distributor
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AD812AR-EBZ