NCP1077BBP130G - ON SEMICONDUCTOR

September, 2018 − Rev. 3 1Publication Order Number:

NCP1076A/D

NCP1075A/B, NCP1076A/B,

NCP1077A/B, NCP1079A/B

Enhanced Off-line Switcher

for Robust and Highly

Efficient Power Supplies

The NCP107xuz products integrate a fixed frequency current mode

controller with a 700 V MOSFET. Available in a two different pin−out

of the very common PDIP−7 package, the NCP107xuz offers a high

level of integration, including soft−start, frequency−jittering,

short−circuit protection, skip−cycle, a maximum peak current

set−point, ramp compensation, and a dynamic self−supply (DSS,

eliminating the need for an auxiliary winding).

Unlike other monolithic solutions, the NCP107xuz is quiet by

nature: during nominal load operation, the part switches at one of the

available frequencies (65, 100 or 130 kHz). When the output power

demand diminishes, the IC automatically enters frequency foldback

mode and provides excellent efficiency at light loads. When the power

demand reduces further, it enters into a skip mode to reduce the

standby consumption down to a no load condition.

Protection features include: a timer to detect an overload or a

short−circuit event, Over−voltage Protection with auto−recovery. Ac

input line voltage detection prevents lethal runaway in low input

voltage conditions (Brown−out) as well as too high an input line (Ac

line Over−voltage Protection). This also allows an Over−power

Protection to compensate all internal delays in high input voltage

conditions and optimize the maximum output current capability.

For improved standby performance, the connection of an auxiliary

winding stops the DSS operation and helps to reduce input power

consumption below 50 mW at high line.

Features

•Built−in 700 V MOSFET with RDS(ON) of 13.5 W

(NCP1075uz), 4.8 W (NCP1076uz/77uz) and 2.9 W

(NCP1079uz)

•Large Creepage Distance Between High Voltage Pins

•Current−mode Fixed Frequency Operation –

65 / 100 / 130 kHz

•Various Options for Maximum Peak Current: see below

table

•Fixed Slope Compensation

•Skip−cycle Operation at Low Peak Currents Only

•Dynamic Self−supply: No Need for an Auxiliary

Winding

•Internal 10 ms Soft−start

•Auto−recovery Output Short−circuit Protection with

T imer−based Detection

•Auto−recovery Over−voltage Protection with Auxiliary

Winding Operation

•Adjustable Brown−out Protection and OVP

•2nd Leading Edge Blanking – Current Protection

(NCP107xuA version only)

•Over Power Protection

•Frequency Jittering for Better EMI Signature

•No Load Input Consumption < 50 mW

•Frequency Foldback to Improve Efficiency at Light

Load

•These are Pb−free Devices

Typical Applications

•Auxiliary / Standby Isolated Power Supplies

•Major Home Appliances Power Supplies

•Power Meter SMPS

•Wide Input Industrial SMPS

PDIP−7

(PDIP−8 LESS PIN 6)

CASE 626A

MARKING

DIAGRAMS

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x = Power Version (5, 6, 7, 9)

u = Pin Connections (A, B)

z = 2nd level OCP enabled/disabled (A, B)

y = Oscillator Frequency 65, 100, 130 (A, B, C

)

A = Assembly Location

WL = Wafer Lot

Y, YY = Year

W, WW = Work Week

G = Pb−Free Package

See detailed ordering and shipping information on page 31 o

this data sheet.

ORDERING INFORMATION

P107xPuzy

AWL

YYWWG

PDIP−7

(PDIP−8 LESS PIN 3)

CASE 626AS

P107xPuzy

AWL

YYWWG

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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PIN CONNECTIONS

GND

BO/AC_OVP

FB DRAIN

VCC

(Top View)

PDIP−7

NCP107xA

GND

BO/AC_OVPFB

DRAIN

VCC

(Top View)

PDIP−7

NCP107xB

PIN FUNCTION DESCRIPTION

Pin No

Pin Name Function Pin Description

PDIP 7 A PDIP 7 B

1 2 VCC IC supply pin This pin is connected to an external capacitor.

The VCC management includes an auto−recov-

ery over−voltage protection.

2 8 BO/AC_OVP Brown−out / Ac Line

Over−voltage protection Detects both input voltage conditions (Brown−

out) and too high an input voltage (Ac line OVP).

Do not leave this pin floating – if this pin is not

used it should be directly connected do GND.

3 5 GND The IC Ground

4 1 FB Feedback signal input By connecting an opto−coupler to this pin, the

peak current set−point is adjusted accordingly to

the output power demand.

5 4 DRAIN Drain connection The internal drain MOSFET connection

6 3 NC This un−connected pin ensures adequate creep-

age distance

7 6 GND The IC Ground

8 7 GND The IC Ground

PRODUCTS INFOS & INDICATIVE MAXIMUM OUTPUT POWER

Product RDS(ON) IPK

230 Vrms +15% 85−265 Vrms

Adapter Open Frame Adapter Open Frame

NCP1075uz 13.5 W400 mA 8.5 W 14 W 6 W 10 W

NCP1076uz / NCP1077uz 4.8 W800 mA 19 W 31 W 14 W 23 W

NCP1079uz 2.9 W1050 mA 25 W 41 W 18 W 30 W

NOTE: Informative values only, with Tamb = 25°C, Tcase = 100°C, PDIP−7 package, Self−supply via Auxiliary winding and circuit mounted

on minimum copper area as recommended.

QUICK SELECTION TABLE

Device Frequency [kHz] RDS(ON) [W]IPK [mA] Package type

NCP1075uz 65, 100, 130* 13.5 400

PDIP−7

(Pb−Free)

NCP1076uz 65, 100, 130* 4.8 650

NCP1077uz 65, 100, 130* 4.8 800

NCP1079uz 65, 100, 130* 2.9 1050

*NOTE: 130 kHz option available in pin connection B only

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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Figure 1. Typical Isolated Application (Flyback Converter), Enable Brown−out, Ac Line OVP and OPP Functions

Figure 2. Typical Isolated Application (Flyback Converter), Disabled Brown−out Function –

Against Line Detection

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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Figure 3. Simplified Internal Circuit Architecture

BO/AC_OVP

GND

DRAIN

VCC VCC

Management

Line Detection

RFB(UP)

TSD

LEB 1

Soft−Start

Current set−point

Ifreeze IPK(0)

Line

detection

enable

Peak current

protection

AC OVP

AC OVP ISTOP

Line

detection

enable

BO enable

Brown−out

OPP

Slope

compensation

DRV

VFB(REF)

Sawtooth

Feedback control

VCC OVP

ISTOP

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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MAXIMUM RATINGS TABLE (All voltages related to GND terminal)

Rating Symbol Value Unit

Power supply voltage, VCC pin, continuous voltage VCC −0.3 to 20 V

Voltage on all pins, except DRAIN and VCC pin Vinmax −0.3 to 10 V

DRAIN voltage BVDSS −0.3 to 700 V

Maximum Current into VCC pin ICC 15 mA

Drain Current Peak during Transformer Saturation (TJ = 150°C):

NCP1075uz

NCP1076uz/77uz

NCP1079uz

Drain Current Peak during Transformer Saturation (TJ = 25°C):

NCP1075uz

NCP1076uz/77uz

NCP1079uz

IDS(PK) 0.9

2.2

3.6

1.5

3.9

6.4

Thermal Resistance Junction−to−Air – PDIP7 0.36 Sq. Inch RθJ−A 77 °C/W

1.0 Sq. Inch 68

Maximum Junction Temperature TJMAX 150 °C

Storage Temperature Range −60 to +150 °C

Human Body Model ESD Capability (All pins except HV pin) per JEDEC JESD22−A114F HBM 2 kV

Human Body Model ESD Capability (Drain pin) per JEDEC JESD22−A114F HBM 1 kV

Charged−Device Model ESD Capability per JEDEC JESD22−C101E CDM 1 kV

Machine Model ESD Capability per JEDEC JESD22−A115−A MM 200 V

Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality

should not be assumed, damage may occur and reliability may be af fected.

1. This device contains latch−up protection and exceeds 100 mA per JEDEC Standard JESD78.

2. Maximum drain current IDS(PK) is obtained when the transformer saturates. It should not be mixed with short pulses that can be seen at turn

on. Figure 4 below provides spike limits the device can tolerate.

<1.5 x IDS(PK)

iD(t)

LEB IDS(PK)

Transformer

Saturation

Figure 4. Spike Limits

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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ELECTRICAL CHARACTERISTICS

(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, VCC = 12 V unless otherwise noted)

Symbol Rating Pin Min Typ Max Unit

SUPPLY SECTION AND VCC MANAGEMENT

VCC(ON) VCC increasing level at which the switcher starts operation 1 (2) 8.0 8.4 8.9 V

VCC(MIN) VCC decreasing level at which the HV current source restarts 1 (2) 6.5 6.9 7.3 V

VCC(OFF) VCC decreasing level at which the switcher stops operation (UVLO) 1 (2) 6.1 6.5 6.9 V

VCC(reset) VCC voltage at which the internal latch is reset (Guaranteed by design) 1 (2) 4 V

ICC1 Internal IC consumption, MOSFET switching (fSW = 65 kHz)

NCP1075uz

NCP1076uz/77uz

NCP1079uz

1 (2) −

−

1.10

1.26

1.40

−

ICC(skip) Internal IC consumption, VFB is 0 V (No switching on MOSFET) 1 (2) − 400 − mA

POWER SWITCH CIRCUIT

RDS(ON) Power Switch Circuit on−state resistance (IDRAIN = 50 mA)

NCP1075uz

TJ = 25°C

TJ = 125°C

NCP1076uz/77uz

TJ = 25°C

TJ = 125°C

NCP1079uz

TJ = 25°C

TJ = 125°C

5 (4)

−

13.5

26.0

4.8

9.3

2.9

5.3

16.8

31.6

6.8

11.6

3.9

7.5

BVDSS Power Switch Circuit & Start−up breakdown voltage

(IDRAIN(OFF) = 120 mA, TJ = 25°C) 5 (4) 700 − − V

IDSS(OFF) Power Switch & Start−up breakdown voltage off−state leakage current

TJ = 125°C (VDS = 700 V) 5 (4) − 85 − mA

Switching characteristics (RL = 50 W, VDS set for IDRAIN = 0.7 x Ilim)

T urn−on time (90% − 10%)

T urn−off time (10% − 90%)

5 (4) −

−20

10 −

−

INTERNAL START−UP CURRENT SOURCE

Istart1 High−voltage current source, VCC = VCC(ON) – 200 mV 5 (4) 4.0 9.0 12.0 mA

Istart2 High−voltage current source, VCC = 0 V 5 (4) − 0.5 − mA

VHV(MIN) Minimum start−up voltage, VCC = 0 V 5 (4) − 21 − V

VCC(TH) VCC Transient level for Istart1 to Istart2 toggling point 1 (2) − 1.6 − V

CURRENT COMPARATOR

IPK Maximum internal current set−point at 50% duty−cycle

FB pin open, TJ = 25°C

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

400

650

800

1050

−

IPK(0) Maximum internal current set−point at beginning of switching cycle

FB pin open, BO/AC_OVP pin voltage v 0.8 V, TJ = 25°C

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

420

690

850

1110

470

765

940

1230

520

840

1030

1350

3. The final switch current is: IPK(0) / (Vin/LP + Sa) x V in/LP + Vin/LP x tprop, with Sa the built−in slope compensation, Vin the input voltage, LP

the primary inductor in a flyback, and tprop the propagation delay.

4. Oscillator frequency is measured with disabled jittering.

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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ELECTRICAL CHARACTERISTICS

(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, VCC = 12 V unless otherwise noted)

Symbol UnitMaxTypMinPinRating

CURRENT COMPARATOR

IPKSW(65) Final switch current with a primary slope of 200 mA/ms,

fSW = 65 kHz (Note 3)

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

450

710

860

1100

−

IPKSW(100) Final switch current with a primary slope of 200 mA/ms,

fSW =100 kHz (Note 3)

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

440

685

825

1040

−

IPKSW(130) Final switch current with a primary slope of 200 mA/ms,

fSW =130 kHz (Note 3)

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

450

685

820

1020

−

IPK(OPP) Maximum internal current set−point at beginning of switching cycle

FB pin open, BO/AC_OVP pin voltage = 2.65 V, TJ = 25°C

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

375

610

750

985

−

tSS Soft−start duration (Guaranteed by design) − − 10 − ms

tprop Propagation delay from current detection to drain OFF state − − 100 − ns

tLEB1 Leading Edge Blanking Duration 1 − − 300 − ns

tLEB2 Leading Edge Blanking Duration 2 (NCP107xuA version only) − − 100 − ns

INTERNAL OSCILLATOR

fOSC(65) Oscillation frequency, 65 kHz version, TJ = 25°C (Note 4) − 59 65 71 kHz

fOSC(100) Oscillation frequency, 100 kHz version, TJ = 25°C (Note 4) − 90 100 110 kHz

fOSC(130) Oscillation frequency, 130 kHz version, TJ = 25°C (Note 4) −117 130 143 kHz

fjitter Frequency jittering in percentage of fOSC − − ±6 − %

fswing Jittering modulation frequency − − 300 − Hz

DMAX Maximum duty−cycle − 64 68 72 %

FEEDBACK SECTION

IFB(fault) FB current for which Fault is detected 4 (1) −−35 −mA

IFB100% FB current for which internal current set−point is 100% (IPK(0))4 (1) −−44 −mA

IFB(freeze) FB current for which internal current set-point is Ifreeze 4 (1) −−90 −mA

VFB(REF) Equivalent pull−up voltage in linear regulation range

(Guaranteed by design) 4 (1) − 3.3 − V

RFB(UP) Equivalent feedback resistor in linear regulation range

(Guaranteed by design) 4 (1) − 19.5 − kΩ

FREQUENCY FOLDBACK & SKIP

IFBfold Start of frequency foldback FB pin current level 4 (1) −−68 −mA

IFBfold(END) End of frequency foldback FB pin current level, fSW = fMIN 4 (1) −−100 −mA

3. The final switch current is: IPK(0) / (Vin/LP + Sa) x V in/LP + Vin/LP x tprop, with Sa the built−in slope compensation, Vin the input voltage, LP

the primary inductor in a flyback, and tprop the propagation delay.

4. Oscillator frequency is measured with disabled jittering.

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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ELECTRICAL CHARACTERISTICS

(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, VCC = 12 V unless otherwise noted)

Symbol UnitMaxTypMinPinRating

FREQUENCY FOLDBACK & SKIP

fMIN The frequency below which skip−cycle occurs, TJ = 25°C (Note 4) − 23 27 31 kHz

IFB(skip) The FB pin current level to enter skip mode 4 (1) −−120 −mA

Ifreeze Internal minimum current set−point (IFB = IFB(freeze))

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

165

270

330

430

−

SLOPE COMPENSATION

Sa(65) The internal slope compensation @ 65 kHz:

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

mA/ms

Sa(100) The internal slope compensation @ 100 kHz:

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

mA/ms

Sa(130) The internal slope compensation @ 130 kHz:

NCP1075uz

NCP1076uz

NCP1077uz

NCP1079uz

−

mA/ms

PROTECTIONS

tSCP Fault validation further to error flag assertion − 35 48 − ms

trecovery OFF phase in fault mode − − 420 − ms

VOVP VCC voltage at which the switcher stops pulsing 1 (5) 17.0 18.0 18.8 V

tOVP The filter of VCC OVP comparator − − 80 − ms

VBO(EN) Brown−out level detection 2 (8) − 50 − mV

VBO(ON) Brown−out level, the switcher starts pulsing, OPP starts to decrease IPK 2 (8) 0.76 0.80 0.84 V

VBO(HYST) Brown−out hysteresis (Guaranteed by design) 2 (8) − 100 − mV

VACOVP(ON) OVP level when the switcher stops pulsing 2 (8) 2.755 2.900 3.045 V

VACOVP(OFF) OVP level when the switcher starts pulsing 2 (8) 2.3 2.6 2.9 V

tBOfilter VBO filter − − 20 − ms

tBO Brown−out timer − − 50 − ms

VHV(EN) The drain pin voltage above which the MOSFET operates. Checked after one

of the following events: TSD, UVLO, SCP, or VCC OVP mode, BO/AC_OVP

pin = 0 V

5 (4) 72 91 110 V

IPK(150) High current protection, percent of max limit IPK (NCP107xuA version only) − − 150 − %

TEMPERATURE MANAGEMENT

TSD Temperature shutdown (Guaranteed by design) − 150 − − °C

TSDHYST Hysteresis in shutdown (Guaranteed by design) − − 20 − °C

3. The final switch current is: IPK(0) / (Vin/LP + Sa) x V in/LP + Vin/LP x tprop, with Sa the built−in slope compensation, Vin the input voltage, LP

the primary inductor in a flyback, and tprop the propagation delay.

4. Oscillator frequency is measured with disabled jittering.

Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product

performance may not be indicated by the Electrical Characteristics if operated under different conditions.

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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TYPICAL CHARACTERISTICS

Figure 5. VCC(on) vs. Temperature Figure 6. VCC(min) vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

806040 100200−20−40

8.25

8.30

8.35

8.40

8.45

8.50

100806040200−20−40

6.80

6.82

6.84

6.86

6.88

6.90

6.92

6.98

Figure 7. VCC(off) vs. Temperature Figure 8. IDSS(off) vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

806040 120200−20−40

6.42

6.45

6.48

6.46

6.47

6.49

8060 12040200−20−40

110

120

130

Figure 9. ICC1(1075uz) vs. Temperature Figure 10. ICC1(1076uz/77uz) vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

806040 100200−20−40

1.00

1.02

1.08

1.06

80 1006040200−20−40

1.18

1.20

1.22

1.24

VCC(on) (V)

VCC(min) (V)

VCC(off) (V)

IDSS(off) (mA)

ICC1(1075uz) (mA)

ICC1(1076uz/77uz) (mA)

120 120

100 100

120 120

6.78

6.94

6.96

6.43

6.44

100

1.04

1.10

1.12

1.14

1.16

1.26

1.28

1.19

1.21

1.23

1.25

1.27

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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TYPICAL CHARACTERISTICS

Figure 11. ICC1(1079uz) vs. Temperature Figure 12. IPK(0)1075uz vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

8060 10040200−20−40

1.21

1.27

1.29

1.31

1.33

1.39

100806040200−20−40

420

430

450

460

Figure 13. IPK(0)1076uz vs. Temperature Figure 14. IPK(0)1077uz vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

8060 10040200−20−40

680

700

740

780

100806040200−20−40

840

860

880

900

920

960

ICC1(1079uz) (mA)

IPK(0)1075uz (mA)

IPK(0)1076uz (mA)

IPK(0)1077uz (mA)

Figure 15. IPK(0)1079uz vs. Temperature Figure 16. ISTART1 vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

100806040200−20−40

1000

1040

1080

1120

1160

1200

806040 100200−20−40

IPK(0)1079uz (mA)

ISTART1 (mA)

440

120 120

120

720

760

120

120 120

940

1.35

1.37

1.25

1.23

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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TYPICAL CHARACTERISTICS

Figure 17. ISTART2 vs. Temperature Figure 18. RDS(on) vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

8060 10040200−20−40

0.25

0.30

0.50

0.45

0.35

0.40

0.65

100806040200−20−40

Figure 19. fOSC65 vs. Temperature Figure 20. fOSC100 vs. Temperature

TEMPERATURE (°C) TEMPERATURE (°C)

8060 12040200−20−40

100806040200−20−40

ISTART2 (mA)

RDS(on) (W)

fOSC65 (kHz)

fOSC100 (kHz)

Figure 21. fOSC130 vs. Temperature

TEMPERATURE (°C)

120806040200−20−40

119

121

125

131

fOSC130 (kHz)

120 120

NCP1075uz

NCP1079uz

NCP1076uz/77uz

120

100

0.60

0.55

61 93

127

Figure 22. DMAX vs. Temperature

TEMPERATURE (°C)

120806040200−20−40

67.1

67.2

67.3

67.5

DMAX (%)

100

67.4

123

129

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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TYPICAL CHARACTERISTICS

TEMPERATURE (°C)

8060 10040200−20−40

350

355

360

365

370

375

380

100806040200−20−40

TEMPERATURE (°C)

8060 10040200−20−40

17.9

18.0

18.2

18.4

100806040200−20−40

tRECOVERY (ms)

tSCP (ms)

VOVP (V)

VHV(en) (V)

TEMPERATURE (°C)

120806040200−20−40

0.785

0.790

0.795

0.800

0.805

0.810

VBO(on) (V)

120

18.1

18.3

100

Figure 23. fMIN vs. Temperature

TEMPERATURE (°C)

806040 100200−20−40

26.0

26.5

27.0

27.5

28.0

fMIN (kHz)

120

Figure 24. tRECOVERY vs. Temperature

Figure 25. tSCP vs. Temperature Figure 26. VOVP vs. Temperature

Figure 27. VHV(en) vs. Temperature Figure 28. VBO(on) vs. Temperature

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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TYPICAL CHARACTERISTICS

Figure 29. VACOVP(on) vs. Temperature

TEMPERATURE (°C)

806040 120200−20−40

2.85

2.89

2.91

2.95

2.97

2.99

VACOVP(on) (V)

100

2.93

2.87

Figure 30. VACOVP(off) vs. Temperature

TEMPERATURE (°C)

8060 10040200−20−40

2.590

2.595

2.600

2.610

2.615

2.620

VACOVP(off) (V)

120

2.605

Figure 31. BVDSS/BVDSS(255C) vs.

Temperature

TEMPERATURE (°C)

120806040200−20−40

0.925

0.950

1.025

1.100

BVDSS/BVDSS(25°C) [−]

1.000

100

0.975

1.050

1.075

Figure 32. Drain Current Peak during

Transformer Saturation vs. Junction

Temperature

TEMPERATURE (°C)

8060 12040200−20−40

IDS(pk) (A)

NCP1075uz

NCP1079uz

NCP1076uz/77uz

100

140

Figure 33. ICC1 vs. VCC

VCC (V)

1413 171211987

1.0

1.1

1.4

1.7

ICC1 (mA)

NCP1075uz

NCP1079uz

NCP1076uz/77uz

1.3

1.5

10 16

1.2

1.6

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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APPLICATION INFORMATION

Introduction

Thanks to ON Semiconductor Very High Voltage

Integrated Circuit technology, the circuit hosts a

high−voltage power MOSFET featuring a 13.5/4.8/2.9 W

RDS(ON) – TJ = 25°C. A n internal current source delivers the

start−up current, necessary to crank the power supply.

•Current−mode operation: The controller uses

current−mode control architecture.

•700 V Power MOSFET: Thanks to ON Semiconductor

Very High Voltage Integrated Circuit technology, the

circuit hosts a high−voltage power MOSFET featuring

a 4.8 and 2.9 W RDS(ON) – TJ = 25°C. This value lets

the designer build a power supply up to 28 W operated

on universal mains. An internal current source delivers

the start−up current, necessary to crank the power

supply.

•Dynamic Self−Supply: This device could be used in an

application without an auxiliary winding to provide

supply voltage via an internal high−voltage current

source.

•Short−circuit protection: By permanently monitoring

the feedback line activity, the IC is able to detect the

presence of a short−circuit, immediately reducing the

output power for a total system protection. A tSCP timer

is started as soon as the feedback current is below

threshold, IFB(fault), which indicates a maximum peak

current condition. If at the end of this timer the fault is

still present, then the device enters a safe,

auto−recovery burst mode, affected by a fixed timer

recurrence, trecovery. Once the short has disappeared, the

controller resumes and goes back to normal operation.

•Built−in VCC Over−Voltage Protection: When the

auxiliary winding is used to bias the VCC pin (no

DSS), an internal comparator is connected to VCC pin.

In case the voltage on the pin exceeds the VOVP level

(18 V typically), the controller immediately stops

switching and awaits a full timer period (trecovery)

before attempting to re−start. If the fault is gone, the

controller resumes operation. If the fault is still there,

e.g. in the case of a broken opto−coupler, the controller

protects the load through a safe burst mode.

•Line detection: An internal comparator monitors the

drain voltage. If the drain voltage is lower than the

internal threshold (VHV(EN)), the internal power switch

is inhibited. This avoids operating at too low an ac

input. Line detection is active, when BO/AC_OVP pin

is grounded.

•Brown−out detection and AC line Over−Voltage

Protection: The BO/AC_OVP input monitors bulk

voltage level via resistive divider and thus assures that

the application is working only for designed bulk

voltage. When BO/AC_OVP pin is connected to

ground, Line detection is inhibited.

•Internal OPP: An internal function using the bulk

voltage to program the maximum current reduction for

a given input voltage. Internal OPP is active when

BO/AC_OVP pin is connected via resistive divider to

the bulk voltage.

•2nd LEB (NCP107xuA only): Second level of current

protection. If peak current is 150% max peak current

limit, then the controller stops switching after three

pulses and waits for an auto−recovery period (trecovery)

before attempting to re−start.

•Frequency jittering: An internal low−frequency

modulation signal varies the pace at which the

oscillator frequency is modulated. This helps spreading

out energy in conducted noise analysis. To improve the

EMI signature at low power levels, the jittering remains

active in frequency foldback mode.

•Soft−Start: A 10 ms soft−start ensures a smooth

start−up sequence, reducing output overshoots.

•Frequency foldback capability: A continuous flow of

pulses is not compatible with no−load/light−load

standby power requirements. To excel in this domain,

the controller observes the feedback current

information and when it reaches a level of IFBfold, the

oscillator then starts to reduce its switching frequency

as the feedback current continues to increase (the power

demand continues to reduce). It can go down to 27 kHz

(typical) reached for a feedback level of IFBfold(END)

(100 mA roughly). At this point, if the power continues

to drop, the controller enters classical skip−cycle mode.

•Skip: If SMPS naturally exhibits a good efficiency at

nominal load, they begin to be less efficient when the

output power demand diminishes. By skipping

un−needed switching cycles, the NCP107xuz

drastically reduces the power wasted during light load

conditions.

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Start−up Sequence

When the power supply is first powered from the mains

outlet, the internal current source (typically 9.2 mA) is

biased and charges up the VCC capacitor from the drain pin.

Once the voltage on this VCC capacitor reaches the VCC(ON)

level (typically 8.4 V), the current source turns off and

pulses are delivered by the output stage: the circuit is awake

and activates the power MOSFET if the bulk voltage is

above VHV(EN) level (Brown−in protection) or voltage on

BO/AC_OVP pin is above VBO(ON) level (Brown−out

protection). Figure 34 details the simplified internal

circuitry.

Being loaded by the circuit consumption, the voltage on

the V CC capacitor goes down. When VCC is below VCC(MIN)

level (7 V typically), it activates the internal current source

to bring VCC toward VCC(ON) level and stops again: a cycle

takes place whose low frequency depends on the VCC

capacitor and the IC consumption. A 1.5 V ripple takes place

on the VCC pin whose average value equals (VCC(ON) +

VCC(MIN))/2. Figure 35 portrays a typical operation of the

DSS.

Rlimit

DRAIN

Istart 1

GND

VCC (ON )

VCC (MIN )

VOVP

CVCC

VCC

VBULK

ICC1

Figure 34. The Internal Arrangement of the

Start−up Circuitry

Figure 35. The Charge / Discharge Cycle Over a 1 mF VCC Capacitor

012345678

V [V]

time [ms]

VCC

8.4 V

VCC(TH)

Startup Duration

Device

Internal

Pulses

6.9 V

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As one can see, even if there is auxiliary winding to

provide energy for VCC, it happens that the device is still

biased by DSS during start−up time or some fault mode

when the voltage on auxiliary winding is not ready yet. The

VCC capacitor shall be dimensioned to avoid VCC crosses

VCC(OFF) level, which stops operation. The ΔV between

VCC(MIN) and VCC(OFF) is 0.5 V. There is no current source

to charge VCC capacitor when driver is on, i.e. drain voltage

is close to zero. Hence the VCC capacitor can be calculated

using

CVCC wICC1 @DMAX

fOSC @DV(eq. 1)

Take the 65 kHz device as an example. CVCC should be

above

CVCC +1.45 @10−3 @0.73

59 @103@0.5 +36 nF

A mar gin that covers the temperature drift and the voltage

drop due to switching inside FET should be considered, an d

thus a capacitor above 0.1 mF is appropriate.

The VCC capacitor has only a supply role and its value

does not impact other parameters such as fault duration or

the frequency sweep period for instance. As one can see on

Figure 34, a n internal OVP comparator protects the switcher

against lethal VCC runaways. This situation can occur if the

feedback loop opto−coupler fails, for instance, and you

would like to protect the converter against an over−voltage

event. In that case, the over−voltage protection (OVP)

circuit immediately stops the output pulses for trecovery

duration (420 ms typically). Then a new start−up attempt

takes place to check whether the fault has disappeared or not.

The OVP paragraph gives more design details on this

particular section.

Fault Condition – Short−circuit on VCC

In some fault situations, a short−circuit can purposely

occur between VCC and GND. In high line conditions

(VHV = 370 V dc) the current delivered by the start−up

device will seriously increase the junction temperature. For

instance, since Istart1 equals 4.9 mA (the min corresponds to

the highest TJ), the device would dissipate

370 x 4.9 x 10−3 = 1.81 W. To avoid this situation, the

controller includes a novel circuitry made of two start−up

levels, Istart1 and Istart2. At power−up, as long as VCC is

below a 1.6 V level, the source delivers Istart2 (around

500 mA typical), then, when VCC reaches 1.6 V, the source

smoothly transitions to Istart1 and delivers its nominal value.

As a result, in case of short−circuit between VCC and GND,

the power dissipation will drop to 370 x 500 x 10−6 =

185 mW. Figure 35 portrays this particular behavior.

The first start−up period is calculated by the formula

C x V = I x t, which implies a 1 x 10−6 x 1.6

/ (500 x 10−6) = 3.2 ms start−up time for the first sequence.

The second sequence is obtained by toggling the source to

8.9 mA with a ΔV of VCC(ON) − VCC(TH) =

8.4 V – 1.6 V = 6.8 V, which finally leads to a second

start−up time of 1 x 10−6 x 6.8 / (8.9 x 10−3) = 0.76 ms.

The total start−up time becomes 3.2 ms + 0.76 ms =

3.96 ms. Please note that this calculation is approximated by

the presence of the knee in the vicinity of the transition.

Fault Condition – Output Short−circuit

As soon as VCC reaches VCC(ON), drive pulses are

internally enabled. If everything is correct, the auxiliary

winding increases the voltage on the VCC pin as the output

voltage rises. During the start−sequence, the controller

smoothly ramps up the peak drain current to maximum

setting, i.e. IPK, which is reached after a typical period of

10 ms. When the output voltage is not regulated, the current

coming through FB pin is below IFBfault level (35 mA

typically), which is not only during the start−up period but

also anytime an overload occurs, an internal error flag is

asserted, IpFlag, indicating that the system has reached its

maximum current limit set−point. The assertion of this flag

triggers a fault counter tSCP (48 ms typically). If at counter

completion, IpFlag remains asserted, all driving pulses are

stopped and the part stays off in trecovery duration (about

420 ms). A new attempt to re−start occurs and will last

48 ms providing the fault is still present. If the fault still

affects the output, a safe burst mode is entered, affected by

a low duty−cycle operation (11%). When the fault

disappears, the power supply quickly resumes operation.

Figure 36 depicts this particular mode:

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Figure 36. In Case of Short−circuit or Overload, the NCP107xuz Protects Itself and the Power Supply Via a Low

Frequency Burst Mode. The V

is Maintained by the Current Source and Self−supplies the Controller.

DRV

internal

420 ms typ.

Fault level 48 ms typ.

Timer

VCC

VCC(MIN)

VCC(ON)

VFB

IpFlag Open loop FB

Auto−recovery Over−voltage Protection

The particular NCP107xuz arrangement offers a simple

way to prevent output voltage runaway when the

opto−coupler fails. As Figure 37 shows, a comparator

monitors the VCC pin. If the auxiliary winding delivers too

much voltage to the CVCC capacitor, then the controller

considers an OVP situation and stops the internal drivers.

When an OVP occurs, all switching pulses are permanently

disabled. After trecovery delay, the circuit resumes

operations. If the failure symptom still exists, e.g. feedback

opto−coupler fails, the device keeps the auto−recovery OVP

mode. We recommend the insertion of a resistor (Rlimit)

between the auxiliary dc level and the VCC pin to protect the

IC against high voltage spikes, which can damage the IC. It

is also recommended to filter out the VCC line to avoid

undesired OVP activations. Rlimit should be carefully

selected to suppress false−triggers of the OVP as we

discussed, but also to avoid disturbing the VCC in low / light

load conditions.

Self−supplying controllers in extremely low−standby

applications of t e n p u z z l e s t h e designe r. Actually, if a SMPS

operated at nominal load can deliver an auxiliary voltage of

an arbitrary 16 V (Vnom), this voltage can drop below 10 V

(Vstby) when entering standby. This is because the

recurrence of the switching pulses expands so much that the

low frequency re−fueling rate of the VCC capacitor is not

enough to keep a proper auxiliary voltage.

VOVP GND

VCC

DRAIN

Shut down

Internal DRV

80 ms

filter

VCC(ON) =8.4V

VCC (MIN ) =6.9V Istart 1

Rlimit D1

CVCC CAUX NAUX

Figure 37. A More Detailed View of the NCP107xuz Offers

Better Insight on How to Properly Wire an Auxiliary Winding

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Figure 38. Describes the Main Signal Variations When the Part Operates in Auto−recovery OVP

VCC

IFB

Timer

DRV

internal

VCC(MIN)

VCC(ON)

VOVP

Fault level 48 ms typ.

420 ms typ.

Soft−start

The NCP107xuz features a 10 ms soft−start which

reduces the power−on stress but also contributes to lower

the output overshoot. Soft−start is running every time when

IC starts switching. It means a first start, a new start after

OVP, TSD, Brown−out, etc. Figure 39 shows a typical

operating waveform. The NCP107xuz features a novel

patented structure which offers a better soft−start ramp,

almost ignoring the start−up pedestal inherent to traditional

current−mode supplies:

DRAIN current

VCC VCC(ON)

Max IPK

10 ms

0V (fresh PON)

Figure 39. The 10 ms Soft−start Sequence

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Jittering

Frequency jittering is a method used to soften the EMI

signature by spreading the energy in the vicinity of the main

switching component. The NCP107xuz offers a ±6%

deviation o f the nominal switching frequency. The sweeping

sawtooth is internally generated and modulates the clock up

and down with a fixed frequency of 300 Hz. Figure 40 shows

the relationship between the jitter ramp and the frequency

deviation. It is not possible to externally disable the jitter.

65 kHz

68.9 kHz

61.1 kHz

Jitter ramp

Internal

sawtooth

adjustable

Figure 40. Modulation Effects on the Clock Signal by the Jittering Sawtooth

Line Detection

When BO/AC_OVP pin is grounded (voltage on this pin

is below VBO(EN)) Figure 2, then an internal comparator

monitors the drain voltage as recovering from one of the

following situations:

•Short−Circuit Protection,

•VCC OVP is Confirmed,

•UVLO

•TSD

If the drain voltage is lower than the internal threshold

VHV(EN) (91 V dc typically), the internal power switch is

inhibited. This avoids operating at too low ac input.

Brown−out Function, Ac Line Over−voltage Protection

The Brown−out circuitry offers a way to protect the

application from operation under too low an input voltage.

Below a given level, the controller blocks the output pulses,

above it, it authorizes them. The internal circuitry, depicted

by Figure 41, of fers a way to observe the high−voltage (HV)

rail.

Figure 41. The Internal Brown−out Configuration

BO/AC_OVP

VBO(ON)

VBO(EN)

Line

detection

disable

BO enable

VAC(OVP)

AC OVP

20μs

filter

20μs

filter

20μs

filter

RUPPER

RLOWER

VBULK

CBO

tBO

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A resistive divider made of RUPPER and RLOWER, brings

a portion of the HV rail on BO/AC_OVP pin. Below the

VBO(EN) = 50 mV is the Brown−out function disabled, over

the VBO(EN) Brown−out function is enable and against Line

detection is inhibited. If voltage on BO/AC_OVP pin is

higher than VBO(ON), switcher starts pulsing. If voltage falls

down under VBO(OFF) − level VBO(ON) minus VBO(HYST),

the switcher waits 50 ms and then stops pulsing, depicted by

Figure 42. Bulk voltage at which IC starts switching is set by

resistive divider.

Figure 42. Brown−out Input Functionality with 50 ms Timer

VBO(OFF)

VBO(ON)

VCC

VBO/AC_OVP

DRV

internal

VCC(MIN)

VCC(ON)

Timer

50 ms

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The IC also includes over−voltage protection. If the voltage on BO/AC_OVP pin exceed VACOVP(ON), the switcher

immediately stops pulsing until the voltage on BO/AC_OVP pin drops under VACOVP(OFF), depicted by Figure 43.

AVOVP(OFF)

VACOVP(ON)

VBO(OFF)

VBO(ON)

VCC

BO/AC_OVP

DRV

internal

VCC(MIN)

VCC(ON)

Figure 43. Brown−out Input Functionality with Ac Line OVP Function

Calculation of the resistive divider:

RLOWER

RUPPER +VBO(ON)

VBULK *VBO(ON) (eq. 2)

If we decide to start pulsing at VBULK(ON) = 113 V dc (80 V rms at ac mains):

RLOWER

RUPPER +VBO(ON)

VBULK(ON) *VBO(ON) +0.8

113 *0.8 [7.1 m

We choose RLOWER = 100 kW

RUPPER +100 @103

7.1 @10−3 +14 MW

Then power losses on resistive divider for worst case (VBULK = 409 V dc)

P+U@I+U2

R+U2

RUPPER )RLOWER +4092

14 @106)100 @103+12 mW (eq. 3)

For VBULK(ON) = 113 V dc will be over−voltage protection (voltage when the switcher stops pulsing):

VBULK(OVP) +VACOVP(ON) @RLOWER )RUPPER

RLOWER +VACOVP(ON) @VBULK(ON)

VBO(ON) +29 @113

0.8 +409 Vdc +290 Vrms (eq. 4)

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VBO(OFF)

VBO(ON)

VCC

O/AC_OVP

Drain

current

VCC(MIN)

VCC(ON)

Timer

50 ms

Soft−start Soft−start

Figure 44. Brown−out Functionality in Soft−start

If voltage on VCC pin is higher than VCC(ON) and voltage

on BO/AC_OVP pin is higher than VBO(ON) then IC starts

pulsing, drain current is increasing for 10 ms (Soft−start).

Brown−out is inhibited during Soft−start, when Soft−start

ended, Brown−out checked if is voltage on BO/AC_OVP

pin higher than VBO(OFF). If the voltage is lower , timer count

50 ms and if the voltage don’t increase over VBO(OFF) then

IC stops switching as one can see on Figure 44.

Frequency Foldback

The reduction of no−load standby power associated with

the need for improving the efficiency, requires to change the

traditional fixed−frequency type of operation. This device

implements a switching frequency folback when the

feedback current passes above a certain level, IFBfold, set

around 68 mA. At this point, the oscillator enters frequency

foldback and reduces its switching frequency.

The internal peak current set−point is following the

feedback current information until its level reaches the

minimal freezing level point of Ifreeze. Below this value, the

peak current set−point is frozen to 30% of the IPK(0). The

only way to further reduce the transmitted power is to

diminish the operating frequency down to fMIN (27 kHz

typically). This value is reached at a feedback current level

of IFBfold(END) (100 mA typically). Below this point, if the

output power continues to decrease, the part enters skip

cycle for the best noise−free performance in no−load

conditions. Figures 45 and 46 depict the adopted scheme for

the part.

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Figure 45. By Observing the Current on the FB pin, the Controller Reduces its

Switching Frequency for an Improved Performance at Light Load

100

120

140

50 60 70 80 90 100

Frequency [kHz]

IFB [mA]

130 kHz

100 kHz

65 kHz

Figure 46. IPK Set−point is Frozen at Lower Power Demand

200

400

600

800

1000

1200

1400

40 50 60 70 80 90 100 110

Current set point [mA]

IFB [mA]

NCP1079uz

NCP1077uz

NCP1076uz

NCP1075uz

Feedback and Skip

The FB pin operates linearly as the absolute value of

feedback current (IFB) is above 40 mA. In this linear

operating range, the dynamic resistance is 19.5 kW typically

(RFB(UP)) and the effective pull up voltage is 3.3 V typically

(VFB(REF)). When IFB is decreased, the FB voltage will

increase to 3.3 V.

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Figure 47 depicts the skip mode block diagram. When the

FB current information reaches IFB(skip), the internal clock

to set the flip−flop is blanked and the internal consumption

of the controller is decreased. The hysteresis of internal skip

comparator is minimized to lower the ripple of the auxiliary

voltage for VCC pin and VOUT of power supply during skip

mode. It easies the design of VCC overload range.

OSC

Jittering

Foldback

IFB(skip)

RFB (UP )

VFB(REF) S

CS comparator

SKIP

DRV stage

Figure 47. Skip Cycle Schematic

Over−power Protection

This function lets you limit the maximum dc output current regardless of the operating input voltage. For a correct operation,

the BO/AC_OVP pin must be connected via a resistive divider to observe the bulk voltage.

IFB to CS setpoint

Ifreeze IPK(0)

Vramp + Vsense

OSC

IFB

MOSFET

RUPPER

RLOWER

VBULK

BO/AC_OVP

2.65 V

VBO (ON )

IPK (0 )

IPK (OPP )

Figure 48. The OPP Circuity Affects the Maximum Peak

Current Set−point in Relationship to the Input Voltage.

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Figure 49. Current Set−point Dependence on BO/AC_OVP Pin Voltage

300

400

500

600

700

800

900

1000

1100

1200

1300

0 0.5 1.0 1.5 2.0 2.5 3.0

Max current set-point [mA]

VBO/ACOVP [V]

NCP1079uz

NCP1077uz

NCP1076uz

NCP1075uz

There are several known ways to implement Over−power

Protection (OPP), all suffering from particular problems.

These problems range from the added consumption burden

on the converter or the skip−cycle disturbance brought by

the current−sense offset. In this case is added consumption

due to resistive divider (Equation 2).

Maximum peak current is reduced internally according t o

bulk voltage. When VBO(OPP) is maximum, the peak current

set−point is reduced by 10%. Bulk voltage at which will be

maximum current peak reduced by 20% (10% in

NCP1075uz):

VBULK(OPP) +VBO(OPP) @VBULK(ON)

VBO(ON) +VBO(OPP) @RLOWER )RUPPER

RLOWER +2.65 @100 @103)14 @106

100 @103+375 Vdc +265 Vrms

(eq. 5)

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Second LEB – Peak Current Protection (NCP107xuA

only)

There is a second level of current protection with 100 ns

propagation delay to prevent IC against high peak current.

If peak current is 150% max peak current limit, then the

controller stops switching after three pulses and waits for an

auto−recovery period (trecovery) before attempting to

re−start.

Slope Compensation and IPK Set−point

In order to let the NCP107xuz operate in CCM with a

duty−cycle above 50%, a fixed slope compensation is

internally applied to the current−mode control.

Below appears a table of the slope compensation level, the

initial current set−point, and the final current set−point of

different versions of switcher.

NCP1075uz NCP1076uz NCP1077uz NCP1079uz

fSW [kHz] 65 100 130 65 100 130 65 100 130 65 100 130

Sa [mA/µs] 9 14 18 15 23 30 18 28 36 24 37 46

IPK (Duty−cycle = 50%) [mA] 400 600 800 1050

IPK(0) [mA] 470 765 940 1230

Figure 50 depicts the variation of IPK set−point vs. the power switcher duty ratio, which is caused by the internal ramp

compensation.

Figure 50. IPK Set−point varies with Power Switch On Time, which is Caused by the Ramp Compensation

200

400

600

800

1000

1200

1400

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

IIPK set-point [mA]

Duty Ratio [%]

NCP1079uz

NCP1077uz

NCP1076uz

NCP1075uz

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Design Procedure

The design of an SMPS around a monolithic device does

not differ from that of a standard circuit using a controller

and a MOSFET. However, one needs to be aware of certain

characteristics specific of monolithic devices. Let us follow

the steps:

VIN,MIN = 90 V rms or 127 V dc once rectified,

assuming a low bulk ripple

VIN,MAX = 265 V rms or 375 V dc

VOUT = 12 V

POUT = 10 W

Operating mode is CCM

η = 0.8

1. The lateral MOSFET body−diode shall never be

forward biased, either during start−up (because of

a large leakage inductance) or in normal operation,

depicted by Figure 51. This condition sets the

maximum voltage that can be reflected during tF

As a result, the flyback voltage which is reflected

on the drain at the switch opening cannot be larger

than the input voltage. When selecting

components, you thus must adopt a turn ratio

which adheres to the following equation:

N@ǒVOUT )VFǓtVIN,MIN (eq. 6)

2. In our case, since we operate from a 127 V dc rail

while delivering 12 V, we can select a reflected

voltage of 120 V dc maximum. Therefore, the turn

ratio Np:Ns must be smaller than

Vreflect

VOUT )VF+120

12 )0.5 +9.6orNp:Nst9.6

Here we choose N = 8 in this case. We will see later

on how it affects the calculation.

Figure 51. The Drain−Source Wave Shall Always be Positive

IPEAK

IVALLEY

Iavg

ILavg

DTsw

Tsw

DIL

Figure 52. Primary Inductance Current

Evolution in CCM

3. Lateral MOSFETs have a poorly doped

body−diode which naturally limits their ability to

sustain the avalanche. A traditional RCD clamping

network shall thus be installed to protect the

MOSFET. In some low power applications,

a simple capacitor can also be used since

VDRAIN,MAX +(eq. 7)

VIN )N@ǒVOUT )VFǓ)IPEAK @LF

CTOT

where LF is the leakage inductance, CTOT the total

capacitance at the drain node (which is increased by

the capacitor you will wire between drain and

source), N the NP:NS turn ratio, VOUT the output

voltage, VF the secondary diode forward drop and

finally, IPEAK the maximum peak current. Worse

case occurs when the SMPS is very close to

regulation, e.g. the VOUT target is almost reached

and IPEAK is still pushed to the maximum. For this

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design, we have selected our maximum voltage

around 650 V (at VIN = 375 V dc). This voltage is

given by the RCD clamp installed from the drain to

the bulk voltage. We will see how to calculate it later

on.

4. Calculate the maximum operating duty−cycle for

this flyback converter operated in CCM:

DMAX +N@ǒVOUT )VFǓ

N@ǒVOUT )VFǓ)VIN,MIN +(eq. 8)

1)VIN,MIN

N@(VOUT)VF)

+0.44

5. To obtain the primary inductance, we have the

choice between two equations:

L+ǒVIN @DǓ2

fSW @K@PIN (eq. 9)

K+DIL

ILavg (eq. 10)

where

and defines the amount of ripple we want in CCM,

depicted by Figure 51.

•Small K: deep CCM, implying a large primary

inductance, a low bandwidth and a large leakage

inductance.

•Large K: approaching DCM where the conduction

losses are worse, but smaller inductance, leading to a

better leakage inductance.

From Equation 9, a K factor of 1 (50% ripple), gives an inductance of:

DIL+VIN @D

L@fSW +127 @0.44

3.8 @10−3 @65 @103+223 mA (eq. 11)

L+(127 @0.44)2

65 k @1@12.75 +3.8 mH

peak−to−peak

The peak current can be evaluated to be:

IPEAK +Iavg

D)DIL

2+98 @10−3

0.44 )223 @10−3

2+335 mA (eq. 12)

On IL, ILavg can also be calculated

ILavg +IPEAK *DIL

2+335 @10−3 *223 @10−3

2+223 mA (eq. 13)

6. Based on the above numbers, we can now evaluate the conduction losses:

ID,RMS +DǒIPEAK2*IPEAK @DIL)DIL2

3Ǔ

Ǹ+0.44ǒ0.3352*0.335 @0.223 )0.2232

3Ǔ

Ǹ+154 mA (eq. 14)

If we take the maximum RDS(ON) for a 125°C junction temperature, i.e. 10.1 W, then conduction losses worse case are:

PCOND +ID,RMS2@RDS(ON) +ǒ154 @10−3Ǔ2@13.6 +323 mW (eq. 15)

7. Off−time and on−time switching losses can be estimated based on the following calculations:

POFF +IPEAK @(VBULK )VCLAMP)@tF

2@TSW +0.335 @(127 )120 @2) @10 @10−9

2@15.4 @10−6 +40 mW (eq. 16)

Where, assume the VCLAMP is equal to 2 times of reflected voltage.

PON +IVALLEY @ǒVBULK )N@(VOUT )VF)Ǔ@tR

6@TSW +0.112 @(127 )100) @20 @10−9

6@15.4 @10−6 +5.5 mW (eq. 17)

It is noted that the overlap of voltage and current seen on MOSFET during turning on and off duration is dependent on the

snubber and parasitic capacitance seen from drain pin. Therefore the tF and tR in Equations 16 and 17 have to be modified after

measuring on the bench.

8. The theoretical total power is then

PMOSFET +323 )40 )5.5 +368.5 mW

9. If the NCP107xuz operates at DSS mode, then the losses caused by DSS mode should be counted as losses of this

device on the following calculation:

PDSS +ICC1 @VIN,MAX +1.5 @10−3 @375 +563 mW (eq. 18)

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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MOSFET Protection

As in any flyback design, it is important to limit the drain excursion to a safe value, e.g. below the MOSFET BVDSS which

is 700 V. Figure 53 a−b−c present possible implementations:

Figure 53. Different Options to Clamp the Leakage Spike

Figure 53a: the simple capacitor limits the voltage

according to the lateral MOSFET body−diode shall never b e

forward biased, either during start−up (because of a large

leakage inductance) or in normal operation as shown by

Figure 51. This condition sets the maximum voltage that can

be reflected during tF. As a result, the flyback voltage which

is reflected on the drain at the switch opening cannot be

larger than the input voltage. When selecting components,

you must adopt a turn ratio which adheres to the following

Equation 6. This option is only valid for low power

applications, e.g. below 5 W, otherwise chances exist to

destroy the MOSFET. After evaluating the leakage

inductance, you can compute C with (Equation 7). Typical

values are between 100 pF and up to 470 pF. Large

capacitors increase capacitive losses...

Figur e 53b: the most standard circuitry is called the RCD

network. You calculate RCLAMP and CCLAMP using the

following formulae:

RCLAMP +2@VCLAMP ǒVCLAMP )(VOUT )VF)@NǓ

LLEAK @ILEAK2@fSW (eq. 19)

CCLAMP +VCLAMP

VRIPPLE @fSW @RCLAMP (eq. 20)

VCLAMP is usually selected 50−80 V above the reflected

value N x (VOUT + VF). The diode needs to be a fast one

and a MUR160 represents a good choice. One major

drawback of the RCD network lies in its dependency upon

the peak current. Worse case occurs when IPEAK and VIN are

maximum and VOUT is close to reach the steady−state value.

Figur e 53c: this option is probably the most expensive of

all three but it offers the best protection degree. If you need

a very precise clamping level, you must implement a Zener

diode or a TVS. There are little technology differences

behind a standard Zener diode and a TVS. However, the die

area is far bigger for a transient suppressor than that of Zener.

A 5 W Zener diode like the 1N5388B will accept 180 W

peak power if it lasts less than 8.3 ms. If the peak current in

the worse case (e.g. when the PWM circuit maximum

current limit works) multiplied by the nominal zener voltage

exceeds these 180 W, then the diode will be destroyed when

the supply experiences overloads. A transient suppressor

like the P6KE200 still dissipates 5 W of continuous power

but is able to accept surges up to 600 W @ 1 ms. Select the

Zener or TVS clamping level between 40 to 80 volts above

the reflected output voltage when the supply is heavily

loaded.

As a good design practice, it is recommended to

implement one of this protection to ensure a maximum drain

pin voltage below 650 V (to have some margin between

drain pin voltage and BVDSS) during most stringent

operating conditions (high VIN and peak power condition).

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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Power Dissipation and Heatsinking

The NCP107xuz welcomes two dissipating terms, the

DSS current−source (when active) and the MOSFET. Thus,

PTOT = PDSS + PMOSFET. It is mandatory to properly

manage the heat generated by losses. If no precaution is

taken, risks exist to trigger the internal thermal shutdown

(TSD). To help dissipating the heat, the PCB designer must

foresee large copper areas around the package. Take the

PDIP−7 package as an example, when surrounded by a

surface approximately 200 mm2 of 35 mm copper, the

maximum power the device can thus evacuate is:

PMAX +TJ(max) *TAMB(max)

RqJA (eq. 21)

which gives around 1300 mW for an ambient of 50°C and

a maximum junction of 150°C. If the surface is not large

enough, the RqJA is growing and the maximum power the

device can evacuate decreases. Figure 54 gives a possible

layout to help drop the thermal resistance.

Figure 54. A Possible PCB Arrangement to Reduce

the Thermal Resistance Junction−to−Ambient

Bill of Material:

C1Bulk capacitor, input dc voltage is

connected to the capacitor

C2, R1, D1Clamping elements

C3VCC capacitor

OK1Opto−coupler

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ORDERING INFORMATION

Device Frequency

[kHz] RDS(ON)

[W]IPK

[mA] 2nd level

OCP Package Type Shipping

NCP1075AAP065G 65 13.5 400 enabled PDIP8 (Less pin#6)

50 Units /

Rail

NCP1075AAP100G 100 13.5 400 enabled PDIP8 (Less pin#6)

NCP1075BAP065G 65 13.5 400 enabled PDIP8 (Less pin#3)

NCP1075BAP100G 100 13.5 400 enabled PDIP8 (Less pin#3)

NCP1075BAP130G 130 13.5 400 enabled PDIP8 (Less pin#3)

NCP1076AAP065G 65 4.8 650 enabled PDIP8 (Less pin#6)

NCP1076AAP100G 100 4.8 650 enabled PDIP8 (Less pin#6)

NCP1076BAP065G 65 4.8 650 enabled PDIP8 (Less pin#3)

NCP1076BAP100G 100 4.8 650 enabled PDIP8 (Less pin#3)

NCP1076BAP130G 130 4.8 650 enabled PDIP8 (Less pin#3)

NCP1077AAP065G 65 4.8 800 enabled PDIP8 (Less pin#6)

NCP1077AAP100G 100 4.8 800 enabled PDIP8 (Less pin#6)

NCP1077BAP065G 65 4.8 800 enabled PDIP8 (Less pin#3)

NCP1077BAP100G 100 4.8 800 enabled PDIP8 (Less pin#3)

NCP1077BAP130G 130 4.8 800 enabled PDIP8 (Less pin#3)

NCP1079AAP065G 65 2.9 1050 enabled PDIP8 (Less pin#6)

NCP1079AAP100G 100 2.9 1050 enabled PDIP8 (Less pin#6)

NCP1079BAP065G 65 2.9 1050 enabled PDIP8 (Less pin#3)

NCP1079BAP100G 100 2.9 1050 enabled PDIP8 (Less pin#3)

NCP1079BAP130G 130 2.9 1050 enabled PDIP8 (Less pin#3)

NCP1075ABP065G 65 13.5 400 disabled PDIP8 (Less pin#6)

NCP1075ABP100G 100 13.5 400 disabled PDIP8 (Less pin#6)

NCP1075BBP065G 65 13.5 400 disabled PDIP8 (Less pin#3)

NCP1075BBP100G 100 13.5 400 disabled PDIP8 (Less pin#3)

NCP1075BBP130G 130 13.5 400 disabled PDIP8 (Less pin#3)

NCP1076ABP065G 65 4.8 650 disabled PDIP8 (Less pin#6)

NCP1076ABP100G 100 4.8 650 disabled PDIP8 (Less pin#6)

NCP1076BBP065G 65 4.8 650 disabled PDIP8 (Less pin#3)

NCP1076BBP100G 100 4.8 650 disabled PDIP8 (Less pin#3)

NCP1076BBP130G 130 4.8 650 disabled PDIP8 (Less pin#3)

NCP1077ABP065G 65 4.8 800 disabled PDIP8 (Less pin#6)

NCP1077ABP100G 100 4.8 800 disabled PDIP8 (Less pin#6)

NCP1077BBP065G 65 4.8 800 disabled PDIP8 (Less pin#3)

NCP1077BBP100G 100 4.8 800 disabled PDIP8 (Less pin#3)

NCP1077BBP130G 130 4.8 800 disabled PDIP8 (Less pin#3)

NCP1079ABP065G 65 2.9 1050 disabled PDIP8 (Less pin#6)

NCP1079ABP100G 100 2.9 1050 disabled PDIP8 (Less pin#6)

NCP1079BBP065G 65 2.9 1050 disabled PDIP8 (Less pin#3)

NCP1079BBP100G 100 2.9 1050 disabled PDIP8 (Less pin#3)

NCP1079BBP130G 130 2.9 1050 disabled PDIP8 (Less pin#3)

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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PACKAGE DIMENSIONS

PDIP−7 (PDIP−8 LESS PIN 6)

CASE 626A

ISSUE C

NOTE 8

TOP VIEW

SEATING

PLANE

0.010 CA

SIDE VIEW

END VIEW

WITH LEADS CONSTRAINED

NOTES:

1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.

2. CONTROLLING DIMENSION: INCHES.

3. DIMENSIONS A, A1 AND L ARE MEASURED WITH THE PACK-

AGE SEATED IN JEDEC SEATING PLANE GAUGE GS−3.

4. DIMENSIONS D, D1 AND E1 DO NOT INCLUDE MOLD FLASH

OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS ARE

NOT TO EXCEED 0.10 INCH.

5. DIMENSION E IS MEASURED AT A POINT 0.015 BELOW DATUM

PLANE H WITH THE LEADS CONSTRAINED PERPENDICULAR

TO DATUM C.

6. DIMENSION eB IS MEASURED AT THE LEAD TIPS WITH THE

LEADS UNCONSTRAINED.

7. DATUM PLANE H IS COINCIDENT WITH THE BOTTOM OF THE

LEADS, WHERE THE LEADS EXIT THE BODY.

8. PACKAGE CONTOUR IS OPTIONAL (ROUNDED OR SQUARE

CORNERS).

NOTE 5

e/2 A2

NOTE 3

MBMNOTE 6

DIM MIN MAX

INCHES

A−−−− 0.210

A1 0.015 −−−−

b0.014 0.022

C0.008 0.014

D0.355 0.400

D1 0.005 −−−−

e0.100 BSC

E0.300 0.325

M−−−− 10

−−− 5.33

0.38 −−−

0.35 0.56

0.20 0.36

9.02 10.16

0.13 −−−

2.54 BSC

7.62 8.26

−−− 10

MIN MAX

MILLIMETERS

E1 0.240 0.280 6.10 7.11

eB −−−− 0.430 −−− 10.92

0.060 TYP 1.52 TYP

A2 0.115 0.195 2.92 4.95

L0.115 0.150 2.92 3.81

°°

NCP1075A/B, NCP1076A/B, NCP1077A/B, NCP1079A/B

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PACKAGE DIMENSIONS

PDIP8 LESS PIN 3

CASE 626AS

ISSUE O

NOTE 6

TOP VIEW

SEATING

PLANE

0.010 CA

SIDE VIEW

END VIEW

WITH LEADS CONSTRAINED DIM MIN MAX

INCHES

A0.155 0.175

A1 0.020 0.040

b2 0.056 0.064

c0.008 0.012

D0.365 0.369

D1 0.005 0.080

e0.100 BSC

E0.300 0.325

M−−−− 10

3.94 4.45

0.51 1.02

1.42 1.63

0.20 0.30

9.27 9.37

0.13 2.03

2.54 BSC

7.62 8.25

−−− 10

MIN MAX

MILLIMETERS

NOTES:

1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.

2. CONTROLLING DIMENSION: INCHES.

3. DIMENSIONS A, A1 AND L ARE MEASURED WITH THE PACK-

AGE SEATED IN JEDEC SEATING PLANE GAUGE GS−3.

4. DIMENSIONS D, D1 AND E1 DO NOT INCLUDE MOLD FLASH

OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS ARE

NOT TO EXCEED 0.10 INCH.

5. DIMENSION E IS MEASURED AT A POINT 0.015 BELOW DATUM

PLANE H WITH THE LEADS CONSTRAINED PERPENDICULAR

TO DATUM C.

6. PACKAGE CONTOUR IS OPTIONAL (ROUNDED OR SQUARE

CORNERS).

E1 0.244 0.260 6.20 6.60

A3 0.015 BSC 0.38 BSC

b0.015 0.020 0.38 0.50

L0.115 0.135 2.92 3.43

°°

NOTE 5

e/2

NOTE 3

MBM

END VIEW

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