LTC1871-1
1
18711fb
TYPICAL APPLICATION
FEATURES
APPLICATIONS
DESCRIPTION
Wide Input Range, No RSENSE
Current Mode Boost,
Flyback and SEPIC Controller
The LTC
®
1871-1 is a wide input range, current mode, boost,
yback or SEPIC controller that drives an N-channel power
MOSFET and requires very few external components. It
eliminates the need for a current sense resistor by utilizing
the power MOSFETs on-resistance, thereby maximizing
effi ciency. Higher output voltage applications are possible
with the LTC1871-1 by connecting the SENSE pin to a
resistor in the source of the power MOSFET.
The IC’s operating frequency can be set with an external
resistor over a 50kHz to 1MHz range, and can be synchro-
nized to an external clock using the MODE/SYNC pin.
The LTC1871-1 differs from the LTC1871 by having a
lower pulse skip threshold, making it ideal for applica-
tions requiring constant frequency operation at light
loads. The lower pulse skip threshold also helps maintain
constant frequency operation in applications with a wide
input voltage range. For applications requiring primary-
to-secondary side isolation, please refer to the LTC1871
datasheet.
The LTC1871-1 is available in the 10-lead MSOP package.
Effi ciency of Figure 1
n High Effi ciency (No Sense Resistor Required)
n
Wide Input Voltage Range: 2.5V to 36V
n
Current Mode Control Provides Excellent
Transient Response
n
High Maximum Duty Cycle (92% Typ)
n
±2% RUN Pin Threshold with 100mV Hysteresis
n
±1% Internal Voltage Reference
n Ultra Low Pulse Skip Threshold for Wide Input
Range Applications
n Micropower Shutdown: IQ = 10µA
n
Programmable Operating Frequency
(50kHz to 1MHz) with One External Resistor
n
Synchronizable to an External Clock Up to 1.3 × fOSC
n
User-Controlled Pulse Skip or Burst Mode
®
Operation
n
Internal 5.2V Low Dropout Voltage Regulator
n
Output Overvoltage Protection
n
Capable of Operating with a Sense Resistor for High
Output Voltage Applications
n Small 10-Lead MSOP Package
n Telecom Power Supplies
n Portable Electronic Equipment
L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation.
No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Figure 1. High Effi ciency 3.3V Input, 5V Output Boost Converter (Bootstrapped)
+
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
LTC1871-1
RT
80.6k
1%
R2
37.4k
1%
R1
12.1k
1% CVCC
4.7µF
X5R
CIN
22µF
6.3V
×2
M1
D1
L1
1µH
RC
22k
CC1
6.8nF
CC2
47pF
COUT1
150µF
6.3V
×4
VIN
3.3V
VOUT
5V
7A
(10A PEAK)
GND
18711 F01a
+
COUT2
22µF
6.3V
X5R
×2
CIN: TAIYO YUDEN JMK325BJ226MM
COUT1: PANASONIC EEFUEOJ151R
COUT2: TAIYO YUDEN JMK325BJ226MM
D1: MBRB2515L
L1: SUMIDA CEP125-H 1R0MH
M1: FAIRCHILD FDS7760A OUTPUT CURRENT (A)
30
EFFICIENCY (%)
90
100
80
50
70
60
40
0.001 0.1 1 10
18711 F01b
0.01
Burst Mode
OPERATION
PULSE-SKIP
MODE
LTC1871-1
2
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PIN CONFIGURATION ABSOLUTE MAXIMUM RATINGS
VIN Voltage ............................................... 0.3V to 36V
INTVCC Voltage ............................................ –0.3V to 7V
INTVCC Output Current .......................................... 50mA
GATE Voltage ............................ –0.3V to VINTVCC + 0.3V
ITH, FB Voltages ....................................... –0.3V to 2.7V
RUN, MODE/SYNC Voltages ....................... –0.3V to 7V
FREQ Voltage ............................................ –0.3V to 1.5V
SENSE Pin Voltage .................................... –0.3V to 36V
Operating Junction Temperature Range (Note 2)
LTC1871E-1 ......................................... –40°C to 85°C
LTC1871I-1 ........................................ –40°C to 125°C
Junction Temperature (Note 3) ............................ 125°C
Storage Temperature Range ................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
(Note 1)
1
2
3
4
5
RUN
ITH
FB
FREQ
MODE/
SYNC
10
9
8
7
6
SENSE
VIN
INTVCC
GATE
GND
TOP VIEW
MS PACKAGE
10-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 120°C/W
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC1871EMS-1#PBF LTC1871EMS-1#TRPBF LTCTV 10-Lead Plastic MSOP –40°C to 85°C
LTC1871IMS-1#PBF LTC1871IMS-1#TRPBF LTCTV 10-Lead Plastic MSOP –40°C to 125°C
LEAD BASED FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC1871EMS-1 LTC1871EMS-1#TR LTCTV 10-Lead Plastic MSOP –40°C to 85°C
LTC1871IMS-1 LTC1871IMS-1#TR LTCTV 10-Lead Plastic MSOP –40°C to 125°C
Consult LTC Marketing for parts specifi ed with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi cations, go to: http://www.linear.com/tapeandreel/
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Main Control Loop
VIN(MIN) Minimum Input Voltage 2.5 V
I-Grade (Note 2) 2.5 V
IQInput Voltage Supply Current (Note 4)
Continuous Mode VMODE/SYNC = 5V, VFB = 1.4V, VITH = 0.75V 550 1000 µA
VMODE/SYNC = 5V, VFB = 1.4V, VITH = 0.75V,
I-Grade (Note 2)
550 1000 µA
Burst Mode Operation, No Load VMODE/SYNC = 0V, VITH = 0V (Note 5) 250 500 µA
VMODE/SYNC = 0V, VITH = 0V (Note 5),
I-Grade (Note 2)
250 500 µA
Shutdown Mode VRUN = 0V 10 20 µA
VRUN = 0V, I-Grade (Note 2) 10 20 µA
ELECTRICAL CHARACTERISTICS
The l denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise specifi ed.
LTC1871-1
3
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ELECTRICAL CHARACTERISTICS
The l denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise specifi ed.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
VRUN+Rising RUN Input Threshold Voltage 1.348 V
VRUNFalling RUN Input Threshold Voltage
1.223
1.198
1.248 1.273
1.298
V
V
VRUN(HYST) RUN Pin Input Threshold Hysteresis 50 100 150 mV
I-Grade (Note 2) 35 100 175 mV
IRUN RUN Input Current 160 nA
VFB Feedback Voltage VITH = 0.4V (Note 5)
1.218
1.212
1.230 1.242
1.248
V
V
VITH = 0.4V (Note 5), I-Grade (Note 2) 1.205 1.255 V
IFB FB Pin Input Current VITH = 0.4V (Note 5) 18 60 nA
ΔVFB
ΔVIN
Line Regulation 2.5V ≤ VIN ≤ 30V 0.002 0.02 %/V
2.5V ≤ VIN ≤ 30V, I-Grade (Note 2) 0.002 0.03 %/V
ΔVFB
ΔVITH
Load Regulation VMODE/SYNC = 0V, VITH = 0.5V to 0.9V (Note 5) 1 –0.1 %
VMODE/SYNC = 0V, VITH = 0.5V to 0.9V (Note 5)
I-Grade (Note 2)
1 –0.1 %
ΔVFB(OV) ΔFB Pin, Overvoltage Lockout VFB(OV) – VFB(NOM) in Percent 2.5 6 10 %
gmError Amplifi er Transconductance ITH Pin Load = ±5µA (Note 5) 650 µmho
VITH(BURST) Burst Mode Operation ITH Pin Voltage Falling ITH Voltage (Note 5) 195 mV
VSENSE(MAX) Maximum Current Sense Input Threshold Duty Cycle < 20% 120 150 180 mV
Duty Cycle < 20%, I-Grade (Note 2) 100 200 mV
ISENSE(ON) SENSE Pin Current (GATE High) VSENSE = 0V 35 50 µA
ISENSE(OFF) SENSE Pin Current (GATE Low) VSENSE = 30V 0.1 5 µA
Oscillator
fOSC Oscillator Frequency RFREQ = 80k 250 300 350 kHz
RFREQ = 80k, I-Grade (Note 2) 250 300 350 kHz
Oscillator Frequency Range 50 1000 kHz
I-Grade (Note 2) 50 1000 kHz
DMAX Maximum Duty Cycle 87 92 97 %
I-Grade (Note 2) 87 92 97 %
fSYNC/fOSC Recommended Maximum Synchronized
Frequency Ratio
fOSC = 300kHz (Note 6) 1.25 1.30
fOSC = 300kHz (Note 6), I-Grade (Note 2) 1.25 1.30
tSYNC(MIN) MODE/SYNC Minimum Input Pulse Width VSYNC = 0V to 5V 25 ns
tSYNC(MAX) MODE/SYNC Maximum Input Pulse Width VSYNC = 0V to 5V 0.8/fOSC ns
VIL(MODE) Low Level MODE/SYNC Input Voltage 0.3 V
I-Grade (Note 2) 0.3 V
VIH(MODE) High Level MODE/SYNC Input Voltage 1.2 V
I-Grade (Note 2) 1.2 V
RMODE/SYNC MODE/SYNC Input Pull-Down Resistance 50 k
VFREQ Nominal FREQ Pin Voltage 0.62 V
LTC1871-1
4
18711fb
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC1871E-1 is guaranteed to meet performance specifi cations
from 0°C to 85°C junction temperature. Specifi cations over the – 40°C
to 85°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC1871I-1 is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
T
J = TA + (PD • 110°C/W)
Note 4: The dynamic input supply current is higher due to power MOSFET
gate charging (QG • fOSC). See Applications Information.
Note 5: The LTC1871-1 is tested in a feedback loop which servos VFB to
the reference voltage with the ITH pin forced to the midpoint of its voltage
range (0.3V ≤ VITH ≤ 1.2V, midpoint = 0.75V).
Note 6: In a synchronized application, the internal slope compensation
gain is increased by 25%. Synchronizing to a signifi cantly higher ratio will
reduce the effective amount of slope compensation, which could result in
subharmonic oscillation for duty cycles greater than 50%.
Note 7: Rise and fall times are measured at 10% and 90% levels.
The l denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise specifi ed.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Low Dropout Regulator
VINTVCC INTVCC Regulator Output Voltage VIN = 7.5V 5.0 5.2 5.4 V
VIN = 7.5V, I-Grade (Note 2) 5.0 5.2 5.4 V
ΔVINTVCC
ΔVIN1
INTVCC Regulator Line Regulation 7.5V ≤ VIN ≤ 15V 8 25 mV
ΔVINTVCC
ΔVIN2
INTVCC Regulator Line Regulation 15V ≤ VIN ≤ 30V 70 200 mV
VLDO(LOAD) INTVCC Load Regulation 0 ≤ IINTVCC ≤ 20mA, VIN = 7.5V 2 –0.2 %
VDROPOUT INTVCC Regulator Dropout Voltage INTVCC Load = 20mA 280 mV
IINTVCC Bootstrap Mode INTVCC Supply RUN = 0V, SENSE = 5V 10 20 µA
I-Grade (Note 2) 30 µA
GATE Driver
trGATE Driver Output Rise Time CL = 3300pF (Note 7) 17 100 ns
tfGATE Driver Output Fall Time CL = 3300pF (Note 7) 8 100 ns
TYPICAL PERFORMANCE CHARACTERISTICS
FB Voltage vs Temp FB Voltage Line Regulation FB Pin Current vs Temperature
TEMPERATURE (°C)
–50
FB VOLTAGE (V)
1.23
1.24
150
18711 G01
1.22
1.21 050 100
–25 25 75 125
1.25
VIN (V)
0
1.229
FB VOLTAGE (V)
1.230
1.231
5101520
18711 G02
25 30 35
TEMPERATURE (°C)
–50
0
FB PIN CURRENT (nA)
10
20
30
40
60
–25 250 50 10075
18711 G03
125 150
50
LTC1871-1
5
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TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Mode IQ vs VIN Shutdown Mode IQ vs Temperature Burst Mode IQ vs VIN
Burst Mode IQ vs Temperature Dynamic IQ vs Frequency
Gate Drive Rise and
Fall Time vs CL
VIN (V)
0
0
SHUTDOWN MODE IQ (µA)
10
20
10 20 30 40
18711 G04
30
TEMPERATURE (°C)
–50
0
SHUTDOWN MODE IQ (µA)
5
10
15
20
–25 0 25 50
18711 G05
75 100 125 150
VIN = 5V
VIN (V)
0
0
Burst Mode IQ (µA)
100
200
300
400
600
10 20
18711 G06
30 40
500
TEMPERATURE (°C)
–50
0
Burst Mode IQ (µA)
200
500
050 75
18711 G07
100
400
300
–25 25 100 125 150
FREQUENCY (kHz)
0
0
IQ (mA)
2
6
8
10
800
18
18711 G08
4
400 1200
600
200 1000
12
14
16
CL = 3300pF
IQ(TOT) = 550µA + Qg • f
CL (pF)
0
0
TIME (ns)
10
20
30
40
60
2000 4000 6000 8000
18711 G09
10000 12000
50
RISE TIME
FALL TIME
RUN Thresholds vs VIN RUN Thresholds vs Temperature RT vs Frequency
VIN (V)
0
1.2
RUN THRESHOLDS (V)
1.3
1.4
10 20 30 40
18711 G10
1.5
TEMPERATURE (°C)
–50
RUN THRESHOLDS (V)
1.30
1.35
150
18711 G11
1.25
1.20 050 100
–25 25 75 125
1.40
FREQUENCY (kHz)
100
RT (kΩ)
300
1000
18711 G12
10
100
200 1000
900
800700600
500
400
0
LTC1871-1
6
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TYPICAL PERFORMANCE CHARACTERISTICS
Frequency vs Temperature
Maximum Sense Threshold
vs Temperature
SENSE Pin Current
vs Temperature
INTVCC Load Regulation INTVCC Line Regulation
INTVCC Dropout Voltage
vs Current, Temperature
RUN (Pin 1): The RUN pin provides the user with an
accurate means for sensing the input voltage and pro-
gramming the start-up threshold for the converter. The
falling RUN pin threshold is nominally 1.248V and the
comparator has 100mV of hysteresis for noise immunity.
When the RUN pin is below this input threshold, the IC
is shut down and the VIN supply current is kept to a low
value (typ 10µA). The Absolute Maximum Rating for the
voltage on this pin is 7V.
ITH (Pin 2): Error Amplifi er Compensation Pin. The
current comparator input threshold increases with this
control voltage. Nominal voltage range for this pin is 0V
to 1.40V.
FB (Pin 3): Receives the feedback voltage from the external
resistor divider across the output. Nominal voltage for
this pin in regulation is 1.230V.
FREQ (Pin 4): A resistor from the FREQ pin to ground
programs the operating frequency of the chip. The nominal
voltage at the FREQ pin is 0.6V.
TEMPERATURE (°C)
–50
275
GATE FREQUENCY (kHz)
280
290
295
300
325
310
050 75
18711 G13
285
315
320
305
–25 25 100 125 150
TEMPERATURE (°C)
–50
140
MAX SENSE THRESHOLD (mV)
145
150
155
160
–25 0 25 50
18711 G14
75 100 125 150
TEMPERATURE (°C)
–50
25
SENSE PIN CURRENT (µA)
30
35
050 75
18711 G15
–25 25 100 125 150
GATE HIGH
VSENSE = 0V
INTVCC LOAD (mA)
0
INTVCC VOLTAGE (V)
5.2
30 50 80
18711 G16
5.1
5.0
10 20 40 60 70
VIN = 7.5V
VIN (V)
0
5.1
INTVCC VOLTAGE (V)
5.2
5.3
10 20 30 40
18711 G17
5.4
51525 35
INTVCC LOAD (mA)
0
0
DROPOUT VOLTAGE (mV)
50
150
200
250
500
350
510
18711 G18
100
400
450
300
15 20
150°C
75°C
125°C
25°C
–50°C
0°C
PIN FUNCTIONS
LTC1871-1
7
18711fb
PIN FUNCTIONS
MODE/SYNC (Pin 5): This input controls the operating
mode of the converter and allows for synchronizing the
operating frequency to an external clock. If the MODE/
SYNC pin is connected to ground, Burst Mode operation
is enabled. If the MODE/SYNC pin is connected to INTVCC,
or if an external logic-level synchronization signal is ap-
plied to this input, Burst Mode operation is disabled and
the IC operates in a continuous mode.
GND (Pin 6): Ground Pin.
GATE (Pin 7): Gate Driver Output.
I
NTVCC (Pin 8): The Internal 5.20V Regulator Output.
The gate driver and control circuits are powered from
this voltage. Decouple this pin locally to the IC ground
with a minimum of 4.7µF low ESR tantalum or ceramic
capacitor.
VIN (Pin 9): Main Supply Pin. Must be closely decoupled
to ground.
SENSE (Pin 10): The Current Sense Input for the Control
Loop. Connect this pin to the drain of the power MOSFET
for VDS sensing and highest effi ciency. Alternatively, the
SENSE pin may be connected to a resistor in the source
of the power MOSFET. Internal leading edge blanking is
provided for both sensing methods.
BLOCK DIAGRAM
+
+
+
1.230V
85mV OV
50k
EA
UV
TO
START-UP
CONTROL
BURST
COMPARATOR
S
R
Q
LOGIC
PWM LATCH
CURRENT
COMPARATOR
0.175V
1.230V
5.2V
+
2.00V
1.230V
SLOPE
1.230V
ILOOP
FB
ITH
+
gm
3
MODE/SYNC
5
FREQ
4
2
INTVCC
8LDO
V-TO-I
OSCV-TO-I
SLOPE
COMPENSATION
BIAS AND
START-UP
CONTROL
VIN
BIAS VREF
IOSC
RLOOP
+
+
C1
SENSE
10
GND
18711 BD
6
GATE
INTVCC
GND
7
VIN
1.248V
9
RUN
C2
1
0.6V
LTC1871-1
8
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OPERATION
Figure 2. Using the SENSE Pin On the LTC1871-1
Main Control Loop
The LTC1871-1 is a constant frequency, current mode
controller for DC/DC boost, SEPIC and fl yback converter
applications. The LTC1871-1 is distinguished from con-
ventional current mode controllers because the current
control loop can be closed by sensing the voltage drop
across the power MOSFET switch instead of across a
discrete sense resistor, as shown in Figure 2. This sensing
technique improves effi ciency, increases power density,
and reduces the cost of the overall solution.
which causes the current comparator C1 to trip at a higher
peak inductor current value. The average inductor current
will therefore rise until it equals the load current, thereby
maintaining output regulation.
The nominal operating frequency of the LTC1871-1 is
programmed using a resistor from the FREQ pin to ground
and can be controlled over a 50kHz to 1000kHz range. In
addition, the internal oscillator can be synchronized to
an external clock applied to the MODE/SYNC pin and can
be locked to a frequency between 100% and 130% of its
nominal value. When the MODE/SYNC pin is left open, it
is pulled low by an internal 50k resistor and Burst Mode
operation is enabled. If this pin is taken above 2V or an
external clock is applied, Burst Mode operation is disabled
and the IC operates in continuous mode. With no load (or
an extremely light load), the controller will skip pulses in
order to maintain regulation and prevent excessive output
ripple.
The RUN pin controls whether the IC is enabled or is in a low
current shutdown state. A micropower 1.248V reference
and comparator C2 allow the user to program the supply
voltage at which the IC turns on and off (comparator C2
has 100mV of hysteresis for noise immunity). With the
RUN pin below 1.248V, the chip is off and the input supply
current is typically only 10µA.
An overvoltage comparator OV senses when the FB pin
exceeds the reference voltage by 6.5% and provides a
reset pulse to the main RS latch. Because this RS latch is
reset-dominant, the power MOSFET is actively held off for
the duration of an output overvoltage condition.
The LTC1871-1 can be used either by sensing the voltage
drop across the power MOSFET or by connecting the
SENSE pin to a conventional shunt resistor in the source
of the power MOSFET, as shown in Figure 2. Sensing the
voltage across the power MOSFET maximizes converter
effi ciency and minimizes the component count, but limits
the output voltage to the maximum rating for this pin (36V).
By connecting the SENSE pin to a resistor in the source
of the power MOSFET, the user is able to program output
voltages signifi cantly greater than 36V.
COUT
VSW
VSW
2a. SENSE Pin Connection for
Maximum Efficiency (VSW < 36V)
VOUT
VIN
GND
LD
+
COUT
RS
18711 F02
2b. SENSE Pin Connection for Precise
Control of Peak Current or for VSW > 36V
VOUT
VIN
GND
LD
+
GATE
GND
VIN
SENSE
GATE
GND
VIN
SENSE
For circuit operation, please refer to the Block Diagram of
the IC and Figure 1. In normal operation, the power MOSFET
is turned on when the oscillator sets the PWM latch and
is turned off when the current comparator C1 resets the
latch. The divided-down output voltage is compared to an
internal 1.230V reference by the error amplifi er EA, which
outputs an error signal at the ITH pin. The voltage on the
ITH pin sets the current comparator C1 input threshold.
When the load current increases, a fall in the FB voltage
relative to the reference voltage causes the ITH pin to rise,
LTC1871-1
9
18711fb
OPERATION
MOSFET RDS(ON). If the ITH pin drops below 0.175V, the
Burst Mode comparator B1 will turn off the power MOSFET
and scale back the quiescent current of the IC to 250µA
(sleep mode). In this condition, the load current will be
supplied by the output capacitor until the ITH voltage rises
above the 50mV hysteresis of the burst comparator. At
light loads, short bursts of switching (where the average
inductor current is 20% of its maximum value) followed
by long periods of sleep will be observed, thereby greatly
improving converter effi ciency. Oscilloscope waveforms
illustrating Burst Mode operation are shown in Figure 3.
Pulse-Skip Mode Operation
With the MODE/SYNC pin tied to a DC voltage above 2V,
Burst Mode operation is disabled. The internal, 0.525V
buffered ITH burst clamp is removed, allowing the ITH
pin to directly control the current comparator from no
load to full load. With no load, the ITH pin is driven below
0.175V, the power MOSFET is turned off and sleep mode
is invoked. Oscilloscope waveforms illustrating this mode
of operation are shown in Figure 4.
When an external clock signal drives the MODE/SYNC
pin at a rate faster than the chip’s internal oscillator, the
oscillator will synchronize to it. In this synchronized mode,
Burst Mode operation is disabled. The constant frequency
associated with synchronized operation provides a more
controlled noise spectrum from the converter, at the ex-
pense of overall system effi ciency of light loads.
Programming the Operating Mode
For applications where maximizing the effi ciency at very
light loads (e.g., <100µA) is a high priority, the current
in the output divider could be decreased to a few micro-
amps and Burst Mode operation should be applied (i.e.,
the MODE/SYNC pin should be connected to ground).
In applications where fi xed frequency operation is more
critical than low current effi ciency, or where the lowest
output ripple is desired, pulse-skip mode operation should
be used and the MODE/SYNC pin should be connected
to the INTVCC pin. This allows discontinuous conduction
mode (DCM) operation down to near the limit defi ned
by the chip’s minimum on-time (about 175ns). Below
this output current level, the converter will begin to skip
cycles in order to maintain output regulation. Figures 3
and 4 show the light load switching waveforms for Burst
Mode and pulse-skip mode operation for the converter
in Figure 1.
Burst Mode Operation
Burst Mode operation is selected by leaving the MODE/
SYNC pin unconnected or by connecting it to ground. In
normal operation, the range on the ITH pin corresponding to
no load to full load is 0.30V to 1.2V. In Burst Mode opera-
tion, if the error amplifi er EA drives the ITH voltage below
0.525V, the buffered ITH input to the current comparator
C1 will be clamped at 0.525V (which corresponds to 25%
of maximum load current). The inductor current peak is
then held at approximately 30mV divided by the power
Figure 3. LTC1871-1 Burst Mode Operation
(MODE/SYNC = 0V) at Low Output Current
Figure 4. LTC1871-1 Low Output Current Operation with
Burst Mode Operation Disabled (MODE/SYNC = INTVCC)
VOUT
50mV/DIV
IL
5A/DIV
10µs/DIV 18711 F03
VIN = 3.3V
VOUT = 5V
IOUT = 500mA
MODE/SYNC = 0V
(Burst Mode OPERATION)
VOUT
50mV/DIV
IL
5A/DIV
2µs/DIV 18711 F04
VIN = 3.3V
VOUT = 5V
IOUT = 500mA
MODE/SYNC = INTVCC
(PULSE-SKIP MODE)
LTC1871-1
10
18711fb
When the oscillators internal logic circuitry detects a
synchronizing signal on the MODE/SYNC pin, the in-
ternal oscillator ramp is terminated early and the slope
compensation is increased by approximately 30%. As
a result, in applications requiring synchronization, it is
recommended that the nominal operating frequency of
the IC be programmed to be about 75% of the external
clock frequency. Attempting to synchronize to too high an
external frequency (above 1.3fO) can result in inadequate
slope compensation and possible subharmonic oscillation
(or jitter).
The external clock signal must exceed 2V for at least 25ns,
and should have a maximum duty cycle of 80%, as shown
in Figure 5. The MOSFET turn on will synchronize to the
rising edge of the external clock signal.
0.6V, and the current that fl ows into the FREQ pin is used
to charge and discharge an internal oscillator capacitor. A
graph for selecting the value of RT for a given operating
frequency is shown in Figure 6.
APPLICATIONS INFORMATION
Figure 5. MODE/SYNC Clock Input and Switching
Waveforms for Synchronized Operation
Figure 6. Timing Resistor (RT) Value
18711 F05
2V TO 7V
MODE/
SYNC
GATE
IL
tMIN = 25ns
0.8T
D = 40%
T T = 1/fO
Programming the Operating Frequency
The choice of operating frequency and inductor value is
a tradeoff between effi ciency and component size. Low
frequency operation improves effi ciency by reducing
MOSFET and diode switching losses. However, lower
frequency operation requires more inductance for a given
amount of load current.
The LTC1871-1 uses a constant frequency architecture that
can be programmed over a 50kHz to 1000kHz range with
a single external resistor from the FREQ pin to ground, as
shown in Figure 1. The nominal voltage on the FREQ pin is
FREQUENCY (kHz)
100
RT (kΩ)
300
1000
18711 F06
10
100
200 1000
900
800700600
500
400
0
INTVCC Regulator Bypassing and Operation
An internal, P-channel low dropout voltage regulator pro-
duces the 5.2V supply which powers the gate driver and
logic circuitry within the LTC1871-1, as shown in Figure 7.
The INTVCC regulator can supply up to 50mA and must be
bypassed to ground immediately adjacent to the IC pins
with a minimum of 4.7µF tantalum or ceramic capacitor.
Good bypassing is necessary to supply the high transient
currents required by the MOSFET gate driver.
For input voltages that don’t exceed 7V (the absolute
maximum rating for this pin), the internal low dropout
regulator in the LTC1871-1 is redundant and the INTVCC
pin can be shorted directly to the VIN pin. With the INTVCC
pin shorted to VIN, however, the divider that programs the
regulated INTVCC voltage will draw 10µA of current from
the input supply, even in shutdown mode. For applications
that require the lowest shutdown mode input supply cur-
rent, do not connect the INTVCC pin to VIN. Regardless of
whether the INTVCC pin is shorted to VIN or not, it is always
necessary to have the driver circuitry bypassed with a
4.7μF tantalum or low ESR ceramic capacitor to ground
immediately adjacent to the INTVCC and GND pins.
In an actual application, most of the IC supply current is
used to drive the gate capacitance of the power MOSFET.
LTC1871-1
11
18711fb
APPLICATIONS INFORMATION
Figure 7. Bypassing the LDO Regulator and Gate Driver Supply
+
+
1.230V
R2 R1
P-CH
5.2V
DRIVER GATE
CVCC
4.7µF
CIN
INPUT
SUPPLY
2.5V TO 30V
GND
PLACE AS CLOSE AS
POSSIBLE TO DEVICE PINS
M1
18711 F07
INTVCC
VIN
GND
LOGIC
As a result, high input voltage applications in which a
large power MOSFET is being driven at high frequencies
can cause the LTC1871-1 to exceed its maximum junc-
tion temperature rating. The junction temperature can be
estimated using the following equations:
I
Q(TOT) ≈ IQ + f • QG
P
IC = VIN • (IQ + f • QG)
T
J = TA + PIC • RTH(JA)
The total quiescent current IQ(TOT) consists of the static
supply current (IQ) and the current required to charge and
discharge the gate of the power MOSFET. The 10-pin MSOP
package has a thermal resistance of RTH(JA) = 120°C/W.
As an example, consider a power supply with VIN = 5V and
VO = 12V at IO = 1A. The switching frequency is 500kHz,
and the maximum ambient temperature is 70°C. The power
MOSFET chosen is the IRF7805, which has a maximum
RDS(ON) of 11m (at room temperature) and a maximum
total gate charge of 37nC (the temperature coeffi cient of
the gate charge is low).
I
Q(TOT) = 600µA + 37nC • 500kHz = 19.1mA
P
IC = 5V • 19.1mA = 95mW
T
J = 70°C + 120°C/W • 95mW = 81.4°C
This demonstrates how signifi cant the gate charge current
can be when compared to the static quiescent current in
the IC.
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in a continuous mode at high VIN. A tradeoff
between the operating frequency and the size of the power
MOSFET may need to be made in order to maintain a reliable
IC junction temperature. Prior to lowering the operating
frequency, however, be sure to check with power MOSFET
manufacturers for their latest-and-greatest low QG, low
RDS(ON) devices. Power MOSFET manufacturing tech-
nologies are continually improving, with newer and better
performance devices being introduced almost yearly.
Output Voltage Programming
The output voltage is set by a resistor divider according
to the following formula:
VO=1.230V 1+R2
R1
The external resistor divider is connected to the output
as shown in Figure 1, allowing remote voltage sensing.
The resistors R1 and R2 are typically chosen so that the
LTC1871-1
12
18711fb
APPLICATIONS INFORMATION
error caused by the current fl owing into the FB pin dur-
ing normal operation is less than 1% (this translates to a
maximum value of R1 of about 250k).
Programming Turn-On and Turn-Off Thresholds
with the RUN Pin
The LTC1871-1 contains an independent, micropower volt-
age reference and comparator detection circuit that remains
active even when the device is shut down, as shown in
Figure 8. This allows users to accurately program an input
voltage at which the converter will turn on and off. The
falling threshold voltage on the RUN pin is equal to the
internal reference voltage of 1.248V. The comparator has
100mV of hysteresis to increase noise immunity.
The turn-on and turn-off input voltage thresholds are
programmed using a resistor divider according to the
following formulas:
VIN(OFF) =1.248V 1+R2
R1
VIN(ON) =1.348V 1+R2
R1
The resistor R1 is typically chosen to be less than 1M.
For applications where the RUN pin is only to be used as
a logic input, the user should be aware of the 7V Absolute
Maximum Rating for this pin! The RUN pin can be con-
nected to the input voltage through an external 1M resistor,
as shown in Figure 8c, for “always on” operation.
Figure 8a. Programming the Turn-On and Turn-Off Thresholds Using the RUN Pin
Figure 8c. External Pull-Up Resistor On
RUN Pin for “Always On” Operation
Figure 8b. On/Off Control Using External Logic
+
RUN
COMPARATOR
VIN
RUN
R2
R1
INPUT
SUPPLY OPTIONAL
FILTER
CAPACITOR
+
GND
18711 F8a
BIAS AND
START-UP
CONTROL
1.248V
µPOWER
REFERENCE
6V
+
RUN
COMPARATOR
1.248V
18711 F08b
RUN
6V
EXTERNAL
LOGIC CONTROL
+
RUN
COMPARATOR
VIN
RUN
R2
1M
INPUT
SUPPLY
+
GND 1.248V
18711 F08c
6V
LTC1871-1
13
18711fb
APPLICATIONS INFORMATION
Application Circuits
A basic LTC1871-1 application circuit is shown in
Figure 1. External component selection is driven by the
characteristics of the load and the input supply. The
rst topology to be analyzed will be the boost converter,
followed by SEPIC (single ended primary inductance
converter).
Boost Converter: Duty Cycle Considerations
For a boost converter operating in a continuous conduction
mode (CCM), the duty cycle of the main switch is:
D=VO+VD–V
IN
VO+VD
where VD is the forward voltage of the boost diode. For
converters where the input voltage is close to the output
voltage, the duty cycle is low and for converters that develop
a high output voltage from a low voltage input supply,
the duty cycle is high. The maximum output voltage for a
boost converter operating in CCM is:
VO(MAX) =VIN(MIN)
1–DMAX
( )
–V
D
The maximum duty cycle capability of the LTC1871-1 is
typically 92%. This allows the user to obtain high output
voltages from low input supply voltages.
Boost Converter: The Peak and Average Input Currents
The control circuit in the LTC1871-1 is measuring the input
current (either by using the RDS(ON) of the power MOSFET
or by using a sense resistor in the MOSFET source), so
the output current needs to be refl ected back to the input
in order to dimension the power MOSFET properly. Based
on the fact that, ideally, the output power is equal to the
input power, the maximum average input current is:
IIN(MAX) =IO(MAX)
1–DMAX
The peak input current is:
IIN(PEAK) =1+
2
IO(MAX)
1–DMAX
The maximum duty cycle, DMAX, should be calculated at
minimum VIN.
Boost Converter: Ripple Current ΔIL and the ‘χ’ Factor
The constant ‘χ’ in the equation above represents the
percentage peak-to-peak ripple current in the inductor,
relative to its maximum value. For example, if 30% ripple
current is chosen, then χ = 0.30, and the peak current is
15% greater than the average.
For a current mode boost regulator operating in CCM,
slope compensation must be added for duty cycles above
50% in order to avoid subharmonic oscillation. For the
LTC1871-1, this ramp compensation is internal. Having an
internally fi xed ramp compensation waveform, however,
does place some constraints on the value of the inductor
and the operating frequency. If too large an inductor is
used, the resulting current ramp (ΔIL) will be small relative
to the internal ramp compensation (at duty cycles above
50%), and the converter operation will approach voltage
mode (ramp compensation reduces the gain of the current
loop). If too small an inductor is used, but the converter
is still operating in CCM (near critical conduction mode),
the internal ramp compensation may be inadequate to
prevent subharmonic oscillation. To ensure good current
mode gain and avoid subharmonic oscillation, it is recom-
mended that the ripple current in the inductor fall in the
range of 20% to 40% of the maximum average current.
For example, if the maximum average input current is
1A, choose a ΔIL between 0.2A and 0.4A, and a value ‘χ
between 0.2 and 0.4.
Boost Converter: Inductor Selection
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the inductor,
the inductor value can be determined using the following
equation:
L=VIN(MIN)
IL f •D
MAX
where:
IL=IO(MAX)
1–DMAX
LTC1871-1
14
18711fb
APPLICATIONS INFORMATION
Figure 9. Discontinuous Mode Waveforms
Remember that boost converters are not short-circuit
protected. Under a shorted output condition, the inductor
current is limited only by the input supply capability. For
applications requiring a step-up converter that is short-
circuit protected, please refer to the applications section
covering SEPIC converters.
The minimum required saturation current of the inductor
can be expressed as a function of the duty cycle and the
load current, as follows:
IL(SAT) 1+
2
IO(MAX)
1–DMAX
The saturation current rating for the inductor should be
checked at the minimum input voltage (which results
in the highest inductor current) and maximum output
current.
Boost Converter: Operating in Discontinuous Mode
Discontinuous mode operation occurs when the load cur-
rent is low enough to allow the inductor current to run out
during the off-time of the switch, as shown in Figure 9.
Once the inductor current is near zero, the switch and diode
capacitances resonate with the inductance to form damped
ringing at 1MHz to 10MHz. If the off-time is long enough,
the drain voltage will settle to the input voltage.
Depending on the input voltage and the residual energy
in the inductor, this ringing can cause the drain of the
power MOSFET to go below ground where it is clamped
by the body diode. This ringing is not harmful to the IC
and it has not been shown to contribute signifi cantly to
EMI. Any attempt to damp it with a snubber will degrade
the effi ciency.
MOSFET DRAIN
VOLTAGE
2V/DIV
INDUCTOR
CURRENT
2A/DIV
2µs/DIV 18711 F09
VIN = 3.3V IOUT = 200mA
VOUT = 5V
Boost Converter: Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High effi ciency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ
®
cores. Actual core loss is independent of core
size for a fi xed inductor value, but is very dependent on
the inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore, copper losses
will increase. Generally, there is a tradeoff between core
losses and copper losses that needs to be balanced.
Ferrite designs have very low core losses and are pre-
ferred at high switching frequencies, so design goals can
concentrate on copper losses and preventing saturation.
Ferrite core material saturates “hard,” meaning that the
inductance collapses rapidly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequently, output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good,
low cost core material for toroids, but is more expensive
than ferrite. A reasonable compromise from the same
manufacturer is Kool Mµ.
Boost Converter: Power MOSFET Selection
The power MOSFET serves two purposes in the LTC1871-1:
it represents the main switching element in the power path,
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFETs
thermal resistances (RTH(JC) and RTH(JA)).
The gate drive voltage is set by the 5.2V INTVCC low drop
regulator. Consequently, logic-level threshold MOSFETs
should be used in most LTC1871-1 applications. If low
input voltage operation is expected (e.g., supplying power
LTC1871-1
15
18711fb
APPLICATIONS INFORMATION
DUTY CYCLE
0
MAXIMUM CURRENT SENSE VOLTAGE (mV)
100
150
0.8
18711 F10
50
00.2 0.4 0.5 1.0
200
Figure 10. Maximum SENSE Threshold Voltage vs Duty Cycle
from a lithium-ion battery or a 3.3V logic supply), then
sublogic-level threshold MOSFETs should be used.
Pay close attention to the BVDSS specifi cations for the
MOSFETs relative to the maximum actual switch voltage in
the application. Many logic-level devices are limited to 30V
or less, and the switch node can ring during the turn-off of
the MOSFET due to layout parasitics. Check the switching
waveforms of the MOSFET directly across the drain and
source terminals using the actual PC board layout (not
just on a lab breadboard!) for excessive ringing.
During the switch on-time, the control circuit limits the
maximum voltage drop across the power MOSFET to about
150mV (at low duty cycle). The peak inductor current
is therefore limited to 150mV/RDS(ON). The relationship
between the maximum load current, duty cycle and the
RDS(ON) of the power MOSFET is:
RDS(ON) VSENSE(MAX) 1–DMAX
1+
2
•I
O(MAX) T
The VSENSE(MAX) term is typically 150mV at low duty
cycle, and is reduced to about 100mV at a duty cycle of
92% due to slope compensation, as shown in Figure 10.
The ρT term accounts for the temperature coeffi cient of
the RDS(ON) of the MOSFET, which is typically 0.4%/°C.
Figure 11 illustrates the variation of normalized RDS(ON)
over tempera
ture for a typical power MOSFET.
Another method of choosing which power MOSFET to
use is to check what the maximum output current is for a
given RDS(ON), since MOSFET on-resistances are available
in discrete values.
IO(MAX) =VSENSE(MAX) 1–DMAX
1+
2
•RDS(ON) T
It is worth noting that the 1 – DMAX relationship between
IO(MAX) and RDS(ON) can cause boost converters with a
wide input range to experience a dramatic range of maxi-
mum input and output current. This should be taken into
consideration in applications where it is important to limit
the maximum current drawn from the input supply.
Calculating Power MOSFET Switching and Conduction
Losses and Junction Temperatures
In order to calculate the junction temperature of the
power MOSFET, the power dissipated by the device must
be known. This power dissipation is a function of the
duty cycle, the load current and the junction temperature
itself (due to the positive temperature coeffi cient of its
RDS(ON)). As a result, some iterative calculation is normally
required to determine a reasonably accurate value. Since
the con
troller is using the MOSFET as both a switching
and a sensing element, care should be taken to ensure
that the converter is capable of delivering the required
load current over all operating conditions (line voltage
and temperature), and for the worst-case specifi cations
JUNCTION TEMPERATURE (°C)
–50
ρT NORMALIZED ON RESISTANCE
1.0
1.5
150
18711 F11
0.5
0050 100
2.0
Figure 11. Normalized RDS(ON) vs Temperature
LTC1871-1
16
18711fb
APPLICATIONS INFORMATION
for VSENSE(MAX) and the RDS(ON) of the MOSFET listed in
the manufacturers data sheet.
The power dissipated by the MOSFET in a boost converter
is:
P
FET
=I
O(MAX)
1–D
MAX
2
•R
DS(ON)
•D
MAX
T
+k•V
O1.85
I
O(MAX)
1–D
MAX
( )
•C
RSS
•f
The fi rst term in the equation above represents the I2R
losses in the device, and the second term, the switching
losses. The constant, k = 1.7, is an empirical factor inversely
related to the gate drive current and has the dimension
of 1/current.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
T
J = TA + PFET • RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(CA)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
Boost Converter: Output Diode Selection
To maximize effi ciency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
ID(PEAK) =IL(PEAK) =1+
2
IO(MAX)
1–DMAX
The power dissipated by the diode is:
P
D = IO(MAX) • VD
and the diode junction temperature is:
T
J = TA + PD • RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
Remember to keep the diode lead lengths short and to
observe proper switch-node layout (see Board Layout
Checklist) to avoid excessive ringing and increased
dissipation.
Boost Converter: Output Capacitor Selection
Contributions of ESR (equivalent series resistance), ESL
(equivalent series inductance) and the bulk capacitance
must be considered when choosing the correct component
for a given output ripple voltage. The effects of these three
parameters (ESR, ESL and bulk C) on the output voltage
ripple waveform are illustrated in Figure 12e for a typical
boost converter.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step and the charging/discharging
Δ
V.
For the purpose of simplicity we will choose 2% for the
maximum output ripple, to be divided equally between the
ESR step and the charging/discharging
Δ
V. This percent-
age ripple will change, depending on the requirements
of the application, and the equations provided below can
easily be modifi ed.
For a 1% contribution to the total ripple voltage, the ESR
of the output capacitor can be determined using the fol-
lowing equation:
ESRCOUT 0.01• VO
IIN(PEAK)
where:
IIN(PEAK)=1+
2
IO(MAX)
1–DMAX
For the bulk C component, which also contributes 1% to
the total ripple:
COUT IO(MAX)
0.01• VO f
LTC1871-1
17
18711fb
APPLICATIONS INFORMATION
For many designs it is possible to choose a single capacitor
type that satisfi es both the ESR and bulk C requirements
for the design. In certain demanding applications, however,
the ripple voltage can be improved signifi cantly by con-
necting two or more types of capacitors in parallel. For
example, using a low ESR ceramic capacitor can minimize
the ESR step, while an electrolytic capacitor can be used
to supply the required bulk C.
Once the output capacitor ESR and bulk capacitance have
been determined, the overall ripple voltage waveform
should be verifi ed on a dedicated PC board (see Board
Layout section for more information on component place-
ment). Lab breadboards generally suffer from excessive
series inductance (due to inter-component wiring), and
these parasitics can make the switching waveforms look
signifi cantly worse than they would be on a properly
designed PC board.
The output capacitor in a boost regulator experiences high
RMS ripple currents, as shown in Figure 12. The RMS
output capacitor ripple current is:
IRMS(COUT) IO(MAX) VO–V
IN(MIN)
VIN(MIN)
Note that the ripple current ratings from capacitor manu-
facturers are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be placed in parallel
to meet size or height requirements in the design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic, at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be placed in parallel in order to meet the ESR or
RMS current handling requirements of the application.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
tested for use in switching power supplies. An excellent
choice is AVX TPS series of surface mount tantalum. Also,
ceramic capacitors are now available with extremely low
ESR, ESL and high ripple current ratings.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical
than the output capacitor, due to the fact that the inductor
is in series with the input and the input current waveform
is continuous (see Figure 12b). The input voltage source
impedance determines the size of the input capacitor,
VIN
LD
SW
12a. Circuit Diagram
12b. Inductor and Input Currents
COUT
VOUT
RL
IIN
IL
12c. Switch Current
ISW
tON
12d. Diode and Output Currents
12e. Output Voltage Ripple Waveform
IO
ID
VOUT
(AC)
tOFF
ΔVESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
ΔVCOUT
Figure 12. Switching Waveforms for a Boost Converter
LTC1871-1
18
18711fb
APPLICATIONS INFORMATION
Table 1. Recommended Component Manufacturers
VENDOR COMPONENTS TELEPHONE WEB ADDRESS
AVX Capacitors (207) 282-5111 avxcorp.com
BH Electronics Inductors, Transformers (952) 894-9590 bhelectronics.com
Coilcraft Inductors (847) 639-6400 coilcraft.com
Coiltronics Inductors (407) 241-7876 coiltronics.com
Diodes, Inc Diodes (805) 446-4800 diodes.com
Fairchild MOSFETs (408) 822-2126 fairchildsemi.com
General Semiconductor Diodes (516) 847-3000 generalsemiconductor.com
International Rectifi er MOSFETs, Diodes (310) 322-3331 irf.com
IRC Sense Resistors (361) 992-7900 irctt.com
Kemet Tantalum Capacitors (408) 986-0424 kemet.com
Magnetics Inc Toroid Cores (800) 245-3984 mag-inc.com
Microsemi Diodes (617) 926-0404 microsemi.com
Murata-Erie Inductors, Capacitors (770) 436-1300 murata.co.jp
Nichicon Capacitors (847) 843-7500 nichicon.com
On Semiconductor Diodes (602) 244-6600 onsemi.com
Panasonic Capacitors (714) 373-7334 panasonic.com
Sanyo Capacitors (619) 661-6835 sanyo.co.jp
Sumida Inductors (847) 956-0667 sumida.com
Taiyo Yuden Capacitors (408) 573-4150 t-yuden.com
TDK Capacitors, Inductors (562) 596-1212 component.tdk.com
Thermalloy Heat Sinks (972) 243-4321 aavidthermalloy.com
Tokin Capacitors (408) 432-8020 nec-tokinamerica.com
Toko Inductors (847) 699-3430 tokoam.com
United Chemicon Capacitors (847) 696-2000 chemi-com.com
Vishay/Dale Resistors (605) 665-9301 vishay.com
Vishay/Siliconix MOSFETs (800) 554-5565 vishay.com
Vishay/Sprague Capacitors (207) 324-4140 vishay.com
Zetex Small-Signal Discretes (631) 543-7100 zetex.com
which is typically in the range of 10µF to 100µF. A low ESR
capacitor is recommended, although it is not as critical as
for the output capacitor.
The RMS input capacitor ripple current for a boost con-
verter is:
IRMS(CIN) =0.3 VIN(MIN)
L•f•D
MAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors!
Burst Mode Operation and Considerations
The choice of MOSFET RDS(ON) and inductor value also
determines the load current at which the LTC1871-1 en-
ters Burst Mode operation. When bursting, the controller
clamps the peak inductor current to approximately:
IBURST(PEAK) =30mV
RDS(ON)
which represents about 20% of the maximum 150mV
SENSE pin voltage. The corresponding average current
depends upon the amount of ripple current. Lower inductor
values (higher ΔIL) will reduce the load current at which
LTC1871-1
19
18711fb
APPLICATIONS INFORMATION
Burst Mode operations begins, since it is the peak current
that is being clamped.
The output voltage ripple can increase during Burst Mode
operation if ΔIL is substantially less than IBURST
. This can
occur if the input voltage is very low or if a very large
inductor is chosen. At high duty cycles, a skipped cycle
causes the inductor current to quickly decay to zero.
However, because ΔIL is small, it takes multiple cycles
for the current to ramp back up to IBURST(PEAK). Dur-
ing this inductor charging interval, the output capacitor
must supply the load current and a signifi cant droop in
the output voltage can occur. Generally, it is a good idea
to choose a value of inductor ΔIL between 25% and 40%
of IIN(MAX). The alternative is to either increase the value
of the output capacitor or disable Burst Mode operation
using the MODE/SYNC pin.
Burst Mode operation can be defeated by connecting the
MODE/SYNC pin to a high logic-level voltage (either with
a control input or by connecting this pin to INTVCC). In
this mode, the burst clamp is removed, and the chip can
operate at constant frequency from continuous conduction
mode (CCM) at full load, down into deep discontinuous
conduction mode (DCM) at light load. Prior to skipping
pulses at very light load (i.e., < 5% of full load), the
controller will operate with a minimum switch on-time
in DCM. Pulse skipping prevents a loss of control of
the output at very light loads and reduces output volt-
age ripple.
Effi ciency Considerations: How Much Does VDS
Sensing Help?
The effi ciency of a switching regulator is equal to the
output power divided by the input power (×100%).
Percent effi ciency can be expressed as:
% Effi ciency = 100% – (L1 + L2 + L3 + …),
where L1, L2, etc. are the individual loss components
as a percentage of the input power. It is often useful to
analyze individual losses to determine what is limiting
the effi ciency and which change would produce the most
improvement. Although all dissipative elements in the
circuit produce losses, four main sources usually account
for the majority of the losses in LTC1871-1 applica-
tion circuits:
1. The supply current into VIN. The VIN current is the
sum of the DC supply current IQ (given in the Electrical
Characteristics) and the MOSFET driver and control
currents. The DC supply current into the VIN pin is typi-
cally about 550µA and represents a small power loss
(much less than 1%) that increases with VIN. The driver
current results from switching the gate capacitance
of the power MOSFET; this current is typically much
larger than the DC current. Each time the MOSFET is
switched on and then off, a packet of gate charge QG
is transferred from INTVCC to ground. The resulting
dQ/dt is a current that must be supplied to the INTVCC
capacitor through the VIN pin by an external supply. If
the IC is operating in CCM:
I
Q(TOT) ≈ IQ = f • QG
P
IC = VIN • (IQ + f • QG)
2. Power MOSFET switching and conduction losses. The
technique of using the voltage drop across the power
MOSFET to close the current feedback loop was chosen
because of the increased effi ciency that results from
not having a sense resistor. The losses in the power
MOSFET are equal to:
P
FET
=I
O(MAX)
1–D
MAX
2
•R
DS(ON)
•D
MAX
T
+k•V
O1.85
I
O(MAX)
1–D
MAX
( )
•C
RSS
•f
The I2R power savings that result from not having a
discrete sense resistor can be calculated almost by
inspection.
To understand the magnitude of the improvement with
this VDS sensing technique, consider the 3.3V input,
5V output power supply shown in Figure 1. The maxi-
mum load current is 7A (10A peak) and the duty cycle
is 39%. Assuming a ripple current of 40%, the peak
inductor current is 13.8A and the average is 11.5A.
With a maximum sense voltage of about 140mV, the
sense resistor value would be 10m, and the power
LTC1871-1
20
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APPLICATIONS INFORMATION
Figure 13. Load Transient Response for a 3.3V Input,
5V Output Boost Converter Application, 0.7A to 7A Step
dissipated in this resistor would be 514mW at maxi-
mum output current. Assuming an effi ciency of 90%,
this sense resistor power dissipation represents 1.3%
of the overall input power. In other words, for this ap-
plication, the use of VDS sensing would increase the
effi ciency by approximately 1.3%.
For more details regarding the various terms in these
equations, please refer to the section Boost Converter:
Power MOSFET Selection.
3. The losses in the inductor are simply the DC input cur-
rent squared times the winding resistance. Expressing
this loss as a function of the output current yields:
P
R(WINDING) =IO(MAX)
1–DMAX
2
•RW
4. Losses in the boost diode. The power dissipation in the
boost diode is:
P
DIODE = IO(MAX) • VD
The boost diode can be a major source of power loss
in a boost converter. For the 3.3V input, 5V output at
7A example given above, a Schottky diode with a 0.4V
forward voltage would dissipate 2.8W, which represents
7% of the input power. Diode losses can become signifi -
cant at low output voltages where the forward voltage
is a signifi cant percentage of the output voltage.
5. Other losses, including CIN and CO ESR dissipation and
inductor core losses, generally account for less than
2% of the total additional loss.
Checking Transient Response
The regulator loop response can be verifi ed by looking at
the load transient response. Switching regulators generally
take several cycles to respond to an instantaneous step
in resistive load current. When the load step occurs, VO
immediately shifts by an amount equal to (ΔILOAD)(ESR),
and then CO begins to charge or discharge (depending on
the direction of the load step) as shown in Figure 13. The
regulator feedback loop acts on the resulting error amp
output signal to return VO to its steady-state value. During
this recovery time, VO can be monitored for overshoot or
ringing that would indicate a stability problem.
IOUT
2V/DIV
VOUT (AC)
100mV/DIV
100µs/DIV 18711 F13
VIN = 3.3V
VOUT = 5V
MODE/SYNC = INTVCC
(PULSE-SKIP MODE)
A second, more severe transient can occur when con-
necting loads with large (>1µF) supply bypass capacitors.
The discharged bypass capacitors are effectively put in
parallel with CO, causing a nearly instantaneous drop in
VO. No regulator can deliver enough current to prevent
this problem if the load switch resistance is low and it is
driven quickly. The only solution is to limit the rise time
of the switch drive in order to limit the inrush current
di/dt to the load.
Boost Converter Design Example
The design example given here will be for the circuit shown
in Figure 1. The input voltage is 3.3V, and the output is 5V
at a maximum load current of 7A (10A peak).
1. The duty cycle is:
D=VO+VD–V
IN
VO+VD
=5+0.4 3.3
5+0.4 =38.9%
2. Pulse-skip operation is chosen so the MODE/SYNC pin
is shorted to INTVCC.
3. The operating frequency is chosen to be 300kHz to
reduce the size of the inductor. From Figure 5, the
resistor from the FREQ pin to ground is 80k.
4. An inductor ripple current of 40% of the maximum load
current is chosen, so the peak input current (which is
also the minimum saturation current) is:
IIN(PEAK) =1+
2
IO(MAX)
1–DMAX
=1.2 7
1– 0.39 =13.8A
LTC1871-1
21
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APPLICATIONS INFORMATION
The inductor ripple current is:
IL=IO(MAX)
1–DMAX
=0.4 7
1– 0.39 =4.6A
And so the inductor value is:
L=VIN(MIN)
IL f •D
MAX =3.3V
4.6A 300kHz 0.39 =0.93μH
The component chosen is a 1µH inductor made by
Sumida (part number CEP125-H 1ROMH) which has
a saturation current of greater than 20A.
5. With the input voltage to the IC bootstrapped to the
output of the power supply (5V), a logic-level MOSFET
can be used. Because the duty cycle is 39%, the maxi-
mum SENSE pin threshold voltage is reduced from its
low duty cycle typical value of 150mV to approximately
140mV. Assuming a MOSFET junction temperature of
125°C, the room temperature MOSFET RDS(ON) should
be less than:
RDS(ON) VSENSE(MAX) 1–DMAX
1+
2
•I
O(MAX) T
=0.140V 1– 0.39
1+0.4
2
•7A1.5
=6.8m
The MOSFET used was the Fairchild FDS7760A, which
has a maximum RDS(ON) of 8m at 4.5V VGS, a BVDSS
of greater than 30V, and a gate charge of 37nC at 5V
VGS.
6. The diode for this design must handle a maximum
DC output current of 10A and be rated for a minimum
reverse voltage of VOUT
, or 5V. A 25A, 15V diode from
On Semiconductor (MBRB2515L) was chosen for its
high power dissipation capability.
7. The output capacitor usually consists of a high valued
bulk C connected in parallel with a lower valued, low
ESR ceramic. Based on a maximum output ripple voltage
of 1%, or 50mV, the bulk C needs to be greater than:
COUT IOUT(MAX)
0.01• VOUT f =
7A
0.01• 5V 300kHz =466μF
The RMS ripple current rating for this capacitor needs
to exceed:
IRMS(COUT) IO(MAX) VO–V
IN(MIN)
VIN(MIN)
=
7A 5V 3.3V
3.3V =5A
To satisfy this high RMS current demand, four 150µF
Panasonic capacitors (EEFUEOJ151R) are required.
In parallel with these bulk capacitors, two 22µF, low
ESR (X5R) Taiyo Yuden ceramic capacitors (JMK-
325BJ226MM) are added for HF noise reduction.
Check the output ripple with a single oscilloscope
probe connected directly across the output capacitor
terminals, where the HF switching currents fl ow.
8. The choice of an input capacitor for a boost converter
depends on the impedance of the source supply and the
amount of input ripple the converter will safely toler-
ate. For this particular design and lab setup a 100µF
Sanyo Poscap (6TPC 100M), in parallel with two 22µF
Taiyo Yuden ceramic capacitors (JMK325BJ226MM)
is required (the input and return lead lengths are kept
to a few inches, but the peak input current is close to
20A!). As with the output node, check the input ripple
with a single oscilloscope probe connected across the
input capacitor terminals.
LTC1871-1
22
18711fb
APPLICATIONS INFORMATION
Figure 14. LTC1871-1 Boost Converter Suggested Layout
Figure 15. LTC1871-1 Boost Converter Layout Diagram
LTC1871-1
M1
VIN
18711 F14
VOUT
SWITCH NODE IS ALSO
THE HEAT SPREADER
FOR L1, M1, D1
L1
RT
RCCC
R3
J1
CIN
COUT
CVCC
R1
R2
PSEUDO-KELVIN
SIGNAL GROUND
CONNECTION
TRUE REMOTE
OUTPUT SENSING
VIAS TO GROUND
PLANE
R4
PIN 1
COUT
BULK C LOW ESR CERAMIC
JUMPER
D1
RUN
ITH
FB
FREQ
MODE/
SYNC
SENSE
VIN
INTVCC
GATE
GND
LTC1871-1
+
R4
J1
10
9
8
7
6
1
2
3
4
5
CVCC
PSEUDO-KELVIN
GROUND CONNECTION
CIN
M1
D1
L1
VIN
GND
18711 F15
VOUT
SWITCH
NODE
COUT
RC
R1
RT
BOLD LINES INDICATE HIGH CURRENT PATHS
R2
CC
R3
+
PC Board Layout Checklist
1. In order to minimize switching noise and improve output
load regulation, the GND pin of the LTC1871-1 should
be connected directly to 1) the negative terminal of the
INTVCC decoupling capacitor, 2) the negative terminal
of the output decoupling capacitors, 3) the
source of
the power MOSFET or the bottom terminal of the sense
resistor, 4) the negative terminal of the input capacitor
and 5) at least one via to the ground plane immediately
adjacent to Pin 6. The ground trace on the top layer of
the PC board should be as wide and short as possible
to minimize series resistance and inductance.
LTC1871-1
23
18711fb
APPLICATIONS INFORMATION
2. Beware of ground loops in multiple layer PC boards.
Try to maintain one central ground node on the board
and use the input capacitor to avoid excess input ripple
for high output current power supplies. If the ground
plane is to be used for high DC currents, choose a path
away from the small-signal components.
3. Place the CVCC capacitor immediately adjacent to the
INTVCC and GND pins on the IC package. This capaci-
tor carries high di/dt MOSFET gate drive currents. A
low ESR and ESL 4.7µF ceramic capacitor works well
here.
4. The high di/dt loop from the bottom terminal of the
output capacitor, through the power MOSFET, through
the boost diode and back through the output capacitors
should be kept as tight as possible to reduce inductive
ringing. Excess inductance can cause increased stress
on the power MOSFET and increase HF noise on the
output. If low ESR ceramic capacitors are used on the
output to reduce output noise, place these capacitors
close to the boost diode in order to keep the series
inductance to a minimum.
5. Check the stress on the power MOSFET by measuring
its drain-to-source voltage directly across the device
terminals (reference the ground of a single scope probe
directly to the source pad on the PC board). Beware
of inductive ringing which can exceed the maximum
specifi ed voltage rating of the MOSFET. If this ringing
cannot be avoided and exceeds the maximum rating
of the device, either choose a higher voltage device
or specify an avalanche-rated power MOSFET. Not all
MOSFETs are created equal (some are more equal than
others).
6. Place the small-signal components away from high
frequency switching nodes. In the layout shown in
Figure 14, all of the small-signal components have
been placed on one side of the IC and all of the power
components have been placed on the other. This also
allows the use of a pseudo-Kelvin connection for the
signal ground, where high di/dt gate driver currents
ow out of the IC ground pin in one direction (to the
bottom plate of the INTVCC decoupling capacitor) and
small-signal currents fl ow in the other direction.
7. If a sense resistor is used in the source of the power
MOSFET, minimize the capacitance between the SENSE
pin trace and any high frequency switching nodes. The
LTC1871-1 contains an internal leading edge blanking
time of approximately 180ns, which should be adequate
for most applications.
8. For optimum load regulation and true remote sensing,
the top of the output resistor divider should connect
independently to the top of the output capacitor (Kelvin
connection), staying away from any high dV/dt traces.
Place the divider resistors near the LTC1871-1 in order
to keep the high impedance FB node short.
9. For applications with multiple switching power convert-
ers connected to the same input supply, make sure
that the input fi lter capacitor for the LTC1871-1 is not
shared with other converters. AC input current from
another converter could cause substantial input volt-
age ripple, and this could interfere with the operation
of the LTC1871-1. A few inches of PC trace or wire
(L ≈ 100nH) between the CIN of the LTC1871-1 and the
actual source VIN should be suffi cient to prevent current
sharing problems.
Figures 16. SEPIC Topology and Current Flow
+
+
+
SW L2 COUT RL
VOUT
VIN
C1 D1
L1
16a. SEPIC Topology
+
+
+
RL
VOUT
VIN
D1
16c. Current Flow During Switch Off-Time
+
+
+
RL
VOUT
VIN
VIN
VIN
16b. Current Flow During Switch On-Time
LTC1871-1
24
18711fb
APPLICATIONS INFORMATION
SEPIC Converter Applications
The LTC1871-1 is also well suited to SEPIC (single-ended
primary inductance converter) converter applications. The
SEPIC converter shown in Figure 16 uses two inductors.
The advantage of the SEPIC converter is the input voltage
may be higher or lower than the output voltage, and the
output is short-circuit protected.
The fi rst inductor, L1, together with the main switch,
resembles a boost converter. The second inductor, L2,
together with the output diode D1, resembles a fl yback or
buck-boost converter. The two inductors L1 and L2 can be
independent but can also be wound on the same core since
identical voltages are applied to L1 and L2 throughout the
switching cycle. By making L1 = L2 and winding them on
the same core the input ripple is reduced along with cost
and size. All of the SEPIC applications information that
follows assumes L1 = L2 = L.
SEPIC Converter: Duty Cycle Considerations
For a SEPIC converter operating in a continuous conduction
mode (CCM), the duty cycle of the main switch is:
D=VO+VD
VIN +VO+VD
where VD is the forward voltage of the diode. For convert-
ers where the input voltage is close to the output voltage
the duty cycle is near 50%.
The maximum output voltage for a SEPIC converter is:
VO(MAX) =VIN +VD
()
DMAX
1–DMAX
–V
D
1
1–DMAX
The maximum duty cycle of the LTC1871-1 is typically
92%.
SEPIC Converter: The Peak and Average
Input Currents
The control circuit in the LTC1871-1 is measuring the input
current (either using the RDS(ON) of the power MOSFET
or by means of a sense resistor in the MOSFET source),
so the output current needs to be refl ected back to the
input in order to dimension the power MOSFET properly.
Based on the fact that, ideally, the output power is equal
to the input power, the maximum input current for a SEPIC
converter is:
I
IN(MAX)
=I
O(MAX)
D
MAX
1–D
MAX
The peak input current is:
I
IN(PEAK)
=1+
2
•I
O(MAX)
D
MAX
1–D
MAX
The maximum duty cycle, DMAX, should be calculated at
minimum VIN.
Figure 17. SEPIC Converter Switching Waveforms
17a. Input Inductor Current
IIN
IL1 SW
ON
SW
OFF
17b. Output Inductor Current
IO
IL2
17c. DC Coupling Capacitor Current
IO
IIN
IC1
17e. Output Ripple Voltage
VOUT
(AC)
ΔVESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
ΔVCOUT
17d. Diode Current
IO
ID1
LTC1871-1
25
18711fb
APPLICATIONS INFORMATION
The constant ‘χ’ represents the fraction of ripple current in
the inductor relative to its maximum value. For example, if
30% ripple current is chosen, then χ = 0.30 and the peak
current is 15% greater than the average.
It is worth noting here that SEPIC converters that operate
at high duty cycles (i.e., that develop a high output volt-
age from a low input voltage) can have very high input
currents, relative to the output current. Be sure to check
that the maximum load current will not overload the input
supply.
SEPIC Converter: Inductor Selection
For most SEPIC applications the equal inductor values
will fall in the range of 10µH to 100µH. Higher values will
reduce the input ripple voltage and reduce the core loss.
Lower inductor values are chosen to reduce physical size
and improve transient response.
Like the boost converter, the input current of the SEPIC
converter is calculated at full load current and minimum
input voltage. The peak inductor current can be signifi cantly
higher than the output current, especially with smaller in-
ductors and lighter loads. The following formulas assume
CCM operation and calculate the maximum peak inductor
currents at minimum VIN:
IL1(PEAK) =1+
2
•I
O(MAX) VO+VD
VIN(MIN)
IL2(PEAK) =1+
2
•I
O(MAX) VIN(MIN) +VD
VIN(MIN)
The ripple current in the inductor is typically 20% to 40%
(i.e., a range of ‘χ’ from 0.20 to 0.40) of the maximum
average input current occurring at VIN(MIN) and IO(MAX) and
ΔIL1 = ΔIL2. Expressing this ripple current as a function of
the output current results in the following equations for
calculating the inductor value:
L=VIN(MIN)
IL f •D
MAX
where:
IL=•I
O(MAX) DMAX
1–DMAX
By making L1 = L2 and winding them on the same core,
the value of inductance in the equation above is replace
by 2L due to mutual inductance. Doing this maintains the
same ripple current and energy storage in the inductors. For
example, a Coiltronix CTX10-4 is a 10µH inductor with two
windings. With the windings in parallel, 10µH inductance is
obtained with a current rating of 4A (the number of turns
hasn’t changed, but the wire diameter has doubled). Split-
ting the two windings creates two 10µH inductors with a
current rating of 2A each. Therefore, substituting 2L yields
the following equation for coupled inductors:
L1=L2=VIN(MIN)
2•IL f •D
MAX
Specify the maximum inductor current to safely handle
IL(PK) specifi ed in the equation above.
The saturation
current rating for the inductor should be checked at the
minimum input voltage (which results in the highest
inductor current) and maximum output current.
SEPIC Converter: Power MOSFET Selection
The power MOSFET serves two purposes in the LTC1871-1:
it represents the main switching element in the power path,
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFETs
thermal resistances (RTH(JC) and RTH(JA)).
The gate drive voltage is set by the 5.2V INTVCC low dropout
regulator. Consequently, logic-level threshold MOSFETs
should be used in most LTC1871-1 applications. If low
input voltage operation is expected (e.g., supplying power
from a lithium-ion battery), then sublogic-level threshold
MOSFETs should be used.
The maximum voltage that the MOSFET switch must
sustain during the off-time in a SEPIC converter is equal
to the sum of the input and output voltages (VO + VIN).
As a result, careful attention must be paid to the BVDSS
specifi cations for the MOSFETs relative to the maximum
actual switch voltage in the application. Many logic-level
LTC1871-1
26
18711fb
APPLICATIONS INFORMATION
devices are limited to 30V or less. Check the switching
waveforms directly across the drain and source terminals
of the power MOSFET to ensure the VDS remains below
the maximum rating for the device.
During the MOSFETs on-time, the control circuit limits
the maximum voltage drop across the power MOSFET to
about 150mV (at low duty cycle). The peak inductor current
is therefore limited to 150mV/RDS(ON). The relationship
between the maximum load current, duty cycle and the
RDS(ON) of the power MOSFET is:
R
DS(ON)
V
SENSE(MAX)
I
O(MAX)
1
1+
2
T
1
V
O
+V
D
V
IN(MIN)
+1
The VSENSE(MAX) term is typically 150mV at low duty cycle
and is reduced to about 100mV at a duty cycle of 92% due
to slope compensation, as shown in Figure 8. The constant
χ’ in the denominator represents the ripple current in the
inductors relative to their maximum current. For example,
if 30% ripple current is chosen, then χ = 0.30. The ρT term
accounts for the temperature coeffi cient of the RDS(ON) of
the MOSFET, which is typically 0.4%/°C. Figure 9 illustrates
the variation of normalized RDS(ON) over temperature for
a typical power MOSFET.
Another method of choosing which power MOSFET to
use is to check what the maximum output current is for a
given RDS(ON) since MOSFET on-resistances are available
in discrete values.
IO(MAX) VSENSE(MAX)
RDS(ON)
1
1+
2
T
1
VO+VD
VIN(MIN)
+1
Calculating Power MOSFET Switching and Conduction
Losses and Junction Temperatures
In order to calculate the junction temperature of the
power MOSFET, the power dissipated by the device must
be known. This power dissipation is a function of the
duty cycle, the load current and the junction temperature
itself. As a result, some iterative calculation is normally
required to determine a reasonably accurate value. Since
the controller is using the MOSFET as both a switching
and a sensing element, care should be taken to ensure
that the converter is capable of delivering the required
load current over all operating conditions (load, line and
temperature) and for the worst-case specifi cations for
VSENSE(MAX) and the RDS(ON) of the MOSFET listed in the
manufacturers data sheet.
The power dissipated by the MOSFET in a SEPIC converter
is:
P
FET =IO(MAX) DMAX
1–DMAX
2
•RDS(ON) •D
MAX T
+k• V
IN(MIN) +VO
()
1.85 •I
O(MAX) DMAX
1–DMAX
•C
RSS •f
The fi rst term in the equation above represents the I2R
losses in the device and the second term, the switching
losses. The constant k = 1.7 is an empirical factor inversely
related to the gate drive current and has the dimension
of 1/current.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
T
J = TA + PFET •RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
This value of TJ can then be used to check the original
assumption for the junction temperature in the iterative
calculation process.
SEPIC Converter: Output Diode Selection
To maximize effi ciency, a fast-switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a SEPIC converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to VIN(MAX) + VO. The average
forward current in normal operation is equal to the output
current, and the peak current is equal to:
I
D(PEAK)
=1+
2
•I
O(MAX)
V
O
+V
D
V
IN(MIN)
+1
The power dissipated by the diode is:
P
D = IO(MAX) • VD
LTC1871-1
27
18711fb
APPLICATIONS INFORMATION
and the diode junction temperature is:
T
J = TA + PD • RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
SEPIC Converter: Output Capacitor Selection
Because of the improved performance of todays electro-
lytic, tantalum and ceramic capacitors, engineers need
to consider the contributions of ESR (equivalent series
resistance), ESL (equivalent series inductance) and the
bulk capacitance when choosing the correct component
for a given output ripple voltage. The effects of these three
parameters (ESR, ESL, and bulk C) on the output voltage
ripple waveform are illustrated in Figure 17 for a typical
coupled-inductor SEPIC converter.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step and the charging/discharging ΔV.
For the purpose of simplicity we will choose 2% for the
maximum output ripple, to be divided equally between the
ESR step and the charging/discharging ΔV. This percentage
ripple will change, depending on the requirements of the
application, and the equations provided below can easily
be modifi ed.
For a 1% contribution to the total ripple voltage, the ESR
of the output capacitor can be determined using the fol-
lowing equation:
ESRCOUT 0.01• VO
IIN(PEAK)
where:
ID(PEAK) =1+
2
•I
O(MAX) VO+VD
VIN(MIN)
+1
For the bulk C component, which also contributes 1% to
the total ripple:
COUT IO(MAX)
0.01• VO f
For many designs it is possible to choose a single capacitor
type that satisfi es both the ESR and bulk C requirements
for the design. In certain demanding applications, however,
the ripple voltage can be improved signifi cantly by con-
necting two or more types of capacitors in parallel. For
example, using a low ESR ceramic capacitor can minimize
the ESR step, while an electrolytic or tantalum capacitor
can be used to supply the required bulk C.
Once the output capacitor ESR and bulk capacitance have
been determined, the overall ripple voltage waveform
should be verifi ed on a dedicated PC board (see Board
Layout section for more information on component place-
ment). Lab breadboards generally suffer from excessive
series inductance (due to inter-component wiring), and
these parasitics can make the switching waveforms look
signifi cantly worse than they would be on a properly
designed PC board.
The output capacitor in a SEPIC regulator experiences
high RMS ripple currents, as shown in Figure 17. The
RMS output capacitor ripple current is:
IRMS(C1) =IO(MAX) VO+VD
VIN(MIN)
Note that the ripple current ratings from capacitor manu-
facturers are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be placed in parallel
to meet size or height requirements in the design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic, at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be placed in parallel in order to meet the ESR or
RMS current handling requirements of the application.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
tested for use in switching power supplies. An excellent
LTC1871-1
28
18711fb
APPLICATIONS INFORMATION
choice is AVX TPS series of surface mount tantalum. Also,
ceramic capacitors are now available with extremely low
ESR, ESL and high ripple current ratings.
SEPIC Converter: Input Capacitor Selection
The input capacitor of a SEPIC converter is less critical
than the output capacitor due to the fact that an inductor
is in series with the input and the input current waveform
is triangular in shape. The input voltage source impedance
determines the size of the input capacitor which is typi-
cally in the range of 10µF to 100µF. A low ESR capacitor
is recommended, although it is not as critical as for the
output capacitor.
The RMS input capacitor ripple current for a SEPIC con-
verter is:
IRMS(CIN) =1
12 IL
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors!
SEPIC Converter: Selecting the DC Coupling Capacitor
The coupling capacitor C1 in Figure 16 sees nearly a rectan-
gular current waveform as shown in Figure 17. During the
switch off-time the current through C1 is IO(VO/VIN) while
approximately –IO ows during the on-time. This current
waveform creates a triangular ripple voltage on C1:
VC1(PP) =IO(MAX)
C1 fVO
VIN +VO+VD
The maximum voltage on C1 is then:
VC1(MAX) =VIN +
VC1(PP)
2
which is typically close to VIN(MAX). The ripple current
through C1 is:
IRMS(C1) =IO(MAX) VO+VD
VIN(MIN)
The value chosen for the DC coupling capacitor normally
starts with the minimum value that will satisfy 1) the RMS
current requirement and 2) the peak voltage requirement
(typically close to VIN). Low ESR ceramic and tantalum
capacitors work well here.
SEPIC Converter Design Example
The design example given here will be for the circuit shown
in Figure 18. The input voltage is 5V to 15V and the output
is 12V at a maximum load current of 1.5A (2A peak).
1. The duty cycle range is:
D=VO+VD
VIN +VO+VD
=45.5% to 71.4%
2. The operating mode chosen is pulse skipping, so the
MODE/SYNC pin is shorted to INTVCC.
3. The operating frequency is chosen to be 300kHz to
reduce the size of the inductors; the resistor from the
FREQ pin to ground is 80k.
4. An inductor ripple current of 40% is chosen, so the peak
input current (which is also the minimum saturation
current) is:
IL1(PEAK) =1+
2
•I
O(MAX) VO+VD
VIN(MIN)
=1+0.4
2
1.5 12 +0.5
5=4.5A
The inductor ripple current is:
IL=•I
O(MAX) DMAX
1–DMAX
=0.4 1.5 0.714
1– 0.714 =1.5A
LTC1871-1
29
18711fb
APPLICATIONS INFORMATION
And so the inductor value is:
L=VIN(MIN)
2•IL f •D
MAX =5
2 1.5 300k 0.714 =4μH
T
he component chosen is a BH Electronics BH510-
1007, which has a saturation current of 8A.
5. With an minimum input voltage of 5V, only logic-level
power MOSFETs should be considered. Because the
maximum duty cycle is 71.4%, the maximum SENSE
pin threshold voltage is reduced from its low duty
cycle typical value of 150mV to approximately 120mV.
Assuming a MOSFET junction temperature of 125°C,
the room temperature MOSFET RDS(ON) should be less
than:
RDS(ON) VSENSE(MAX)
IO(MAX)
1
1+
2
T
1
VO+VD
VIN(MIN)
+1
=0.12
1.5 1
1.2 1.5 1
12.5
5
+1
=12.7m
For a SEPIC converter, the switch BVDSS rating must be
greater than VIN(MAX) + VO, or 27V. This comes close to
an IRF7811W, which is rated to 30V, and has a maximum
room temperature RDS(ON) of 12m at VGS = 4.5V.
Figure 18a. 4.5V to 15V Input, 12V/2A Output SEPIC Converter
Figure 18b. SEPIC Effi ciency vs Output Current
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT
80.6k
1%
R1
12.1k
1%
R2
105k
1%
R3
1M
CVCC
4.7µF
X5R
CIN
47µF
M1
CIN, COUT1: KEMET T495X476K020AS
CDC, COUT2: TAIYO YUDEN TMK432BJ106MM
D1: INTERNATIONAL RECTIFIER 30BQ040
D1
L1*
L2*
RC
33k
CC1
6.8nF
CC2
47pF
COUT1
47µF
20V
×2
VIN
4.5V to 15V
VOUT
12V
1.5A
(2A PEAK)
GND
18711 F018a
+
COUT2
10µF
25V
X5R
×2
L1, L2: BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)
M1: INTERNATIONAL RECTIFIER IRF7811W
CDC
10µF
25V
X5R
+
OUTPUT CURRENT (A)
50
EFFICIENCY (%)
55
60
90
85
80
75
70
65
95
0.001 0.1 1 10
18711 F18b
45
0.01
100
VIN = 4.5V
VIN = 15V
VIN = 12V
VO = 12V
MODE = INTVCC
LTC1871-1
30
18711fb
APPLICATIONS INFORMATION
Figure 19. LTC1871-1 SEPIC Converter Load Step Response
VOUT (AC)
200mV/DIV
IOUT
0.5A/DIV
50µs/DIV 18711 F19
VIN = 15V
VOUT = 12V
VOUT (AC)
200mV/DIV
IOUT
0.5A/DIV
50µs/DIV
VIN = 4.5V
VOUT = 12V
6. The diode for this design must handle a maximum
DC output current of 2A and be rated for a minimum
reverse voltage of VIN + VOUT
, or 27V. A 3A, 40V diode
from International Rectifi er (30BQ040) is chosen for its
small size, relatively low forward drop and acceptable
reverse leakage at high temp.
7. The output capacitor usually consists of a high valued
bulk C connected in parallel with a lower valued, low ESR
ceramic. Based on a maximum output ripple voltage of
1%, or 120mV, the bulk C needs to be greater than:
COUT IOUT(MAX)
0.01• VOUT f =
1.5A
0.01• 12V 300kHz =41μF
The RMS ripple current rating for this capacitor needs
to exceed:
IRMS(COUT) IO(MAX) VO
VIN(MIN)
=
1.5A 12V
5V=2.3A
To satisfy this high RMS current demand, two 47µF
Kemet capacitors (T495X476K020AS) are required. As a
result, the output ripple voltage is a low 50mV to 60mV.
In parallel with these tantalums, two 10µF, low ESR (X5R)
Taiyo Yuden ceramic capacitors (TMK432BJ106MM) are
added for HF noise reduction. Check the output ripple
with a single oscilloscope probe connected directly
across the output capacitor terminals, where the HF
switching currents fl ow.
8. The choice of an input capacitor for a SEPIC converter
depends on the impedance of the source supply and the
amount of input ripple the converter will safely toler-
ate. For this particular design and lab setup, a single
47µF Kemet tantalum capacitor (T495X476K020AS) is
adequate. As with the output node, check the input ripple
with a single oscilloscope probe connected across the
input capacitor terminals. If any HF switching noise is
observed it is a good idea to decouple the input with
a low ESR, X5R ceramic capacitor as close to the VIN
and GND pins as possible.
9. The DC coupling capacitor in a SEPIC converter is cho-
sen based on its RMS current requirement and must be
rated for a minimum voltage of VIN plus the AC ripple
voltage. Start with the minimum value which satisfi es
the RMS current requirement and then check the ripple
voltage to ensure that it doesn’t exceed the DC rating.
IRMS(CI) IO(MAX) VO+VD
VIN(MIN)
=1.5A 12V +0.5V
5V =2.4A
For this design a single 10µF, low ESR (X5R) Taiyo
Yuden ceramic capacitor (TMK432BJ106MM) is
adequate.
LTC1871-1
31
18711fb
TYPICAL APPLICATIONS
2.5V to 3.3V Input, 5V/2A Output Boost Converter
Output Effi ciency at 2.5V and 3.3V Input
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT
80.6k
1%
R1
12.1k
1%
R2
37.4k
1%
CVCC
4.7µF
X5R
CIN
47µF
6.3V
M1
CIN: SANYO POSCAP 6TPA47M
COUT1: SANYO POSCAP 6TPB150M
COUT2: TAIYO YUDEN JMK316BJ106ML
CVCC: TAIYO YUDEN LMK316BJ475ML
D1
L1
1.8µH
RC
22k
CC1
6.8nF
CC2
47pF
COUT1
150µF
6.3V
×2
VIN
2.5V to 3.3V
VOUT
5V
2A
GND
18711 TA01a
+
COUT2
10µF
6.3V
X5R
×2
D1: INTERNATIONAL RECTIFIER 30BQ015
L1: TOKO DS104C2 B952AS-1R8N
M1: SILICONIX/VISHAY Si9426
+
OUTPUT CURRENT (A)
65
EFFICIENCY (%)
95
100
60
55
90
75
85
80
70
0.001 0.1 1 10
18711 TA01b
50
0.01
LTC1871-1
32
18711fb
TYPICAL APPLICATIONS
18V to 27V Input, 28V Output, 400W 2-Phase, Low Ripple, Synchronized RF Base Station Power Supply (Boost)
5V to 12V Input, ±12V/0.2A Output SEPIC Converter with Undervoltage Lockout
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT1
150k
5%
RS1
0.007Ω
1W
EXT CLOCK
INPUT (200kHz)
R2
8.45k
1%
CVCC1
4.7µF
X5R
CIN2
2.2µF
35V
X5R
M1
D1
L2
5.6µH
L5*
0.3µH
CC1
47pF
CFB1
47pF
VIN
18V to 27V
GND
VOUT
28V
14A
18711 TA04
COUT1
2.2µF
35V
X5R
×3
COUT2
330µF
50V
CIN1: SANYO 50MV330AX
CIN2, 3: TAIYO YUDEN GMK325BJ225MN
COUT2, 4, 5: SANYO 50MV330AX
COUT1, 3, 6: TAIYO YUDEN GMK325BJ225MN
CVCC1, 2: TAIYO YUDEN LMK316BJ475ML
L1 TO L4: SUMIDA CEP125-5R6MC-HD
L5: SUMIDA CEP125-0R3NC-ND
D1, D2: ON SEMICONDUCTOR MBR2045CT
M1, M2: INTERNATIONAL RECTIFIER IRLZ44NS
R1
93.1k
1%
L1
5.6µH
*L5, COUT5 AND
COUT6 ARE AN
OPTIONAL SECONDARY
FILTER TO REDUCE
OUTPUT RIPPLE FROM
<500mVP-P TO <100mVP-P
+
COUT5*
330µF
50V
×4COUT6*
2.2µF
35V
X5R
CIN1
330µF
50V
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT2
150k
5%
R3
12.1k
1%
RC
22k
R4
261k
1%
CVCC2
4.7µF
X5R
CIN3
2.2µF
35V
X5R
M2
D2
L4
5.6µH
CC2
47pF
CC3
6.8nF
CFB2
47pF
COUT3
2.2µF
35V
X5R
×3
COUT4
330µF
50V
L3
5.6µH
+
+
RS2
0.007Ω
1W
+
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT
60.4k
1% RS
0.02Ω
R4
127Ω
1%
R2
54.9k
1%
R3
1.10k
1%
CVCC
4.7µF
10V
X5R
CIN1
1µF
16V
X5R
M1
D1
L1*
RC
22k
CC1
6.8nF
CC2
100pF
CDC1
4.7µF
16V
X5R
VIN
5V to 12V
VOUT1
12V
0.4A
GND
VOUT2
–12V
0.4A
18711 TA03
COUT1
4.7µF
16V
X5R
×3
COUT2
4.7µF
16V
X5R
×3
D1, D2: MBS120T3
L1 TO L3: COILTRONICS VP1-0076 (*COUPLED INDUCTORS)
M1: SILICONIX/VISHAY Si4840
R1
127k
1%
CIN2
47µF
16V
AVX
CDC2
4.7µF
16V
X5R
+
L2*
L3*
D2
NOTE:
1. VIN UVLO+ = 4.47V
VIN UVLO = 4.14V
LTC1871-1
33
18711fb
TYPICAL APPLICATIONS
4.5V to 28V Input, 5V/2A Output SEPIC Converter with Undervoltage Lockout and Soft-Start
Soft-Start Load Step Response at VIN = 4.5V
Load Step Response at VIN = 28V
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT
162k
1%
C2
1µF
X5R NOTES:
1. VIN UVLO+ = 4.17V
VIN UVLO = 3.86V
2. SOFT-START dVOUT/dt = 5V/6ms
R4
49.9k
1%
R3
154k
1%
Q1
R1
115k
1%
R2
54.9k
1%
CVCC
4.7µF
10V
X5R
CIN1
2.2µF
35V
X5R
CIN2
22µF
35V
M1
D1
L1*
L2*
RC
12k
CC1
8.2nF
C1
4.7nF
CC2
47pF
R6
750Ω
R5
100Ω
COUT1
330µF
6.3V
VIN
4.5V to 28V
VOUT
5V
2A
(3A TO 4A PEAK)
GND
18711 TA02a
+
COUT2
22µF
6.3V
X5R
CDC
2.2µF
25V
X5R
×3
+
CIN1, CDC: TAIYO YUDEN GMK325BJ225MN
CIN2: AVX TPSE226M035R0300
COUT1: SANYO 6TPB330M
COUT2: TAIYO YUDEN JMK325BJ226MN
CVCC: LMK316BJ475ML
D1: INTERNATIONAL RECTIFIER 30BQ040
L1, L2: BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)
M1: SILICONIX/VISHAY Si4840
Q1: PHILIPS BC847BF
VOUT
1V/DIV
1ms/DIV 18711 TA02b
VOUT
100mV/DIV
(AC)
2.2A
0.5A
250µs/DIV 18711 TA02c
IOUT
1A/DIV
(DC)
VOUT
100mV/DIV
(AC)
2.2A
0.5A
250µs/DIV 18711 TA02d
IOUT
1A/DIV
(DC)
LTC1871-1
34
18711fb
TYPICAL APPLICATIONS
5V to 15V Input, –5V/5A Output Positive-to-Negative Converter with Undervoltage Lockout and Level-Shifted Feedback
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
1
2
3
4
5
10
9
8
7
6
LTC1871-1
RT
80.6k
1%
R1
154k
1%
R2
68.1k
1%
C1
1nF
CVCC
4.7µF
10V
X5R
C2
10nF
R4
10k
1% R5
40.2k
1%
R3
10k
1%
4
3
2
6
1
CIN
47µF
16V
X5R
M1
CIN: TDK C5750X5R1C476M
CDC: TDK C5750X7R1E226M
COUT: TDK C5750X5R0J107M
CVCC: TAIYO YUDEN LMK316BJ475ML
D1
L1* L2*
RC
10k
CC1
10nF
CC2
330pF
VIN
5V to 15V
VOUT
–5V
5A
GND
18711 TA05
COUT
100µF
6.3V
X5R
×2
D1: ON SEMICONDUCTOR MBRB2035CT
L1, L2: COILTRONICS VP5-0053 (*3 WINDINGS IN PARALLEL
FOR THE PRIMARY, 3 IN PARALLEL FOR SECONDARY)
M1: INTERNATIONAL RECTIFIER IRF7822
CDC
22µF
25V
X7R
+
LT1783
LTC1871-1
35
18711fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
PACKAGE DESCRIPTION
MS Package
10-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1661 Rev E)
MSOP (MS) 0307 REV E
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
12345
4.90 ± 0.152
(.193 ± .006)
0.497 ± 0.076
(.0196 ± .003)
REF
8910 76
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ± 0.127
(.035 ± .005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ± 0.038
(.0120 ± .0015)
TYP
0.50
(.0197)
BSC
0.1016 ± 0.0508
(.004 ± .002)
LTC1871-1
36
18711fb
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007
LT 0108 REV B • PRINTED IN USA
RELATED PARTS
TYPICAL APPLICATION
High Power SLIC Supply with Undervoltage Lockout
(Also See the LTC3704 Data Sheet)
+
RUN
ITH
FB
FREQ
MODE/SYNC
SENSE
VIN
INTVCC
GATE
GND
LTC1871-1
RT
120k f = 200kHz
*COILTRONICS VP5-0155
(PRIMARY = 3 WINDINGS IN PARALLEL)
C1
4.7µF
X5R
+CIN
220µF
16V
TPS
C3
10µF
25V
X5R
IRL2910
RS
0.012Ω
C8
0.1µF
D3
10BQ060
5
VIN
7V TO 12V
T1*
1, 2, 3
RC
82k
CC1
1nF
CC2
100pF
CR
1nF
R1
49.9k
1%
R2
150k
1%
D4
10BQ060
6
D2
10BQ060
4
C4
10µF
25V
X5R
COUT
3.3µF
100V
GND
VOUT1
–24V
200mA
VOUT2
–72V
200mA
C5
10µF
25V
X5R
4
3
18711 TA06
2
6
1
10k
RF1
10k
1%
RF2
196k
1%
C2
4.7µF
50V
X5R
+
LT1783
PART NUMBER DESCRIPTION COMMENTS
LT
®
1619 Current Mode PWM Controller 300kHz Fixed Frequency, Boost, SEPIC, Flyback Topology
LTC1624 Current Mode DC/DC Controller SO-8; 300kHz Operating Frequency; Buck, Boost, SEPIC Design;
VIN Up to 36V
LTC1700 No RSENSE Synchronous Step-Up Controller Up to 95% Effi ciency, Operation as Low as 0.9V Input
LTC1871 Wide Input Range Controller No RSENSE, 5V Gate Drive, Current Mode Control
LTC1871-7 Wide Input Range Controller No RSENSE, 7V Gate Drive, Current Mode Control
LTC1872 SOT-23 Boost Controller Delivers Up to 5A, 550kHz Fixed Frequency, Current Mode
LT1930 1.2MHz, SOT-23 Boost Converter Up to 34V Output, 2.6V ≤ VIN ≤ 16V, Miniature Design
LT1931 Inverting 1.2MHz, SOT-23 Converter Positive-to-Negative DC/DC Conversion, Miniature Design
LTC3401/LTC3402 1A/2A 3MHz Synchronous Boost Converters Up to 97% Effi ciency, Very Small Solution, 0.5V ≤ VIN ≤ 5V
LTC3704 Positive-to-Negative DC/DC Controller No RSENSE, Current Mode Control, 50kHz to 1MHz
LT3782 2-Phase Step-Up DC/DC Controller 6V ≤ VIN ≤ 40V; 4A Gate Drive, 150kHz to 500kHz