REV. D
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
AMP01*
Low Noise, Precision
Instrumentation Amplifier
GENERAL DESCRIPTION
The AMP01 is a monolithic instrumentation amplifier designed
for high-precision data acquisition and instrumentation applica-
tions. The design combines the conventional features of an
instrumentation amplifier with a high current output stage. The
output remains stable with high capacitance loads (1 µF), a
unique ability for an instrumentation amplifier. Consequently,
the AMP01 can amplify low level signals for transmission
through long cables without requiring an output buffer. The output
stage may be configured as a voltage or current generator.
Input offset voltage is very low (20 µV), which generally elimi-
nates the external null potentiometer. Temperature changes
have minimal effect on offset; TCV
IOS
is typically 0.15 µV/°C.
Excellent low-frequency noise performance is achieved with a
minimal compromise on input protection. Bias current is very
low, less than 10 nA over the military temperature range. High
common-mode rejection of 130 dB, 16-bit linearity at a gain of
1000, and 50 mA peak output current are achievable simulta-
neously. This combination takes the instrumentation amplifier
one step further towards the ideal amplifier.
AC performance complements the superb dc specifications. The
AMP01 slews at 4.5 V/µs into capacitive loads of up to 15 nF,
settles in 50 µs to 0.01% at a gain of 1000, and boasts a healthy
26 MHz gain-bandwidth product. These features make the
AMP01 ideal for high speed data acquisition systems.
Gain is set by the ratio of two external resistors over a range of
0.1 to 10,000. A very low gain temperature coefficient of
10 ppm/°C is achievable over the whole gain range. Output
voltage swing is guaranteed with three load resistances; 50 Ω,
500 Ω, and 2 kΩ. Loaded with 500 Ω, the output delivers
±13.0 V minimum. A thermal shutdown circuit prevents de-
struction of the output transistors during overload conditions.
The AMP01 can also be configured as a high performance op-
erational amplifier. In many applications, the AMP01 can be
used in place of op amp/power-buffer combinations.
PIN CONFIGURATIONS
18-Lead Cerdip
TOP VIEW
(Not to Scale)
18
17
16
15
14
13
12
11
10
1
2
3
4
5
6
7
8
9AMP01
OUTPUT
REFERENCE
RG
RG
–IN
VOOS NULL
SENSE
TEST PIN*
VOOS NULL
–VOP
V–
+IN
VIOS NULL
VIOS NULL
RS
V+
+VOP
RS
*MAKE NO ELECTRICAL CONNECTION
AMP01 BTC/883
28-Terminal LCC
NC = NO CONNECT
TOP VIEW
(Not to Scale)
28 27123426
25
21
22
23
24
19
20
5
6
7
8
9
10
11
12 13 14 15 16 17 18
AMP01
NC
VOOS NULL
NC
VOOS NULL
NC
TEST PIN*
NC
VIOS NULL
NC
RS
RS
+VOP
NC
V+
–IN
RG
RG
NC
+IN
NC
VIOS NULL
SENSE
REF
OUT
NC
–VOP
NC
V–
*MAKE NO ELECTRICAL CONNECTION
20-Lead SOIC
TOP VIEW
(Not to Scale)
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
AMP01
–VOP
OUTPUT
REFERENCE
TEST PIN*
–IN
VOOS NULL
SENSE
TEST PIN*
VOOS NULL
V–
V+
+VOP
TEST PIN*
+IN
VIOS NULL
RS
RS
RGRG
VIOS NULL
*MAKE NO ELECTRICAL CONNECTION
FEATURES
Low Offset Voltage: 50 ␮V Max
Very Low Offset Voltage Drift: 0.3 ␮V/ⴗC Max
Low Noise: 0.12 ␮V p-p (0.1 Hz to 10 Hz)
Excellent Output Drive: ⴞ10 V at ⴞ50 mA
Capacitive Load Stability: to 1 ␮F
Gain Range: 0.1 to 10,000
Excellent Linearity: 16-Bit at G = 1000
High CMR: 125 dB min (G = 1000)
Low Bias Current: 4 nA Max
May Be Configured as a Precision Op Amp
Output-Stage Thermal Shutdown
Available in Die Form
*Protected under U.S. Patent Numbers 4,471,321 and 4,503,381.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 World Wide Web Site: http://www.analog.com
Fax: 781/326-8703 © Analog Devices, Inc., 1999
REV. D–2–
AMP01–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, unless otherwise noted)
AMP01A AMP01B
Parameter Symbol Conditions Min Typ Max Min Typ Max Units
OFFSET VOLTAGE
Input Offset Voltage V
IOS
T
A
= +25°C 2050 40100 µV
–55°C ≤ T
A
≤ +125°C 4080 60150 µV
Input Offset Voltage Drift TCV
IOS
–55°C ≤ T
A
≤ +125°C 0.15 0.3 0.3 1.0 µV/°C
Output Offset Voltage V
OOS
T
A
= +25°C1326mV
–55°C ≤ T
A
≤ +125°C36610mV
Output Offset Voltage Drift TCV
OOS
R
G
=
∞
–55°C ≤ T
A
≤ +125°C 2050 50120 µV/°C
Offset Referred to Input PSR G = 1000 120 130 110 120 dB
vs. Positive Supply G = 100 110 130 100 120 dB
V+ = +5 V to +15 V G = 10 95 110 90 100 dB
G = 1 75 90 70 80 dB
–55°C ≤ T
A
≤ +125°C
G = 1000 120 130 110 120 dB
G = 100 110 130 100 120 dB
G = 10 95 110 90 100 dB
G = 1 75 90 70 80 dB
Offset Referred to Input PSR G = 1000 105 125 105 115 dB
vs. Negative Supply G = 100 90 105 90 95 dB
V– = –5 V to –15 V G = 10 70 85 70 75 dB
G = 1 50 65 50 60 dB
–55°C ≤ T
A
≤ +125°C
G = 1000 105 125 105 115 dB
G = 100 90 105 90 95 dB
G = 10 70 85 70 75 dB
G = 1 50 85 50 60 dB
Input Offset Voltage Trim
Range V
S
= ±4.5 V to ±18 V
1
±6±6mV
Output Offset Voltage Trim
Range V
S
= ±4.5 V to ±18 V
1
±100 ±100 mV
INPUT CURRENT
Input Bias Current I
B
T
A
= +25°C1426nA
–55°C ≤ T
A
≤ +125°C410615nA
Input Bias Current Drift TCI
B
–55°C ≤ T
A
≤ +125°C 40 50 pA/°C
Input Offset Current I
OS
T
A
= +25°C 0.2 1.0 0.5 2.0 nA
–55°C ≤ T
A
≤ +125°C 0.5 3.0 1.0 6.0 nA
Input Offset Current Drift TCI
OS
–55°C ≤ T
A
≤ +125°C 3 5 pA/°C
INPUT
Input Resistance R
IN
Differential, G = 1000 1 1 GΩ
Differential, G ≤ 100 10 10 GΩ
Common Mode, G = 1000 20 20 GΩ
Input Voltage Range IVR T
A
= +25°C
2
±10.5 ±10.5 V
–55°C ≤ T
A
≤ +125°C±10.0 ±10.0 V
Common-Mode Rejection CMR V
CM
= ±10 V, 1 kΩ
Source Imbalance
G = 1000 125 130 115 125 dB
G = 100 120 130 110 125 dB
G = 10 100 120 95 110 dB
G = 1 85 100 75 90 dB
–55°C ≤ T
A
≤ +125°C
G = 1000 120 125 110 120 dB
G = 100 115 125 105 120 dB
G = 10 95 115 90 105 dB
G = 1 80 95 75 90 dB
NOTES
1
V
IOS
and V
OOS
nulling has minimal affect on TCV
IOS
and TCV
OOS
respectively.
2
Refer to section on common-mode rejection.
Specifications subject to change without notice.
ELECTRICAL CHARACTERISTICS
AMP01E AMP01F/G
Parameter Symbol Conditions Min Typ Max Min Typ Max Units
OFFSET VOLTAGE
Input Offset Voltage V
IOS
T
A
= +25°C 2050 40100 µV
T
MIN
≤ T
A
≤ T
MAX
40 80 60 150 µV
Input Offset Voltage Drift TCV
IOS
T
MIN
≤ T
A
≤ T
MAX1
0.15 0.3 0.3 1.0 µV/°C
Output Offset Voltage V
OOS
T
A
= +25°C1326mV
T
MIN
≤ T
A
≤ T
MAX
36 610 mV
Output Offset Voltage Drift TCV
OOS
R
G
=
∞
1
T
MIN
≤ T
A
≤ T
MAX
20 100 50 120 µV/°C
Offset Referred to Input PSR G = 1000 120 130 110 120 dB
vs. Positive Supply G = 100 110 130 100 120 dB
V+ = +5 V to +15 V G = 10 95 110 90 100 dB
G = 1 75 90 70 80 dB
T
MIN
≤ T
A
≤ T
MAX
G = 1000 120 130 110 120 dB
G = 100 110 130 100 120 dB
G = 10 95 110 90 100 dB
G = 1 75 90 70 80 dB
Offset Referred to Input PSR G = 1000 110 125 105 115 dB
vs. Negative Supply G = 100 95 105 90 95 dB
V– = –5 V to –15 V G = 10 75 85 70 75 dB
G = 1 55 65 50 60 dB
T
MIN
≤ T
A
≤ T
MAX
G = 1000 110 125 105 115 dB
G = 100 95 105 90 95 dB
G = 10 75 85 70 75 dB
G = 1 55 85 50 60 dB
Input Offset Voltage Trim
Range V
S
= ±4.5 V to ±18 V
2
±6±6mV
Output Offset Voltage Trim
Range V
S
= ±4.5 V to ±18 V
2
±100 ±100 mV
INPUT CURRENT
Input Bias Current I
B
T
A
= +25°C1426mV
T
MIN
≤ T
A
≤ T
MAX
410 615 mV
Input Bias Current Drift TCI
B
T
MIN
≤ T
A
≤ T
MAX
40 50 pA/°C
Input Offset Current I
OS
T
A
= +25°C 0.2 1.0 0.5 2.0 mV
T
MIN
≤ T
A
≤ T
MAX
0.5 3.0 1.0 6.0 mV
Input Offset Current Drift TCI
OS
T
MIN
≤ T
A
≤ T
MAX
3 5 pA/°C
INPUT
Input Resistance R
IN
Differential, G = 1000 1 1 GΩ
Differential, G ≤ 100 10 10 GΩ
Common Mode, G = 1000 20 20 GΩ
Input Voltage Range IVR T
A
= +25°C
3
±10.5 ±10.5 V
T
MIN
≤ T
A
≤ T
MAX
±10.0 ±10.0 V
Common-Mode Rejection CMR V
CM
= ±10 V, 1 kΩ
Source Imbalance
G = 1000 125 130 115 125 dB
G = 100 120 130 110 125 dB
G = 10 100 120 95 110 dB
G = 1 85 100 75 90 dB
T
MIN
≤ T
A
≤ T
MAX
G = 1000 120 125 110 120 dB
G = 100 115 125 105 120 dB
G = 10 95 115 90 105 dB
G = 1 80 95 75 90 dB
NOTES
1
Sample tested.
2
V
IOS
and V
OOS
nulling has minimal affect on TCV
IOS
and TCV
OOS
, respectively.
3
Refer to section on common-mode rejection.
Specifications subject to change without notice.
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, –25ⴗC ≤
T
A
≤ +85ⴗC for E, F
grades, 0ⴗC ≤ T
A
≤ +70ⴗC for G grade, unless otherwise noted)
AMP01
–3–
REV. D
AMP01
–4– REV. D
ELECTRICAL CHARACTERISTICS
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, unless otherwise noted)
AMP01A/E AMP01B/F/G
Parameter Symbol Conditions Min Typ Max Min Typ Max Units
GAIN
Gain Equation Accuracy G =
20 ×R
S
R
G
0.3 0.6 0.5 0.8 %
Accuracy Measured
from G = 1 to 1000
Gain Range G 0.1 10k 0.1 10k V/V
Nonlinearity G = 1000
1
0.0007 0.005 0.0007 0.005 %
G = 100
1
0.005 0.005 %
G = 10
1
0.005 0.007 %
G = 1
1
0.010 0.015 %
Temperature Coefficient G
TC
1 ≤ G ≤ 1000
1, 2
5 10 5 15 ppm°C
OUTPUT RATING
Output Voltage Swing V
OUT
R
L
= 2 kΩ±13.0 ±13.8 ±13.0 ±13.8 V
R
L
= 500 Ω±13.0 ±13.5 ±13.0 ±13.5 V
R
L
= 50 Ω±2.5 ±4.0 ±2.5 ±4.0 V
R
L
= 2 kΩ Over Temp. ±12.0 ±13.8 ±12.0 ±13.8 V
R
L
= 500 Ω
3
±12.0 ±13.5 ±12.0 ±13.5 V
Positive Current Limit Output-to-Ground Short 60 100 120 60 100 120 mA
Negative Current Limit Output-to-Ground Short 60 90 120 60 90 120 mA
Capacitive Load Stability 1 ≤ G ≤ 1000
No Oscillations
1
0.1 1 0.1 1 µF
Thermal Shutdown
Temperature Junction Temperature 165 165 °C
NOISE
Voltage Density, RTI e
n
f
O
= 1 kHz
e
n
G = 1000 5 5 nV/√Hz
e
n
G = 100 10 10 nV/√Hz
e
n
G = 10 59 59 nV/√Hz
e
n
G = 1 540 540 nV/√Hz
Noise Current Density, RTI i
n
f
O
= 1 kHz, G = 1000 0.15 0.15 pA/√Hz
Input Noise Voltage e
n
p-p 0.1 Hz to 10 Hz
e
n
p-p G = 1000 0.12 0.12 µV p-p
e
n
p-p G = 100 0.16 0.16 µV
p-p
e
n
p-p G = 10 1.4 1.4 µV p-p
e
n
p-p G = 1 13 13 µV p-p
Input Noise Current i
n
p-p 0.1 Hz to 10 Hz, G = 1000 2 2 pA p-p
DYNAMIC RESPONSE
Small-Signal G = 1 570 570 kHz
Bandwidth (–3 dB) BW G = 10 100 100 kHz
G = 100 82 82 kHz
G = 1000 26 26 kHz
Slew Rate SR G = 10 3.5 4.5 3.0 4.5 V/µs
Settling Time t
S
To 0.01%, 20 V step
G = 1 12 12 µs
G = 10 13 13 µs
G = 100 15 15 µs
G = 1000 50 50 µs
NOTES
1
Guaranteed by design.
2
Gain tempco does not include the effects of gain and scale resistor tempco match.
3
–55°C ≤ T
A
≤ +125°C for A/B grades, –25°C ≤ T
A
≤ +85°C for E/F grades, 0°C ≤ T
A
≤ 70°C for G grades.
Specifications subject to change without notice.
ORDERING GUIDE
Model Temperature Range Package Description Package Option
AMP01AX –55°C to +125°C 18-Lead Cerdip Q-18
AMP01AX/883C –55°C to +125°C 18-Lead Cerdip Q-18
AMP01BTC/883C –55°C to +125°C 28-Terminal LCC E-28A
AMP01BX –55°C to +125°C 18-Lead Cerdip Q-18
AMP01BX/883C –55°C to +125°C 18-Lead Cerdip Q-18
AMP01EX –25°C to +85°C 18-Lead Cerdip Q-18
AMP01FX –25°C to +85°C 18-Lead Cerdip Q-18
AMP01GBC Die
AMP01GS 0°C to +70°C 20-Lead SOIC R-20
AMP01GS-REEL 0°C to +70°C13" Tape and Reel R-20
AMP01NBC Die
5962-8863001VA* –55°C to +125°C 18-Lead Cerdip Q-18
5962-88630023A* –55°C to +125°C 28-Terminal LCC E-28A
5962-8863002VA* –55°C to +125°C 18-Lead Cerdip Q-18
*Standard military drawing available.
ELECTRICAL CHARACTERISTICS
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, unless otherwise noted)
AMP01A/E AMP01B/F/G
Parameter Symbol Conditions Min Typ Max Min Typ Max Units
SENSE INPUT
Input Resistance R
IN
35 50 65 35 50 65 kΩ
Input Current I
IN
Referenced to V– 280 280 µA
Voltage Range (Note 1) –10.5 +15 –10.5 +15 V
REFERENCE INPUT
Input Resistance R
IN
35 50 65 35 50 65 kΩ
Input Current I
IN
Referenced to V– 280 280 µA
Voltage Range (Note 1) –10.5 +15 –10.5 +15 V
Gain to Output 1 1 V/V
POWER SUPPLY –25°C ≤ T
A
≤ +85°C for E/F Grades, –55°C ≤ T
A
≤ +125°C for A/B Grades
Supply Voltage Range V
S
+V linked to +V
OP
±4.5 ±18 ±4.5 ±18 V
V
S
–V linked to –V
OP
±4.5 ±18 ±4.5 ±18 V
Quiescent Current I
Q
+V linked to +V
OP
3.0 4.8 3.0 4.8 mA
I
Q
–V linked to –V
OP
3.4 4.8 3.4 4.8 mA
NOTE
1
Guaranteed by design.
Specifications subject to change without notice.
AMP01
–5–
REV. D
1. RG
2. RG
3. –INPUT
4. VOOS NULL
5. VOOS NULL
6. TEST PIN*
7. SENSE
8. REFERENCE
9. OUTPUT
10. V– (OUTPUT)
11. V–
12. V+
13. V+ (OUTPUT)
14. RS
15. RS
16. VIOS NULL
17. VIOS NULL
18. +INPUT
*MAKE NO ELECTRICAL CONNECTION
DICE CHARACTERISTICS
Die Size 0.111 × 0.149 inch, 16,539 sq. mils
(2.82 × 3.78 mm, 10.67 sq. mm)
AMP01
–6– REV. D
WAFER TEST LIMITS
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, unless otherwise noted)
AMP01NBC AMP01GBC
Parameter Symbol Conditions Limit Limit Units
Input Offset Voltage V
IOS
60 120 µV max
Output Offset Voltage V
OOS
4 8 mV max
Offset Referred to Input PSR V+ = +5 V to +15 V dB min
vs. Positive Supply G = 1000 120 110 dB min
G = 100 110 100 dB min
G = 10 95 90 dB min
G = 1 75 70 dB min
Offset Referred to Input PSR V– = –5 V to –15 V dB min
vs. Negative Supply G = 1000 105 105 dB min
G = 100 90 90 dB min
G = 10 70 70 dB min
G = 1 50 50 dB min
Input Bias Current I
B
48nA max
Input Offset Current I
OS
13nA max
Input Voltage Range IVR Guaranteed by CMR Tests ±10 ±10 V min
Common Mode Rejection CMR V
CM
= ±10 V dB min
G = 1000 125 115 dB min
G = 100 120 110 dB min
G = 10 100 95 dB min
G = 1 85 75 dB min
Gain Equation Accuracy G =
20 ×RS
RG
0.6 0.8 % max
Output Voltage Swing V
OUT
R
L
= 2 kΩ±13 ±13 V min
V
OUT
R
L
= 500 Ω±13 ±13 V min
V
OUT
R
L
= 50 Ω±2.5 ±2.5 V min
Output Current Limit Output to Ground Short ±60 ±60 mA min
Output Current Limit Output to Ground Short ±120 ±120 mA max
Quiescent Current I
Q
+V Linked to +V
OP
4.8 4.8 mA max
–V Linked to –V
OP
4.8 4.8 mA max
NOTE
Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed
for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.
VIOS
NULL
RGAIN
RSCALE
VOOS
NULL
R1
47.5kV
R2
2.5kVR4
2.5kV
R3
47.5kV
A1
A3A2
Q1 Q2
250V
250V
–IN
+IN
REFERENCE
V+
+VOP
OUTPUT
–VOP
SENSE
V–
Figure 1. Simplified Schematic
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AMP01 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
AMP01
–7–
REV. D
ELECTRICAL CHARACTERISTICS
(@ V
S
= ⴞ15 V, R
S
= 10 k⍀, R
L
= 2 k⍀, T
A
= +25ⴗC, unless otherwise noted)
AMP01NBC AMP01GBC
Parameter Symbol Conditions Typical Typical Units
Input Offset Voltage Drift TCV
IOS
0.15 0.30 µV/°C
Output Offset Voltage Drift TCV
OOS
R
G
=
∞
20 50 µV/°C
Input Bias Current Drift TCI
B
40 50 pA/°C
Input Offset Current Drift TCI
OS
3 5 pA/°C
Nonlinearity G = 1000 0.0007 0.0007 %
Voltage Noise Density e
n
G = 1000
f
O
= 1 kHz 5 5 nV/√Hz
Current Noise Density i
n
G = 1000
f
O
= 1 kHz 0.15 0.15 pA/√Hz
Voltage Noise e
n
p-p G = 1000
0.1 Hz to 10 Hz 0.12 0.12 µV p-p
Current Noise i
n
p-p G = 1000 2 2 pA p-p
0.1 Hz to 10 Hz
Small-Signal Bandwidth (–3 dB) BW G = 1000 26 26 kHz
Slew Rate SR G = 10 4.5 4.5 V/µs
Settling Time t
S
To 0.01%, 20 V Step
G = 1000 50 50 µs
NOTE
Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed
for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.
AMP01
–8– REV. D
TEMPERATURE – 8C
–75 –50 150
–25 0 25 75 100 12550
INPUT OFFSET VOLTAGE – mV
50
40
–40
0
–10
–20
–30
30
10
20
VS = 615V
Figure 2. Input Offset Voltage
vs. Temperature
POWER SUPPLY VOLTAGE – Volts
OUTPUT OFFSET VOLTAGE CHANGE – mV
2.5
2.0
–1.0 065625610 615 620
1.0
0.5
0
–0.5
1.5
TA = +258C
Figure 5. Output Offset Voltage
Change vs. Supply Voltage
TEMPERATURE – 8C
INPUT OFFSET CURRENT – nA
0.8
–0.6
–75 –50 150
–25 025 75 100 12550
0.6
0.2
0.0
–0.2
–0.4
0.4
VS = 615V
Figure 8. Input Offset Current
vs. Temperature
POWER SUPPLY VOLTAGE – Volts
INPUT OFFSET VOLTAGE – mV
8
6
–6 065620610 615
2
0
–2
–4
4
TA = +258C
UNIT NO.
1
2
3
4
Figure 3. Input Offset Voltage
vs. Supply Voltage
TEMPERATURE – 8C
INPUT BIAS CURRENT – nA
5
–2
–75 –50 150
–25 025 75 100 12550
4
2
1
0
–1
3
VS = 615V
Figure 6. Input Bias Current
vs. Temperature
VOLTAGE GAIN – G
COMMON-MODE REJECTION – dB
140
1001 10k
120
10 100 1k
130
110
VS = 615V
TA = +258C
Figure 9. Common-Mode Rejection
vs. Voltage Gain
TEMPERATURE – 8C
–75 –50 150
–25 0 25 75 100 12550
OUTPUT OFFSET VOLTAGE – mV
5
4
–4
0
–1
–2
–3
3
1
2
–5
VS = 615V
Figure 4. Output Offset Voltage
vs. Temperature
POWER SUPPLY VOLTAGE – Volts
INPUT BIAS CURRENT – nA
2.0
1.5
–1.5 065620610 615
0.5
0
–0.5
–1.0
1.0
TA = +258C
Figure 7. Input Bias Current
vs. Supply Voltage
FREQUENCY – Hz
COMMON-MODE REJECTION – dB
140
01 10 100k100 1k 10k
120
100
80
60
40
20
VCM = 2V p-p
VS = 615V
TA = +258C
G = 1000
G = 100
G = 1
G = 10
Figure 10. Common-Mode Rejection
vs. Frequency
–Typical Performance Characteristics
AMP01
–9–
REV. D
TEMPERATURE – 8C
COMMON-MODE INPUT VOLTAGE – Volts
16
0
–75 –50 150
–25 025 75 100 12550
14
8
6
4
2
12
10
VDM = 0
VS = 615V
VS = 610V
VS = 65V
Figure 11. Common-Mode Voltage
Range vs. Temperature
LOAD RESISTANCE – V
OUITPUT VOLTAGE – Volts
18
10
10k
14
10 100 1k
16
12
8
6
4
2
0
VS = 615V
Figure 14. Maximum Output Voltage
vs. Load Resistance
FREQUENCY – Hz
VOLTAGE GAIN – dB
80
0
100 1M
40
1k 10k 100k
60
20
–20
–40 101
VS = 615V
TA = +258C
G = 1000
G = 100
G = 10
G = 1
Figure 17. Closed-Loop Voltage
Gain vs. Frequency
FREQUENCY – Hz
POWER SUPPLY REJECTION – dB
140
01 10 100k100 1k 10k
120
100
80
60
40
20
VS = 615V
TA = +258C
DVS = 61V
G = 1000
G = 100
G = 10
G = 1
Figure 12. Positive PSR
vs. Frequency
FREQUENCY – Hz
PEAK-TO-PEAK AMPLITUDE – Volts
30
10
100 1M
20
1k 10k 100k
25
15
VS = 615V
RL = 2kV
5
0
Figure 15. Maximum Output Swing
vs. Frequency
FREQUENCY – Hz
TOTAL HARMONIC DISTORTION – %
0.08
0.04
10k
0.06
10 100 1k
0.07
0.05
0.03
0.02
0.01
0
G = 1000
G = 100
G = 10
G = 1
VS = 615V
RL = 600V
VOUT = 20V p-p
Figure 18. Total Harmonic Distortion
vs. Frequency
FREQUENCY – Hz
POWER SUPPLY REJECTION – dB
140
01 10 100k100 1k 10k
120
100
80
60
40
20
VS = 615V
TA = +258C
DVS = 61V
G = 1000
G = 100
G = 10
G = 1
Figure 13. Negative PSR
vs. Frequency
FREQUENCY – Hz
OUTPUT IMPEDANCE – V
100
10 100 1M1k 10k 100k
10
1.0
0.1
0.01
0.001
VS = 615V
IOUT = 20mA p-p
G = 1000
G = 1
Figure 16. Closed-Loop Output
Impedance vs. Frequency
LOAD RESISTANCE – V
TOTAL HARMONIC DISTORTION – %
0.02
0.01
10k100 1k
0
VS = 615V
G = 100
f = 1kHz
VOUT = 20V p-p
Figure 19. Total Harmonic Distortion
vs. Load Resistance
AMP01
–10– REV. D
LOAD CAPACITANCE – F
SLEW RATE – V/ms
6
2
100p 1m
4
1n 10n 100n
5
3
VS = 615V
1
0
Figure 21. Slew Rate vs.
Load Capacitance
VOLTAGE GAIN – G
1k
11k
10
1 10 100
100
VS = 615V
f = 1kHz
VOLTAGE NOISE – nV/ Hz
Figure 24. RTI Voltage Noise
Density vs. Gain
TEMPERATURE – 8C
POSITIVE SUPPLY CURRENT – mA
6
0–75 –50 150–25 0 25 75 100 12550
5
4
3
2
1
VS = 615V
Figure 27. Positive Supply Current
vs. Temperature
VOLTAGE GAIN – G
SLEW RATE – V/ms
6
3
1k
4
1 10 100
5
2
1
0
VS = 615V
Figure 20. Slew Rate vs.
Voltage Gain
FREQUENCY – Hz
VOLTAGE NOISE – nV/ Hz
15
5
1 10k
10
10 100 1k
0
G = 1000
Figure 23. Voltage Noise Density
vs. Frequency
POWER SUPPLY VOLTAGE – Volts
NEGATIVE SUPPLY CURRENT – mA
–8
–7
0065620
610 615
–4
–3
–2
–1
–6
–5
TA = +258C
Figure 26. Negative Supply Current
vs. Supply Voltage
VOLTAGE GAIN – G
SETTLING TIME – ms
70
40
1k
50
1 10 100
60
30
20
10
VS = 615V
20V STEP
Figure 22. Settling Time to 0.01%
vs. Voltage Gain
POWER SUPPLY VOLTAGE – Volts
POSITIVE SUPPLY CURRENT – mA
8
7
0065620
610 615
4
3
2
1
6
5
TA = +258C
Figure 25. Positive Supply Current
vs. Supply Voltage
TEMPERATURE – 8C
NEGATIVE SUPPLY CURRENT – mA
–6
0–75 –50 150–25 0 25 75 100 12550
–5
–4
–3
–2
–1
VS = 615V
VSENSE = VREF = 0V
Figure 28. Negative Supply Current
vs. Temperature
AMP01
–11–
REV. D
GAIN
The AMP01 uses two external resistors for setting voltage gain
over the range 0.1 to 10,000. The magnitudes of the scale resis-
tor, R
S
, and gain-set resistor, R
G
, are related by the formula:
G = 20 × R
S
/R
G
, where G is the selected voltage gain (refer to
Figure 29).
REFERENCE OUTPUT
V+
V–
RS
RG
+IN
–IN
VOLTAGE GAIN, G = 20 3 RS
RG
( )
SENSE
AMP01
14 15 13 12 79
8
11
10
3
2
1
18
Figure 29. Basic AMP01 Connections for Gains
0.1 to 10,000
The magnitude of R
S
affects linearity and output referred errors.
Circuit performance is characterized using R
S
= 10 kΩ when
operating on ±15 volt supplies and driving a ±10 volt output. R
S
may be reduced to 5 kΩ in many applications particularly when
operating on ±5 volt supplies or if the output voltage swing is
limited to ±5 volts. Bandwidth is improved with R
S
= 5 kΩ and
this also increases common-mode rejection by approximately
6 dB at low gain. Lowering the value below 5 kΩ can cause
instability in some circuit configurations and usually has no
advantage. High voltage gains between two and ten thousand
would require very low values of R
G
. For R
S
= 10 kΩ and
A
V
= 2000 we get R
G
= 100 Ω; this value is the practical lower
limit for R
G
. Below 100 Ω, mismatch of wirebond and resistor
temperature coefficients will introduce significant gain tempco
errors. Therefore, for gains above 2,000, R
G
should be kept
constant at 100 Ω and R
S
increased. The maximum gain of
10,000 is obtained with R
S
set to 50 kΩ.
Metal-film or wirewound resistors are recommended for best
results. The absolute values and TCs are not too important,
only the ratiometric parameters.
AC amplifiers require good gain stability with temperature and
time, but dc performance is unimportant. Therefore, low cost
metal-film types with TCs of 50 ppm/°C are usually adequate
for R
S
and R
G
. Realizing the full potential of the AMP01’s offset
voltage and gain stability requires precision metal-film or wire-
wound resistors. Achieving a 15 ppm/°C gain tempco at all gains
requires R
S
and R
G
temperature coefficient matching to
5 ppm/°C or better.
INPUT AND OUTPUT OFFSET VOLTAGES
Instrumentation amplifiers have independent offset voltages
associated with the input and output stages. While the initial
offsets may be adjusted to zero, temperature variations will
cause shifts in offsets. Systems with auto-zero can correct for
offset errors, so initial adjustment would be unnecessary. How-
ever, many high-gain applications don’t have auto zero. For
these applications, both offsets can be nulled, which has mini-
mal effect on TCV
IOS
and TCV
OOS
The input offset component is directly multiplied by the ampli-
fier gain, whereas output offset is independent of gain. There-
fore, at low gain, output-offset errors dominate, while at high
gain, input-offset errors dominate. Overall offset voltage, V
OS
,
referred to the output (RTO) is calculated as follows;
V
OS
(RTO) = (V
IOS
×
G) + V
OOS
(1)
where V
IOS
and V
OOS
are the input and output offset voltage
specifications and G is the amplifier gain. Input offset nulling
alone is recommended with amplifiers having fixed gain above
50. Output offset nulling alone is recommended when gain is
fixed at 50 or below.
In applications requiring both initial offsets to be nulled, the
input offset is nulled first by short-circuiting R
G
, then the output
offset is nulled with the short removed.
The overall offset voltage drift TCV
OS
, referred to the output, is
a combination of input and output drift specifications. Input
offset voltage drift is multiplied by the amplifier gain, G, and
summed with the output offset drift;
TCV
OS
(RTO) = (TCV
IOS
× G) + TCV
OOS
(2)
where TCV
IOS
is the input offset voltage drift, and TCV
OOS
is
the output offset voltage specification. Frequently, the amplifier
drift is referred back to the input (RTI), which is then equiva-
lent to an input signal change;
TCV
OS
(RTI) = TCV
IOS
TCV
OOS
G
(3)
For example, the maximum input-referred drift of an AMP01 EX
set to G = 1000 becomes;
TCV
OS
(RTI ) = 0.3
µ
V/°C +
100
1000
µVC/°
= 0.4
µ
V/°C max
INPUT BIAS AND OFFSET CURRENTS
Input transistor bias currents are additional error sources that
can degrade the input signal. Bias currents flowing through the
signal source resistance appear as an additional offset voltage.
Equal source resistance on both inputs of an IA will minimize
offset changes due to bias current variations with signal voltage
and temperature. However, the difference between the two bias
currents, the input offset current, produces a nontrimmable
error. The magnitude of the error is the offset current times the
source resistance.
A current path must always be provided between the differential
inputs and analog ground to ensure correct amplifier operation.
Floating inputs, such as thermocouples, should be grounded
close to the signal source for best common-mode rejection.
AMP01
–12– REV. D
VOLTAGE GAIN
1M
1 10k
RESISTANCE – V
10k
10 100 1k
VS = 615V
100k
1k
100
RS
RG
Figure 30. R
G
and R
S
Selection
Gain accuracy is determined by the ratio accuracy of R
S
and R
G
combined with the gain equation error of the AMP01 (0.6%
max for A/E grades).
All instrumentation amplifiers require attention to layout so
thermocouple effects are minimized. Thermocouples formed
between copper and dissimilar metals can easily destroy the
TCV
OS
performance of the AMP01 which is typically
0.15 µV/°C. Resistors themselves can generate thermoelectric
EMF’s when mounted parallel to a thermal gradient. “Vishay”
resistors are recommended because a maximum value for ther-
moelectric generation is specified. However, where thermal
gradients are low and gain TCs of 20 ppm–50 ppm are suffi-
cient, general-purpose metal-film resistors can be used for R
G
and R
S
.
COMMON-MODE REJECTION
Ideally, an instrumentation amplifier responds only to the dif-
ference between the two input signals and rejects common-
mode voltages and noise. In practice, there is a small change in
output voltage when both inputs experience the same common-
mode voltage change; the ratio of these voltages is called the
common-mode gain. Common-mode rejection (CMR) is the
logarithm of the ratio of differential-mode gain to common-
mode gain, expressed in dB. CMR specifications are normally
measured with a full-range input voltage change and a specified
source resistance unbalance.
The current-feedback design used in the AMP01 inherently
yields high common-mode rejection. Unlike resistive feedback
designs, typified by the three-op-amp IA, the CMR is not de-
graded by small resistances in series with the reference input. A
slight, but trimmable, output offset voltage change results from
resistance in series with the reference input.
The common-mode input voltage range, CMVR, for linear
operation may be calculated from the formula:
CMVR = ±
IVR –|VOUT
|
2G
(4)
IVR is the data sheet specification for input voltage range; V
OUT
is the maximum output signal; G is the chosen voltage gain. For
example, at +25°C, IVR is specified as ±10.5 volt minimum
with ±15 volt supplies. Using a ±10 volt maximum swing out-
put and substituting the figures in (4) simplifies the formula to:
CMVR = ±
10.5 – 5
G
(5)
For all gains greater than or equal to 10, CMVR is ±10 volt
minimum; at gains below 10, CMVR is reduced.
ACTIVE GUARD DRIVE
Rejection of common-mode noise and line pick-up can be im-
proved by using shielded cable between the signal source and
the IA. Shielding reduces pick-up, but increases input capaci-
tance, which in turn degrades the settling-time for signal
changes. Further, any imbalance in the source resistance be-
tween the inverting and noninverting inputs, when capacitively
loaded, converts the common-mode voltage into a differential
voltage. This effect reduces the benefits of shielding. AC
common-mode rejection is improved by “bootstrapping” the
input cable capacitance to the input signal, a technique called
“guard driving.” This technique effectively reduces the input
capacitance. A single guard-driving signal is adequate at gains
above 100 and should be the average value of the two inputs.
The value of external gain resistor R
G
is split between two resis-
tors R
G1
and R
G2
; the center tap provides the required signal to
drive the buffer amplifier (Figure 31).
GROUNDING
The majority of instruments and data acquisition systems have
separate grounds for analog and digital signals. Analog ground
may also be divided into two or more grounds which will be tied
together at one point, usually the analog power-supply ground.
In addition, the digital and analog grounds may be joined, nor-
mally at the analog ground pin on the A-to-D converter. Fol-
lowing this basic grounding practice is essential for good circuit
performance (Figure 32).
Mixing grounds causes interactions between digital circuits and
the analog signals. Since the ground returns have finite resis-
tance and inductance, hundreds of millivolts can be developed
between the system ground and the data acquisition compo-
nents. Using separate ground returns minimizes the current flow
in the sensitive analog return path to the system ground point.
Consequently, noisy ground currents from logic gates do not
interact with the analog signals.
Inevitably, two or more circuits will be joined together with their
grounds at differential potentials. In these situations, the differ-
ential input of an instrumentation amplifier, with its high CMR,
can accurately transfer analog information from one circuit to
another.
SENSE AND REFERENCE TERMINALS
The sense terminal completes the feedback path for the instru-
mentation amplifier output stage and is normally connected
directly to the output. The output signal is specified with re-
spect to the reference terminal, which is normally connected to
analog ground.
AMP01
–13–
REV. D
VOLTAGE GAIN, G = 20 3 RS
RG1
( )
AV = 500 WITH COMPONENTS SHOWN
*
+15V
C1
0.047mF
+C5
10mF
R5
*
*
SENSE
OUTPUT
*SOLDER LINK
REFERENCE
*
GROUND
R3
C4
0.047mF
+C6
10mFC2
0.047mF
VR1
100kV
VR2
100kV
RG1
400V
+IN
–IN
RG3
200V
RG2
200V
741
+15V
–15V
GUARD
DRIVE
R2
1MVR1
1MV
SIGNAL
GROUND
RS
10kV
C3
0.047mF
–15V
R4
RS
NC
*
AMP01
RS
VIOS
NULL
VOOS
NULL
V+
V–
RG
RG
15
14 613
12 7
18
1
2
3
16
17 4510 11 8
9
7
6
43
2
Figure 31. AMP01 Evaluation Circuit Showing Guard-Drive Connection
HOLD
CAPACITOR
C = 0.047mF CERAMIC CAPACITORS
DIGITAL
DATA
OUTPUT
ANALOG
GROUND DIGITAL
GROUND
ADC
CC
7
8
9
OUTPUT
REFERENCE
SMP-11
SAMPLE AND HOLD
C
+
4.7mF
DIGITAL
GROUND
CC
0V +5V
DIGITAL
POWER SUPPLY
–15V+15V 0V
ANALOG
POWER SUPPLY
CC
AMP01
Figure 32. Basic Grounding Practice
AMP01
–14– REV. D
If heavy output currents are expected and the load is situated
some distance from the amplifier, voltage drops due to track or
wire resistance will cause errors. Voltage drops are particularly
troublesome when driving 50 Ω loads. Under these conditions,
the sense and reference terminals can be used to “remote sense”
the load as shown in Figure 33. This method of connection puts
the I×R drops inside the feedback loop and virtually eliminates
the error. An unbalance in the lead resistances from the sense
and reference pins does not degrade CMR, but will change the
output offset voltage. For example, a large unbalance of 3 Ω will
change the output offset by only 1 mV.
DRIVING 50 ⍀ LOADS
Output currents of 50 mA are guaranteed into loads of up to
50 Ω and 26 mA into 500 Ω. In addition, the output is stable
and free from oscillation even with a high load capacitance. The
combination of these unique features in an instrumentation
amplifier allows low-level transducer signals to be conditioned
and directly transmitted through long cables in voltage or cur-
rent form. Increased output current brings increased internal
dissipation, especially with 50 Ω loads. For this reason, the
power-supply connections are split into two pairs; pins 10 and
13 connect to the output stage only and pins 11 and 12 provide
power to the input and following stages. Dual supply pins allow
dropper resistors to be connected in series with the output stage
so excess power is dissipated outside the package. Additional
decoupling is necessary between pins 10 and 13 to ground to
maintain stability when dropper resistors are used. Figure 34
shows a complete circuit for driving 50 Ω loads.
AMP01
+IN
–IN
RG
18
1
2
3
14
15 12 13 79
8
10
11
V–
V+
SENSE
REFERENCE
*
*
OUTPUT
GROUND
TWISTED
PAIRS REMOTE
LOAD
IN4148 DIODES ARE OPTIONAL. DIODES LIMIT THE OUTPUT
VOLTAGE EXCURSION IF SENSE AND/OR REFERENCE LINES
BECOME DISCONNECTED FROM THE LOAD.
*
RS
Figure 33. Remote Load Sensing
+IN
–IN
RGAMP01
14
15
12
13 79
8
10
11
18
1
2
3
RS
5kV
R1
130V
1W
C1
0.047mF
0.047mF
+15V
SENSE
REFERENCE
VOUT
63V MAX
50V
LOAD
C2
0.047mF
R2
130V
1W 0.047mF
–15V
POWER BANDWIDTH, G = 100, 130kHz
POWER BANDWIDTH, G = 10, 200kHz
T.H.D.~0.04% @ 1kHz, 2Vrms
VOLTAGE GAIN, G =
( )
20 3 RS
RG
RESISTERS R1 AND R2 REDUCE IC DISSIPATION
Figure 34. Driving 50
Ω
Loads
AMP01
–15–
REV. D
HEATSINKING
To maintain high reliability, the die temperature of any IC
should be kept as low as practicable, preferably below 100°C.
Although most AMP01 application circuits will produce very
little internal heat — little more than the quiescent dissipation
of 90 mW—some circuits will raise that to several hundred
milliwatts (for example, the 4-20 mA current transmitter appli-
cation, Figure 37). Excessive dissipation will cause thermal
shutdown of the output stage thus protecting the device from
damage. A heatsink is recommended in power applications to
reduce the die temperature.
Several appropriate heatsinks are available; the Thermalloy
6010B is especially easy to use and is inexpensive. Intended for
dual-in-line packages, the heatsink may be attached with a
cyanoacrylate adhesive. This heatsink reduces the thermal resis-
tance between the junction and ambient environment to ap-
proximately 80°C/W. Junction (die) temperature can then be
calculated by using the relationship:
P
d
=
T
J
–T
A
θ
JA
where T
J
and T
A
are the junction and ambient temperatures
respectively, θ
JA
is the thermal resistance from junction to ambi-
ent, and P
d
is the device’s internal dissipation.
OVERVOLTAGE PROTECTION
Instrumentation amplifiers invariably sit at the front end of
instrumentation systems where there is a high probability of
exposure to overloads. Voltage transients, failure of a trans-
ducer, or removal of the amplifier power supply while the signal
source is connected may destroy or degrade the performance of
an unprotected amplifier. Although it is impractical to protect
an IC internally against connection to power lines, it is relatively
easy to provide protection against typical system overloads.
The AMP01 is internally protected against overloads for gains
of up to 100. At higher gains, the protection is reduced and
some external measures may be required. Limited internal over-
load protection is used so that noise performance would not be
significantly degraded.
AMP01 noise level approaches the theoretical noise floor of the
input stage which would be 4 nV/√Hz at 1 kHz when the gain is
set at 1000. Noise is the result of shot noise in the input devices
and Johnson noise in the resistors. Resistor noise is calculated
from the values of R
G
(200 Ω at a gain of 1000) and the input
protection resistors (250 Ω). Active loads for the input transis-
tors contribute less than 1 nV/√Hz of noise. The measured noise
level is typically 5 nV/√Hz.
Diodes across the input transistor’s base-emitter junctions,
combined with 250 Ω input resistors and R
G
, protect against
differential inputs of up to ±20 V for gains of up to 100. The
diodes also prevent avalanche breakdown that would degrade
the I
B
and I
OS
specifications. Decreasing the value of R
G
for
gains above 100 limits the maximum input overload protection
to ±10 V.
External series resistors could be added to guard against higher
voltage levels at the input, but resistors alone increase the input
noise and degrade the signal-to-noise ratio, especially at high
gains.
Protection can also be achieved by connecting back-to-back
9.1 V Zener diodes across the differential inputs. This technique
does not affect the input noise level and can be used down to a
gain of 2 with minimal increase in input current. Although
voltage-clamping elements look like short circuits at the limiting
voltage, the majority of signal sources provide less than 50 mA,
producing power levels that are easily handled by low-power
Zeners.
Simultaneous connection of the differential inputs to a low
impedance signal above 10 V during normal circuit operation is
unlikely. However, additional protection involves adding 100 Ω
current-limiting resistors in each signal path prior to the voltage
clamp, the resistors increase the input noise level to just
5.4 nV/√Hz (refer to Figure 35).
Input components, whether multiplexers or resistors, should be
carefully selected to prevent the formation of thermocouple
junctions that would degrade the input signal.
VOUT
+15V
+IN
–IN
AMP01
9.1V 1W
ZENERS
100V
1W*
100V
1W*
OPTIONAL PROTECTION
RESISTORS, SEE TEXT.
*LINEAR INPUT RANGE,
65V MAXIMUM
DIFFERENTIAL PROTECTION
TO 630V
–15V
Figure 35. Input Overvoltage Protection for Gains
2 to 10,000
POWER SUPPLY CONSIDERATIONS
Achieving the rated performance of precision amplifiers in a
practical circuit requires careful attention to external influences.
For example, supply noise and changes in the nominal voltage
directly affect the input offset voltage. A PSR of 80 dB means
that a change of 100 mV on the supply, not an uncommon
value, will produce a 10 µV input offset change. Consequently,
care should be taken in choosing a power unit that has a low
output noise level, good line and load regulation, and good
temperature stability.
AMP01
–16– REV. D
6IOUT
R1
100V
R2
200V
ROUT
TRIM
SENSE
79
8
REFERENCE
10
11
15
14
12
13
18
1
2
3
RG
2kV
+IN
–IN
VIN
RS
2kV
–15V
0.047mF
0.047mF
+15V COMPLIANCE, TYPICALLY 610V
LINEARITY ~0.01%
OUTPUT RESISTANCE AT 20mA ~5MV
POWER BANDWIDTH (–3dB) ~60kHz
INTO 500V LOAD
IOUT = VIN
( )
20 3 RS
RG 3 R1
AMP01
R1 = 100V FOR IOUT = 620mA
VIN = 6100mV FOR 620mA FULL SCALE
RG
RG
RS
RS
V+
V–
Figure 36. High Compliance Bipolar Current Source with 13-Bit Linearity
ROUT TRIM
79
8
10
11
15
14
12
13
18
1
2
3
RG
2.75kV
+IN
–IN
0V
RS
2kV
0.047mF
AMP01
COMPLIANCE OF IOUT, +20V WITH +30V SUPPLY (OUTPUT w.r.t. 0V)
DIFFERENTIAL INPUT OF 100mV FOR 16mA SPAN
OUTPUT RESISTANCE ~5MV AT IOUT = 20mA
LINEARITY 0.01% OF SPAN
0.047mF
R5
2.21kV
R6
500V
ZERO TRIM
R2
200V
REF-02
R3
100V
4
6
2
R1
100V
+15V
TO +30V
IOUT
4mA TO 20mA
–5V
ALL RESISTORS 1% METAL FILM
RS
RS
RG
RGV–
V+
R4
100V
Figure 37. 13-Bit Linear 4–20 mA Transmitter Constructed by Adding a Voltage Reference.
Thermocouple Signals Can Be Accepted Without Preamplification.
AMP01
–17–
REV. D
+
+IN
–IN
RG
14
15
12
13
79
8
10
11
18
1
2
3
10kV
SENSE
REFERENCE
0.047mF
VOLTAGE GAIN, G = 100
POWER BANDWIDTH (–3dB), 60kHz
QUIESCENT CURRENT, 4mA
LINEARITY~0.01% @ FULL OUTPUT INTO 10V
10mF
+
0.047mF
100V
0.047mF
2N4921
2N4918
+15V
VOUT
(610V INTO 10V)
–15V
AMP01
RS
RS
V+
V–
RG
RG
GND
Figure 38. Adding Two Transistors Increases Output Current to
±
1 A Without Affecting the Quiescent Current of 4 mA.
Power Bandwidth is 60 kHz.
0.047mF
OUT
100kV100kV
+IN
–IN
IC2
+15V
–15V
7
6
4
3
2
RS
10kV
RS
AMP01
RS
VIOS
NULL
VOOS
NULL
V+
V–
RG
RG
15
14
13
12
7
18
1
2
3
16
17 45
10
11
8
9
SENSE
REFERENCE
GND
0.047mF
–15V
++++
10
8
6
4
3
IC1
+15V
G1 G10 G100 G1000
57911
TTL COMPATIBLE INPUTS
27kV
2.7kV
12
131412
47kV
47kV
47kV
47kV
200kV20kV2kV196V
Q4
Q5
Q3
Q2
Q1
+15V
Q1, Q2...........J110
Q3, Q4, Q5....J107
IC1 ...............CMP-04
IC2 ...............OP15GZ
LINEARITY~0.005%, G = 10 AND 100
~0.02%, G = 1 AND 1000
GAIN ACCURACY, UNTRIMMED~0.5%
SETTLING TIME TO 0.01%, ALL GAINS,
LESS THAN 75ms
GAIN SWITCHING TIME, LESS THAN 100ms
Figure 39. The AMP01 Makes an Excellent Programmable-Gain Instrumentation Amplifier. Combined Gain-Switching
and Settling Time to 13 Bits Falls Below 100
µ
s. Linearity Is Better than 12 Bits over a Gain Range 1 to 1000.
AMP01
–18– REV. D
15
14
18
1
2
3
RG
–IN
RS
10kV
MAXIMUM OUTPUT, 20V p-p INTO 600V
T.H.D. 0.01% @ 1kHz, 20V p-p INTO 600V, G = 10
( )
20 3 RS
RG
VOLTAGE GAIN, G =
AMP01
RG
RG
RS
RS
+
RL
DIFFERENTIAL
OUTPUT
OUTPUT
COMMON-MODE
REFERENCE
(65V MAX)
OP37
2
3
7
6
4
470pF
1.5kV
*5kV
*5kV
*MATCHED TO 0.1%
7
8
9
13
10
12
11
0V
V+
V–
–15V
+15V
0V
0.047mF
0.047mF
0V
SENSE
+IN
REFERENCE
Figure 40. A Differential Input Instrumentation Amplifier with Differential Output Replaces a Transformer in Many
Applications. The Output will Drive a 600
Ω
Load at Low Distortion, (0.01%).
18
1
2
3
R1
390V
CLOSED-LOOP VOLTAGE GAIN MUST BE
GREATER THAN 50 FOR STABLE OPERATION
( )
R2
R3
1 +
VOLTAGE GAIN, G =
VIN
AMP01
RG
RG
RS
RS
7
8
9
13
10
12
11
V+
V–
VOUT
+15V
NC NC –15V
0.047mF+10mF
R2
4.95kV
R3
50V
RL
CL
15
14
POWER BANDWIDTH (–3dB)~150kHz
TOTAL HARMONIC DISTORTION~0.006%
@1kHz, 20V p-p INTO 500V // 1000pF
0.047mF+10mF
REF
SENSE
NC = NO CONNECT
Figure 41. Configuring the AMP01 as a Noninverting Operational Amplifier Provides Exceptional Performance. The
Output Handles Low Load Impedances at Very Low Distortion, 0.006%.
AMP01
–19–
REV. D
18
1
2
3
R4
VIN
7
8
9
10
11
VOUT
R1 = R2
GAIN (G)
R3 = R1 // R2
R4 = 1.5kV @ G = 1
1.2kV @ G = 10
120V @ G = 100 AND 1000 –15V
+10mF
15
14
20V p-p INTO 500V // 1000pF.
TOTAL HARMONIC DISTORTION:
<0.005% @ 1kHz, VOUT = 20V p-p
NC NC R2
220kV
0.01mF
4.7kV
R3
+15V
AMP01
RG
RG
RS
RS
SENSE
V–
REF
0.047mF0.047mF
10mF
+
G = 1 TO 1000
R1
12
13
V–
Figure 42. The Inverting Operational Amplifier Configuration has Excellent Linearity over the Gain Range 1 to 1000, Typically
0.005%. Offset Voltage Drift at Unity Gain Is Improved over the Drift in the Instrumentation Amplifier Configuration.
18
1
2
3
RG
3kV
VIN
AMP01
RG
RG
REF
SENSE
7
8
9
13
10
12
11
V+
V–
VOUT
+15V
NC
NC –15V
0.047mF+10mF
R2
4.7kV
RL
CL
15
14
POWER BANDWIDTH (–3dB)~60kHz
TOTAL HARMONIC DISTORTION~0.001%
@1kHz, 20V p-p INTO 500V // 1000pF
NC = NO CONNECT
0.047mF+10mF
680pF
0.01mF
R3
330V
R1
4.7kV
RS
RS
Figure 43. Stability with Large Capacitive Loads Combined with High Output Current Capability make the AMP01 Ideal
for Line Driving Applications. Offset Voltage Drift Approaches the TCV
IOS
Limit, (0.3
µ
V/
°
C).
AMP01
–20– REV. D
18
1
2
3
9
13
10
12 11
V–
200kV20kV2kV200V
G1
G10 G100
G1000
V+
RG
RG
7
8
1/2 OP215
+
–
84
V–
V+
1
2
3
1.82kV
1mF
16.2kV
OUTPUT
1/2 OP215
+
–
1.62MV
9.09kV
G1000
G1,10,100
100V1kV
8
AMP01
14
15
10kV
RS
RS
RG
RG16.2kV
1mF
en
(G = 1, 10, 100) = eOUT
1000 3 G
en
(G = 1000) = eOUT
100 3 G
1mF
5
6
7
Figure 44. Noise Test Circuit (0.1 Hz to 10 Hz)
18
1
2
3
9
13
10
12
11
200kV
0.1% 20kV
0.1% 2kV
0.1% 200V
0.1%
G1G10 G100
G1000
RG
RG
7
88
14
15
0.047mF 0.047mF
V+ V–
10kV
0.1%
10V
0.1%
102V
0.1%
1.1kV
0.1%
G1000
G100
G1
G10
10kV
0.1% 2kV
0.1%
RG
RG
AMP01
RS
RS
1.91kV
0.1%
200V
10T
2 3 HSCH-1001
VIN
20V p-p VOUT
Figure 45. Settling-Time Test Circuit
AMP01
–21–
REV. D
Figure 46. Instrumentation Amplifier with Autozero
+18V
–18V
10kV
SENSE
AMP01
14 15
13
12
79
8
11 10
3
2
1
18
0.047mF
VOUT
10kV
0.047mF
RSRS
RG
RG
Figure 47. Burn-In Circuit
AMP01
–22– REV. D
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
18-Lead Cerdip
(Q-18)
18
19
10 0.310 (7.87)
0.220 (5.59)
PIN 1
0.005 (0.13) MIN 0.098 (2.49) MAX
SEATING
PLANE
0.023 (0.58)
0.014 (0.36)
0.200 (5.08)
MAX
0.960 (24.38) MAX
0.150
(3.81)
MIN
0.070 (1.78)
0.030 (0.76)
0.200 (5.08)
0.125 (3.18) 0.100
(2.54)
BSC
0.060 (1.52)
0.015 (0.38)
158
08
0.320 (8.13)
0.290 (7.37)
0.015 (0.38)
0.008 (0.20)
20-Lead SOIC
(R-20)
SEATING
PLANE
0.0118 (0.30)
0.0040 (0.10) 0.0192 (0.49)
0.0138 (0.35)
0.1043 (2.65)
0.0926 (2.35)
0.0500
(1.27)
BSC 0.0125 (0.32)
0.0091 (0.23)
0.0500 (1.27)
0.0157 (0.40)
88
08
0.0291 (0.74)
0.0098 (0.25) 3 458
20 11
101
0.5118 (13.00)
0.4961 (12.60)
0.4193 (10.65)
0.3937 (10.00)
0.2992 (7.60)
0.2914 (7.40)
PIN 1
28-Terminal Ceramic Leadless Chip Carrier
(E-28A)
1
28 5
11
12
18
26
BOTTOM
VIEW
19
4
25 0.028 (0.71)
0.022 (0.56)
458 TYP
0.015 (0.38)
MIN
0.055 (1.40)
0.045 (1.14)
0.050
(1.27)
BSC
0.075
(1.91)
REF
0.011 (0.28)
0.007 (0.18)
R TYP
0.095 (2.41)
0.075 (1.90)
0.150
(3.51)
BSC
0.300 (7.62)
BSC
0.200
(5.08)
BSC
0.075
(1.91)
REF
0.458 (11.63)
0.442 (11.23)
SQ 0.458
(11.63)
MAX
SQ
0.100 (2.54)
0.064 (1.63)
0.088 (2.24)
0.054 (1.37)
C3103b–0–12/99
PRINTED IN U.S.A.