OPA 634 OPA634 OPA635 www.ti.com Wideband, Single-Supply OPERATIONAL AMPLIFIERS TM FEATURES DESCRIPTION The OPA634 and OPA635 are low-power, voltagefeedback, high-speed amplifiers designed to operate on +3V or +5V single-supply voltages. Operation on 5V or +10V supplies is also supported. The input range extends below ground and to within 1.2V of the positive supply. Using complementary common-emitter outputs provides an output swing to within 30mV of ground and 140mV of positive supply. The high output drive current, low differential gain and phase errors make them ideal for single-supply composite video line driving. HIGH BANDWIDTH: 150MHz (G = +2) +3V TO +10V SUPPLY OPERATION ZERO POWER DISABLE (OPA635) INPUT RANGE INCLUDES GROUND 4.8V OUTPUT SWING ON +5V SUPPLY HIGH OUTPUT CURRENT: 80mA HIGH SLEW RATE: 250V/s LOW INPUT VOLTAGE NOISE: 5.6nV/HZ AVAILABLE IN SOT23 PACKAGE APPLICATIONS SINGLE-SUPPLY ADC INPUT BUFFERS SINGLE SUPPLY VIDEO LINE DRIVERS WIRELESS LAN IF AMPLIFIERS CCD IMAGING CHANNELS LOW-POWER ULTRASOUND +3V Disable 2.26k 374 DIS VIN 100 +3V Pwrdn ADS900 10-Bit 20Msps OPA635 Low-distortion operation is ensured by the high gain bandwidth product (140MHz) and slew rate (250V/s). This makes the OPA634 and OPA635 ideal input buffer stages to 3V and 5V CMOS converters. Unlike other low-power, single-supply operational amplifiers, distortion performance improves as the signal swing is decreased. A low 5.6nV input voltage noise supports wide dynamic range operation. Multiplexing or system power reduction can be achieved using the high-speed disable line with the OPA635. Power dissipation can be reduced to zero by taking the disable line HIGH. The OPA634 and OPA635 are available in an industry standard SO-8 package. The OPA634 is also available in an ultra-small SOT23-5 package, while the OPA635 is available in the SOT23-6. Where lower supply current and speed are required, consider the OPA631 and OPA632. 22pF 562 750 RELATED PRODUCTS DESCRIPTION Copyright (c) 1999, Texas Instruments Incorporated SINGLES DUALS Medium Speed, No Disable With Disable OPA631 OPA632 OPA2631 -- High Speed, No Disable With Disable OPA634 OPA635 OPA2634 -- SBOS097A Printed in U.S.A. February, 2001 SPECIFICATIONS: VS = +5V At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 1). OPA634U, N OPA635U, N TYP GUARANTEED CONDITIONS +25C +25C 0C to 70C -40C to +85C UNITS MIN/ MAX TEST LEVEL(1) G = +2, VO 0.5Vp-p G = +5, VO 0.5Vp-p G = +10, VO 0.5Vp-p G +10 VO 0.5Vp-p G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2Vp-p, f = 5MHz f > 1MHz f > 1MHz 150 36 16 140 5 250 2.4 2.4 15 63 5.6 2.8 0.10 0.16 100 24 11 100 -- 170 3.4 3.5 19 56 6.2 3.8 -- -- 84 20 10 82 -- 125 4.7 4.5 22 51 7.3 4.2 -- -- 78 18 8 75 -- 115 5.2 4.8 23 50 7.7 5 -- -- MHz MHz MHz MHz dB V/s ns ns ns dBc nV/Hz pA/Hz % degrees min min min min typ min max max max min max max typ typ B B B B C B B B B B B B C C 66 3 -- 25 0.6 -- 63 7 -- 47 2.25 -- 60 8 -- 57 2.6 -- 53 10 4.6 84 4.5 15 dB mV V/C A A nA/C min max max max max max A A B A A B -0.24 3.8 78 -0.1 3.5 73 -0.05 3.45 71 -0.01 3.4 63 V V dB max min min B A A 10 || 2.1 400 || 1.2 -- -- -- -- -- -- k || pF k || pF typ typ C C Current Output, Sourcing Current Output, Sinking Short-Circuit Current (output shorted to either supply) Closed-Loop Output Impedance G = +2, f 100kHz 0.03 0.1 4.86 4.65 80 100 100 0.2 0.07 0.14 4.8 4.55 50 73 -- -- 0.08 0.15 4.75 4.5 45 59 -- -- 0.09 0.22 4.7 4.4 20 18 -- -- V V V V mA mA mA max max min min min min typ typ A A A A A A C C DISABLE (OPA635 only) On Voltage (device enabled Low) Off Voltage (device disabled High) On Disable Current (DIS pin) Off Disable Current (DIS pin) Disabled Quiescent Current Disable Time Enable Time Off Isolation f = 5MHz, Input to Output 1.0 4.0 70 0 0 100 60 70 1.0 4.1 110 -- 30 -- -- -- 1.0 4.2 120 -- 40 -- -- -- 1.0 4.3 120 -- 50 -- -- -- V V A A A ns ns dB max min max typ max typ typ typ A A A C A C C C Input Referred -- -- 12 12 55 2.7 10.5 12.7 11.3 52 2.7 10.5 13.2 9.75 50 2.7 10.5 13.5 8.5 49 V V mA mA dB min max max min min B A A A A -40 to +85 C typ C 125 150 C/W C/W typ typ C C PARAMETER AC PERFORMANCE (Figure 1) Small-Signal Bandwidth Gain Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Spurious Free Dynamic Range Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase DC PERFORMANCE Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Offset Current Input Offset Current Drift INPUT Least Positive Input Voltage Most Positive Input Voltage Common-Mode Rejection (CMRR) Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power Supply Rejection Ratio (PSRR) THERMAL CHARACTERISTICS Specification: U, N Thermal Resistance U SO-8 N SOT23-5, SOT23-6 VCM = 2.0V VCM = 2.0V Input Referred RL = 1k to 2.5V RL = 150 to 2.5V RL = 1k to 2.5V RL = 150 to 2.5V NOTE: (1) Test Levels: (A) 100% tested at 25C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. 2 OPA634, OPA635 SBOS097A SPECIFICATIONS: VS = +3V At TA = 25C, G = +2 and RL = 150 to VS/2, unless otherwise noted (see Figure 2). OPA634U, N OPA635U, N TYP PARAMETER AC PERFORMANCE (Figure 2) Small-Signal Bandwidth Gain Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Spurious Free Dynamic Range Input Voltage Noise Input Current Noise DC PERFORMANCE Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Offset Current Input Offset Current Drift INPUT Least Positive Input Voltage Most Positive Input Voltage Common-Mode Rejection (CMRR) Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage CONDITIONS +25C +25C 0C to 70C -40C to +85C UNITS G = +2, VO 0.5Vp-p G = +5, VO 0.5Vp-p G = +10, VO 0.5Vp-p G +10 VO 0.5Vp-p 1V Step 0.5V Step 0.5V Step 1V Step VO = 1Vp-p, f = 5MHz f > 1MHz f > 1MHz 110 39 16 150 5 215 2.8 3.0 14 65 5.6 2.8 77 24 12 100 -- 160 4.3 4.4 30 56 6.2 3.7 65 20 10 85 -- 123 4.5 4.6 32 52 7.3 4.2 58 19 8 80 -- 82 6.3 6.0 38 47 7.7 4.4 MHz MHz MHz MHz dB V/s ns ns ns dBc nV/Hz pA/Hz min min min min typ min max max max min max max B B B B C B B B B B B B 67 1.5 -- 25 0.6 -- 61 4 -- 45 2 -- 57 5 -- 59 2.3 -- 53 6 46 64 4 40 dB mV V/C A A nA/C min max max max max max A A B A A B -0.25 1.8 75 -0.1 1.6 65 -0.05 1.55 62 -0.01 1.5 59 V V dB max min min B A A 10 || 2.1 400 || 1.2 -- -- -- -- -- -- k || p k || p typ typ C C 0.035 0.06 2.9 2.8 45 65 100 0.2 0.043 0.16 2.86 2.70 35 30 -- -- 0.045 0.19 2.85 2.69 30 27 -- -- 0.06 0.43 2.45 2.65 12 10 -- -- V V V V mA mA mA max max min min min min typ typ A A A A A A C C f = 5MHz, Input to Output 1.0 1.8 66 0 0 100 60 70 0.5 1.9 100 -- 30 -- -- -- 0.5 2.1 110 -- 40 -- -- -- 0.5 2.2 110 -- 50 -- -- -- V V A A A ns ns dB max min max typ max typ typ typ A A A C A C C C Input Referred -- -- 10.8 10.8 50 2.7 10.5 11.5 10.1 49 2.7 10.5 11.8 8.6 45 2.7 10.5 12 8.0 44 V V mA mA dB min max max min min B A A A A -40 to +85 C typ C 125 150 C/W C/W typ typ C C VCM = 1.0V VCM = 1.0V Input Referred RL = 1k to 1.5V RL = 150 to 1.5V RL = 1k to 1.5V RL = 150 to 1.5V Current Output, Sourcing Current Output, Sinking Short Circuit Current (output shorted to either supply) Closed-Loop Output Impedance Figure 2, f < 100kHz DISABLE (OPA635 only) On Voltage (device enabled Low) Off Voltage (device disabled High) On Disable Current (DIS pin) Off Disable Current (DIS pin) Disabled Quiescent Current Disable Time Enable Time Off Isolation POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power Supply Rejection Ratio (PSRR) THERMAL CHARACTERISTICS Specification: U, N Thermal Resistance U SO-8 N SOT23-5, SOT23-6 GUARANTEED MIN/ TEST MAX LEVEL(1) NOTE: (1) Test Levels: (A) 100% tested at 25C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. OPA634, OPA635 SBOS097A 3 PIN CONFIGURATIONS Top View--OPA634, OPA635 NC 1 8 DIS (OPA635 only) Inverting Input 2 7 +VS Non-Inverting Input 3 6 Output GND 4 5 NC SO Top View--OPA635 Top View--OPA634 Non-Inverting Input 3 4 1 6 +VS GND 2 5 DIS Inverting Input Non-Inverting Input 3 4 Inverting Input 6 SOT23-6 4 6 SOT23-5 A35 1 3 2 B34 1 4 2 Output +VS 3 GND 6 5 1 2 Output Pin Orientation/Package Marking Pin Orientation/Package Marking ELECTROSTATIC DISCHARGE SENSITIVITY ABSOLUTE MAXIMUM RATINGS Power Supply ................................................................................ +11VDC Internal Power Dissipation .................................... See Thermal Analysis Differential Input Voltage .................................................................. 1.2V Input Voltage Range ............................................................... -0.5 to +VS Storage Temperature Range: P, U, N ........................... -40C to +125C Lead Temperature (soldering, 10s) .............................................. +300C Junction Temperature (TJ ) ........................................................... +175C Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown Corporation recommends that all integrated circuits be handled and stored using appropriate ESD protection methods. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER OPA634U SO-8 Surface-Mount 182 -40C to +85C OPA634U " " " " SOT23-5 331 -40C to +85C B34 " " " " SO-8 Surface-Mount 182 -40C to +85C OPA635U " " " " SOT23-6 332 -40C to +85C A35 " " " " " OPA634N " OPA635U " OPA635N " SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER(1) TRANSPORT MEDIA OPA634U OPA634U/2K5 OPA634N/250 OPA634N/3K Rails Tape and Reel Tape and Reel Tape and Reel OPA635U OPA635U/2K5 OPA635N/250 OPA635N/3K Rails Tape and Reel Tape and Reel Tape and Reel NOTE: (1) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 3000 pieces of "OPA635N/3K" will get a single 3000-piece Tape and Reel. 4 OPA634, OPA635 SBOS097A TYPICAL PERFORMANCE CURVES: VS = +5V At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 1). SMALL-SIGNAL FREQUENCY RESPONSE LARGE-SIGNAL FREQUENCY RESPONSE 6 12 VO = 0.2Vp-p 9 0 6 G = +5 -3 Gain (dB) Normalized Gain (dB) VO = 0.2Vp-p G = +2 3 -6 -9 G = +10 -12 3 0 VO = 1Vp-p -3 VO = 2Vp-p -6 VO = 4Vp-p -15 -9 -18 -12 1 10 100 300 1 10 Frequency (MHz) VO = 2Vp-p Input and Output Voltage (500mV/div) Input and Output Voltage (50mV/div) VO = 200mVp-p VO VIN VO VIN Time (10ns/div) Time (10ns/div) DISABLE FEEDTHROUGH vs FREQUENCY LARGE-SIGNAL DISABLE/ENABLE RESPONSE -35 VDIS -45 Feedthrough (dB) VO OPA635 only VDIS = +5V -40 Output Voltage (250mV/div) Disable Voltage (1V/div) 300 LARGE-SIGNAL PULSE RESPONSE SMALL-SIGNAL PULSE RESPONSE VIN = 0.5V OPA635 only 100 Frequency (MHz) -50 -55 -60 -65 -70 -75 -80 -85 Time (50ns/div) 1 10 100 1000 Frequency (MHz) OPA634, OPA635 SBOS097A 5 TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 1). -40 -40 -45 -45 -50 -55 -60 -65 -70 RL = 150 -75 RL = 250 -80 -85 0.1 1 -70 RL = 500 -75 -80 -90 0.1 1 4 Output Voltage (Vp-p) 10MHz 2nd-HARMONIC DISTORTION vs OUTPUT VOLTAGE 10MHz 3rd-HARMONIC DISTORTION vs OUTPUT VOLTAGE -45 3rd-Harmonic Distortion (dBc) -40 -45 -50 -55 RL = 250 -60 -65 RL = 150 -75 RL = 250 -65 4 -80 RL = 500 RL = 500 -50 -55 RL = 250 -60 -65 RL = 150 -70 -75 -80 -85 -90 -90 0.1 1 0.1 4 1 4 Output Voltage (Vp-p) Output Voltage (Vp-p) 20MHz 2nd-HARMONIC DISTORTION vs OUTPUT VOLTAGE 20MHz 3rd-HARMONIC DISTORTION vs OUTPUT VOLTAGE -40 -40 RL = 250 -50 -55 -45 RL = 500 3rd-Harmonic Distortion (dBc) -45 2nd-Harmonic Distortion (dBc) -60 -40 -70 RL = 150 -55 Output Voltage (Vp-p) -85 RL = 150 -60 -65 -70 -75 -80 -85 RL = 500 -50 -55 RL = 250 -60 -65 RL = 150 -70 -75 -80 -85 -90 -90 0.1 1 Output Voltage (Vp-p) 6 -50 -85 RL = 500 -90 2nd-Harmonic Distortion (dBc) 5MHz 3rd-HARMONIC DISTORTION vs OUTPUT VOLTAGE 3rd-Harmonic Distortion (dBc) 2nd-Harmonic Distortion (dBc) 5MHz 2nd-HARMONIC DISTORTION vs OUTPUT VOLTAGE 4 0.1 1 4 Output Voltage (Vp-p) OPA634, OPA635 SBOS097A TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 1). 3rd-HARMONIC DISTORTION vs FREQUENCY 2nd-HARMONIC DISTORTION vs FREQUENCY -40 VO = 2Vp-p RL = 100 -45 VO = 2Vp-p RL = 100 -45 3rd-Harmonic Distortion (dBc) 2nd-Harmonic Distortion (dBc) -40 -50 -55 -60 -65 -70 G = +2 -75 -80 G = +5 -85 -50 -55 -60 -65 -70 G = +2 -75 G = +5 -80 G = +10 -85 G = +10 -90 -90 1 10 1 20 10 TWO-TONE, 3rd-ORDER INTERMODULATION SPURIOUS HARMONIC DISTORTION vs LOAD RESISTANCE -40 -40 Harmonic Distortion (dBc) -50 3rd-Order Spurious Level (dBc) VO = 2Vp-p fO = 5MHz -45 3rd-Harmonic Distortion -55 -60 -65 -70 -75 -80 2nd-Harmonic Distortion -85 -90 fO = 20MHz -45 -50 fO = 10MHz -55 -60 -65 -70 -75 fO = 5MHz -80 Load Power at Matched 50 Load -85 -90 100 200 300 400 500 -16 -14 -12 RL () -10 -8 -6 -4 -2 0 Single-Tone Load Power (dBm) CMRR AND PSRR vs FREQUENCY INPUT NOISE DENSITY vs FREQUENCY 80 100 CMRR 75 70 Voltage Noise (nV/Hz) Current Noise (pA/Hz) Rejection Ratio, Input Referred (dB) 20 Frequency (MHz) Frequency (MHz) 65 PSRR 60 55 50 45 40 10 Voltage Noise, eni = 5.6nV/Hz Current Noise, ini = 2.8pA/Hz 35 30 1 100 1k 10k 100k Frequency (Hz) OPA634, OPA635 SBOS097A 1M 10M 100 1k 10k 100k 1M 10M Frequency (Hz) 7 TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 1). RS vs CAPACITIVE LOAD 1000 FREQUENCY RESPONSE vs CAPACITIVE LOAD 2 CL = 1000pF 1 VO = 0.2Vp-p Normalized Gain (dB) 0 RS () 100 10 CL = 10pF -1 CL = 100pF -2 -3 -4 RS -5 OPA63x VO CL -6 -7 1 100 1000 CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY 100k 1M 10M 100M G = +1 RF = 25 10 1 0.1 1G 1k 10k 100k Input Offset Voltage 4.0 40 3.5 35 3.0 30 25 Input Bias Current 2.0 20 1.5 15 10X Input Offset Current 10 0.5 5 0.0 0 20 40 Temperature (C) 8 60 80 100 160 Sinking Output Current 14 Power-Supply Current (mA) 45 Input Bias Current (A) 4.5 0 100M 16 10X Input Offset Current (A) Input Offset Voltage (mV) 50 -20 10M SUPPLY AND OUTPUT CURRENT vs TEMPERATURE INPUT DC ERRORS vs TEMPERATURE -40 1M Frequency (Hz) 5.0 2.5 300 100 Output Impedance () 0 -30 -60 -90 -120 -150 -180 -210 -240 -270 -300 -330 -360 Frequency (Hz) 1.0 100 OPEN-LOOP GAIN AND PHASE Open-Loop Gain 10k 10 Frequency (MHz) Open-Loop Phase 1k 1 Capacitive Load (pF) 140 Quiescent Supply Current 12 120 10 100 Sourcing Output Current 8 80 6 60 4 40 2 20 0 0 -40 -20 0 20 40 60 80 100 Temperature (C) OPA634, OPA635 SBOS097A Output Current (mA) 10 Open-Loop Phase () Open-Loop Gain (dB) +VS/2 -8 1 100 90 80 70 60 50 40 30 20 10 0 -10 -20 1k TYPICAL PERFORMANCE CURVES: VS = +3V At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 2). LARGE-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 12 6 VO = 0.2Vp-p VO = 0.2Vp-p G = +2 9 6 0 G = +5 -3 Gain (dB) Normalized Gain (dB) 3 -6 -9 G = +10 -12 3 VO = 1Vp-p 0 VO = 2Vp-p -3 -6 -9 -15 -12 -18 1 10 100 1 300 10 2nd-HARMONIC DISTORTION vs FREQUENCY -40 VO = 1Vp-p RL = 100 -50 -55 -60 -65 -70 G = +2 -75 G = +5 -80 -85 VO = 1Vp-p RL = 100 -45 3rd-Harmonic Distortion (dBc) -45 2nd-Harmonic Distortion (dBc) 300 3rd-HARMONIC DISTORTION vs FREQUENCY -40 -50 -55 -60 -65 G = +2 -70 G = +5 -75 G = +10 -80 -85 G = +10 -90 -90 1 10 20 1 Frequency (MHz) HARMONIC DISTORTION vs LOAD RESISTANCE TWO-TONE, 3rd-ORDER INTERMODULATION SPURIOUS -40 -50 3rd-Order Spurious Level (dBc) VO = 1Vp-p fO = 5MHz -45 10 Frequency (MHz) -40 Harmonic Distortion (dBc) 100 Frequency (MHz) Frequency (MHz) 3rd-Harmonic Distortion -55 -60 -65 -70 -75 -80 2nd-Harmonic Distortion -85 -90 20 fO = 20MHz -45 -50 -55 -60 fO = 10MHz -65 -70 -75 -80 fO = 5MHz -85 Load Power at Matched 50 Load -90 100 200 300 RL () OPA634, OPA635 SBOS097A 400 500 -16 -14 -12 -10 -8 -6 -4 Single-Tone Load Power (dBm) 9 TYPICAL PERFORMANCE CURVES: VS = +3V (Cont.) At TA = 25C, G = +2, RF = 750, and RL = 150 to VS/2, unless otherwise noted (see Figure 2). FREQUENCY RESPONSE vs CAPACITIVE LOAD RS vs CAPACITIVE LOAD 1000 2 CL = 1000pF 1 VO = 0.2Vp-p Normalized Gain (dB) 0 RS () 100 10 CL = 10pF -1 CL = 100pF -2 -3 -4 RS -5 OPA63x VO CL -6 -7 1k +VS/2 -8 1 1 10 100 1 1000 10 100 300 Frequency (MHz) Capacitive Load (pF) OUTPUT SWING vs LOAD RESISTANCE 3.0 1.0 0.9 Left Scale 2.8 Maximum VO 2.7 0.8 0.7 2.6 0.6 2.5 0.5 2.4 0.4 2.3 Right Scale 2.2 Minimum VO 2.1 2.0 50 0.3 0.2 Minimum Output Voltage (V) Maximum Output Voltage (V) 2.9 0.1 0.0 1000 100 RL () 10 OPA634, OPA635 SBOS097A APPLICATIONS INFORMATION WIDEBAND VOLTAGE FEEDBACK OPERATION The OPA634 and OPA635 are unity-gain stable, very highspeed voltage feedback op amps designed for single supply operation (+3V to +10V). The input stage supports input voltages below ground, and to within 1.2V of the positive supply. The complementary common-emitter output stage provides an output swing to within 30mV of ground and 140mV of the positive supply. They are compensated to provide stable operation with a wide range of resistive loads. The OPA635's internal disable circuitry is designed to minimize supply current when disabled. Figure 1 shows the AC-coupled, gain of +2 configuration used for the +5V Specifications and Typical Performance Curves. For test purposes, the input impedance is set to 50 with a resistor to ground. Voltage swings reported in the Specifications are taken directly at the input and output pins. For the circuit of Figure 1, the total effective load on the output at high frequencies is 150 || 1500. The disable pin needs to be driven by a low impedance source, such as a CMOS inverter. The 1.50k resistors at the non-inverting input provide the common-mode bias voltage. Their parallel combination equals the DC resistance at the inverting input, (RF), minimizing the DC offset. Figure 2 shows the DC-coupled, gain of +2 configuration used for the +3V Specifications and Typical Performance Curves. For test purposes, the input impedance is set to 50 with a resistor to ground. Though not strictly a "rail-to-rail" design, these parts come very close, while maintaining excellent performance. They will deliver 2.8Vp-p on a single +3V supply with 70MHz bandwidth. The 374 and 2.26k resistors at the input level-shift VIN so that VOUT is within the allowed output voltage range when VIN = 0. See the typical performance curves for information on driving capacitive loads. +VS = 3V 6.8F + 2.26k 0.1F 374 VIN DIS (OPA635 only) 57.6 VOUT OPA63x RL 150 RG 562 RF 750 +VS 2 +VS = 5V 6.8F + FIGURE 2. DC-Coupled Signal--Resistive Load to Supply Midpoint. 0.1F 1.50k 0.1F VIN DIS (OPA635 only) 53.6 1.50k VOUT OPA63x RL 150 z 0.1F RG 750 RF 750 +VS 2 FIGURE 1. AC-Coupled Signal--Resistive Load to Supply Midpoint. OPA634, OPA635 SBOS097A SINGLE SUPPLY ADC CONVERTER INTERFACE The front page shows a DC-coupled, single-supply ADC (Analog-to-Digital Converter) driver circuit. Many systems are now requiring +3V supply capability of both the ADC and its driver. The OPA635 provides excellent performance in this demanding application. Its large input and output voltage ranges, and low distortion, support converters such as the ADS900 shown in this figure. The input level-shifting circuitry was designed so that VIN can be between 0V and 0.5V, while delivering an output voltage of 1V to 2V for the ADS900. Both the OPA635 and ADS900 have power reduction pins with the same polarity for those systems that need to conserve power. 11 DC LEVEL SHIFTING Figure 3 shows a DC-coupled non-inverting amplifier that level-shifts the input up to accommodate the desired output voltage range. Given the desired signal gain (G), and the amount VOUT needs to be shifted up (VOUT) when VIN is at the center of its range, the following equations give the resistor values that produce the desired performance. Start by setting R4 between 200 and 1.5k. NG = G + VOUT/VS R1 = R4/G NON-INVERTING AMPLIFIER WITH REDUCED PEAKING Figure 4 shows a non-inverting amplifier that reduces peaking at low gains. The resistor RC compensates the OPA634 or OPA635 to have higher Noise Gain (NG), which reduces the AC response peaking (typically 5dB at G = +1 without RC) without changing the DC gain. VIN needs to be a low impedance source, such as an op amp. The resistor values are low to reduce noise. Using both RT and RF helps minimize the impact of parasitic impedances. R2 = R4/(NG - G) R3 = R4/(NG -1) RT where: VIN NG = 1 + R4/R3 RC VOUT = (G)VIN + (NG - G)VS VOUT OPA63x Make sure that VIN and VOUT stay within the specified input and output voltage ranges. RG RF +VS R2 FIGURE 4. Compensated Non-Inverting Amplifier. R1 VIN OPA63x R3 VOUT R4 The Noise Gain can be calculated as follows: G1 = 1 + RF RG G2 = 1 + R T + R F / G1 RC NG = G1G 2 FIGURE 3. DC Level-Shifting Circuit. The front page circuit is a good example of this type of application. It was designed to take VIN between 0V and 0.5V, and produce VOUT between 1V and 2V, when using a +3V supply. This means G = 2.00, and VOUT = 1.50V - G * 0.25V = 1.00V. Plugging into the above equations (with R4 = 750) gives: NG = 2.33, R1 = 375, R2 = 2.25k, and R3 = 563. The resistors were changed to the nearest standard values. 12 A unity gain buffer can be designed by selecting RT = RF = 20.0 and RC = 40.2 (do not use RG ). This gives a Noise Gain of 2, so its response will be similar to the Characteristics Plots with G = +2. Decreasing RC to 20.0 will increase the Noise Gain to 3, which typically gives a flat frequency response, but with less bandwidth. The circuit in Figure 1 can be redesigned to have less peaking by increasing the noise gain to 3. This is accomplished by adding RC = 2.55k between the op amps inputs. OPA634, OPA635 SBOS097A DESIGN-IN TOOLS DEMONSTRATION BOARDS Two PC boards are available to assist in the initial evaluation of circuit performance using the OPA634 and OPA635 in their three package styles. These are available free as an unpopulated PC board delivered with descriptive documentation. The summary information for these boards is shown in Table I. PRODUCT PACKAGE OPA63xU OPA63xN SO-8 SOT23-5 SOT23-6 " BOARD PART NUMBER LITERATURE REQUEST NUMBER DEM-OPA68xU DEM-OPA6xxN MKT-351 MKT-348 " " TABLE I. Demo Board Summary Information. Contact the Texas Instruments Technical Applications Support Line at 1-972-644-5580 to request any of these boards. OPERATING SUGGESTIONS OPTIMIZING RESISTOR VALUES Since the OPA634 and OPA635 are voltage feedback op amps, a wide range of resistor values may be used for the feedback and gain setting resistors. The primary limits on these values are set by dynamic range (noise and distortion) and parasitic capacitance considerations. For a non-inverting unity gain follower application, the feedback connection should be made with a 20 resistor, not a direct short (see Figure 4 with RG = ). This will isolate the inverting input capacitance from the output pin and improve the frequency response flatness. Usually, for G > 1 application, the feedback resistor value should be between 200 and 1.5k. Below 200, the feedback network will present additional output loading which can degrade the harmonic distortion performance. Above 1.5k, the typical parasitic capacitance (approximately 0.2pF) across the feedback resistor may cause unintentional band-limiting in the amplifier response. A good rule of thumb is to target the parallel combination of RF and RG (Figure 1) to be less than approximately 400. The combined impedance RF || RG interacts with the inverting input capacitance, placing an additional pole in the feedback network and thus, a zero in the forward response. Assuming a 3pF total parasitic on the inverting node, holding RF || RG <400 will keep this pole above 130MHz. By itself, this constraint implies that the feedback resistor RF can increase to several k at high gains. This is acceptable as long as the pole formed by RF and any parasitic capacitance appearing in parallel is kept out of the frequency range of interest. BANDWIDTH VERSUS GAIN: NON-INVERTING OPERATION Voltage feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory, this relationship is described by the Gain Bandwidth Product (GBP) shown in the specifications. Ideally, dividing GBP by the non-inverting signal gain (also called the Noise Gain, or NG) will predict the closed-loop bandwidth. In practice, this only holds true when the phase margin approaches 90, as it does in high gain configurations. At low gains (increased feedback factors), most amplifiers will exhibit a more complex response with lower phase margin. The OPA634 and OPA635 are compensated to give a slightly peaked response in a non-inverting gain of 2 (Figure 1). This results in a typical gain of +2 bandwidth of 150MHz, far exceeding that predicted by dividing the 140MHz GBP by 2. Increasing the gain will cause the phase margin to approach 90 and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +10, the 16MHz bandwidth shown in the Typical Specifications is close to that predicted using the simple formula and the typical GBP. The OPA634 and OPA635 exhibit minimal bandwidth reduction going to +3V single supply operation as compared with +5V supply. This is because the internal bias control circuitry retains nearly constant quiescent current as the total supply voltage between the supply pins is changed. INVERTING AMPLIFIER OPERATION Since the OPA634 and OPA635 are general-purpose, wideband voltage- feedback op amps, all of the familiar op amp application circuits are available to the designer. Figure 5 shows a typical inverting configuration where the I/O impedances and signal gain from Figure 1 are retained in an inverting circuit configuration. Inverting operation is one of the more common requirements and offers several performance benefits. The inverting configuration shows improved slew rate and distortion. It also allows the input to be biased at VS/2 without any headroom issues. The output voltage can be independently moved to be within the output voltage range with coupling capacitors, or bias adjustment resistors. +5V + 0.1F 2RT 1.5k 0.1F 50 Source 2RT 1.5k RG 0.1F 374 DIS 6.8F RO 50 OPA63x 50 Load RF 750 RM 57.6 FIGURE 5. AC-Coupled, Gain of -2 Example Circuit. OPA634, OPA635 SBOS097A 13 In the inverting configuration, three key design consideration must be noted. The first is that the gain resistor (RG) becomes part of the signal channel input impedance. If input impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long PC board trace, or other transmission line conductor), RG may be set equal to the required termination value and RF adjusted to give the desired gain. This is the simplest approach and results in optimum bandwidth and noise performance. However, at low inverting gains, the resultant feedback resistor value can present a significant load to the amplifier output. For an inverting gain of 2, setting RG to 50 for input matching eliminates the need for RM but requires a 100 feedback resistor. This has the interesting advantage of the noise gain becoming equal to 2 for a 50 source impedance--the same as the non-inverting circuits considered above. However, the amplifier output will now see the 100 feedback resistor in parallel with the external load. In general, the feedback resistor should be limited to the 200 to 1.5k range. In this case, it is preferable to increase both the RF and RG values, as shown in Figure 5, and then achieve the input matching impedance with a third resistor (RM) to ground. The total input impedance becomes the parallel combination of RG and RM. The second major consideration, touched on in the previous paragraph, is that the signal source impedance becomes part of the noise gain equation and hence influences the bandwidth. For the example in Figure 5, the RM value combines in parallel with the external 50 source impedance (at high frequencies), yielding an effective driving impedance of 50 || 576 = 26.8. This impedance is added in series with RG for calculating the noise gain. The resultant is 2.87 for Figure 5, as opposed to only 2 if RM could be eliminated as discussed above. The bandwidth will therefore be lower for the gain of -2 circuit of Figure 5 (NG = +2.87) than for the gain of +2 circuit of Figure 1. The third important consideration in inverting amplifier design is setting the bias current cancellation resistors on the non-inverting input (a parallel combination of RT = 750). If this resistor is set equal to the total DC resistance looking out of the inverting node, the output DC error, due to the input bias currents, will be reduced to (Input Offset Current) times RF. With the DC blocking capacitor in series with RG, the DC source impedance looking out of the inverting mode is simply RF = 750 for Figure 5. To reduce the additional high-frequency noise introduced by this resistor, and powersupply feedthrough, RT is bypassed with a capacitor. As long as RT < 400, its noise contribution will be minimal. As a minimum, the OPA634 and OPA635 require an RT value of 50 to damp out parasitic-induced peaking--a direct short to ground on the non-inverting input runs the risk of a very high-frequency instability in the input stage. 14 OUTPUT CURRENT AND VOLTAGE The OPA634 and OPA635 provide outstanding output voltage capability. Under no-load conditions at +25C, the output voltage typically swings closer than 140mV to either supply rail; the guaranteed over temperature swing is within 300mV of either rail (VS = +5V). The minimum specified output voltage and current specifications over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold start-up will the output current and voltage decrease to the numbers shown in the guaranteed tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBE's (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the available output voltage and current will always be greater than that shown in the over-temperature specifications, since the output stage junction temperatures will be higher than the minimum specified operating ambient. To maintain maximum output stage linearity, no output short-circuit protection is provided. This will not normally be a problem, since most applications include a series matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to the adjacent positive power-supply pin (8-pin packages) will, in most cases, destroy the amplifier. If additional short-circuit protection is required, consider a small series resistor in the power-supply leads. This will, under heavy output loads, reduce the available output voltage swing. DRIVING CAPACITIVE LOADS One of the most demanding and yet very common load conditions for an op amp is capacitive loading. Often, the capacitive load is the input of an ADC--including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA634 and OPA635 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. The Typical Performance Curves show the recommended RS versus capacitive load and the resulting frequency response at the load. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA634 and OPA635. Long PC board traces, unmatched cables, and connections to multiple devices can easily exceed this value. Always consider this effect carefully, and add the recommended series resistor as close as possible to the output pin (see Board Layout Guidelines section). OPA634, OPA635 SBOS097A The criterion for setting this RS resistor is a maximum bandwidth, flat frequency response at the load. For a gain of +2, the frequency response at the output pin is already slightly peaked without the capacitive load, requiring relatively high values of RS to flatten the response at the load. Increasing the noise gain will also reduce the peaking (see Figure 4). The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 1 shows the general form for the output noise voltage using the terms shown in Figure 6: (1) DISTORTION PERFORMANCE Dividing this expression by the noise gain (NG = (1 + RF/RG)) will give the equivalent input-referred spot noise voltage at the non-inverting input, as shown in Equation 2: (2) The OPA634 and OPA635 provide good distortion performance into a 150 load. Relative to alternative solutions, it provides exceptional performance into lighter loads and/or operating on a single +3V supply. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd harmonic will dominate the distortion with a negligible 3rd harmonic component. Focusing then on the 2nd harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network; in the non-inverting configuration (Figure 1) this is sum of RF + RG, while in the inverting configuration, only RF needs to be included in parallel with the actual load. NOISE PERFORMANCE High slew rate, unity gain stable, voltage-feedback op amps usually achieve their slew rate at the expense of a higher input noise voltage. The 5.6nV/Hz input voltage noise for the OPA634 and OPA635 is, however, much lower than comparable amplifiers. The input-referred voltage noise, and the two input-referred current noise terms (2.8pA/Hz), combine to give low output noise under a wide variety of operating conditions. Figure 6 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/Hz or pA/Hz. ENI EO OPA63x RS IBN RF 4kT RG RG FIGURE 6. Noise Analysis Model. OPA634, OPA635 SBOS097A IBI (E NI 2 ) + ( I BN R S ) + 4kTR S NG 2 + ( I BI R F ) + 4kTR F NG 2 2 I R 2 4kTR F 2 E N = E NI 2 + ( I BN R S ) + 4kTR S + BI F + NG NG Evaluating these two equations for the circuit and component values shown in Figure 1 will give a total output spot noise voltage of 12.5nV/Hz and a total equivalent input spot noise voltage of 6.3nV/Hz. This is including the noise added by the resistors. This total input-referred spot noise voltage is not much higher than the 5.6nV/Hz specification for the op amp voltage noise alone. This will be the case as long as the impedances appearing at each op amp input are limited to the previously recommend maximum value of 400, and the input attenuation is low. DC ACCURACY AND OFFSET CONTROL The balanced input stage of a wideband voltage-feedback op amp allows good output DC accuracy in a wide variety of applications. The power-supply current trim for the OPA634 and OPA635 gives even tighter control than comparable products. Although the high-speed input stage does require relatively high input bias current (typically 25A out of each input terminal), the close matching between them may be used to reduce the output DC error caused by this current. This is done by matching the DC source resistances appearing at the two inputs. Evaluating the configuration of Figure 1 (which has matched DC input resistances), using worst-case +25C input offset voltage and current specifications, gives a worst-case output offset voltage equal to: (NG = non-inverting signal gain at DC) (NG * VOS(MAX)) (RF * IOS(MAX)) = (1 * 7.0mV) (750 * 2.25A) = 8.7mV ERS 4kTRS EO = 4kTRF 4kT = 1.6E -20J at 290K A fine scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are available for introducing DC offset control into an op amp circuit. Most of these techniques are based on adding a DC current through the feedback resistor. In selecting an offset trim method, one key consideration is the impact on the desired signal path frequency response. If the signal path is 15 intended to be non-inverting, the offset control is best applied as an inverting summing signal to avoid interaction with the signal source. If the signal path is intended to be inverting, applying the offset control to the non-inverting input may be considered. Bring the DC offsetting current into the inverting input node through resistor values that are much larger than the signal path resistors. This will insure that the adjustment circuit has minimal effect on the loop gain and hence the frequency response. output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for resistive load connected to mid-supply (VS/2), be at a maximum when the output is fixed at a voltage equal to VS/4 or 3VS/4. Under this condition, PDL = VS2/(16 * RL), where RL includes feedback network loading. DISABLE OPERATION As a worst-case example, compute the maximum TJ using an OPA635 (SOT23-6 package) in the circuit of Figure 1 operating at the maximum specified ambient temperature of +85C and driving a 150 load at mid-supply. The OPA635 provides a disable feature that may be used either to reduce system power or to implement a simple channel multiplexing operation. To disable, the control pin must be asserted HIGH. Figure 7 shows a simplified internal circuit for the disable control feature. Note that it is the power in the output stage, and not into the load, that determines internal power dissipation. PD = 10V * 13.5mA + 52/(16 * (150 || 1500)) = 144mW Maximum TJ = +85C + (0.14W * 150C/W) = 107C. Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower guaranteed junction temperatures. The highest possible internal dissipation will occur if the load requires current to be forced into the output at high output voltages or sourced from the output at low output voltages. This puts a high current through a large internal voltage drop in the output transistors. +VS Q1 50k IS Control VDIS FIGURE 7. Simplified Disable Control Circuit (OPA635). In normal operation, base current to Q1 is provided through the 50k resistor. One key parameter in disable operation is the output glitch when switching in and out of the disabled mode. The transition edge rate (dv/dt) of the DIS control line will influence this glitch. Adding a simple RC filter into the DIS pin from a higher speed logic line will reduce the glitch. If extremely fast transition logic is used, a 1k series resistor will provide adequate bandlimiting using just the parasitic input capacitance on the DIS pin while still ensuring adequate logic level swing. THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation, as described below. In no case should the maximum junction temperature be allowed to exceed 175C. Operating junction temperature (TJ) is given by TA + PD*JA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the 16 BOARD LAYOUT GUIDELINES Achieving optimum performance with a high frequency amplifier like the OPA634 and OPA635 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (<0.25") from the power-supply pins to high frequency 0.1F decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. Each power-supply connection should always be decoupled with one of these capacitors. An optional supply decoupling capacitor (0.1F) across the two power supplies (for bipolar operation) will improve 2ndharmonic distortion performance. Larger (2.2F to 6.8F) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. OPA634, OPA635 SBOS097A c) Careful selection and placement of external components will preserve the high-frequency performance. Resistors should be a very low reactance type. Surfacemount resistors work best and allow a tighter overall layout. Metal film or carbon composition axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board traces as short as possible. Never use wirewound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as non-inverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal film or surfacemount resistors have approximately 0.2pF in shunt with the resistor. For resistor values >1.5k, this parasitic capacitance can add a pole and/or zero below 500MHz that can effect circuit operation. Keep resistor values as low as possible consistent with load driving considerations. The 750 feedback used in the typical performance specifications is a good starting point for design d) Connections to other wideband devices on the board may be made with short direct traces or through on-board transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the typical performance curve "Recommended RS vs Capacitive Load". Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA634 and OPA635 are nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin) If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50 environment is normally not necessary on board, and in fact, a higher impedance environment will improve distortion as shown in the distortion versus load plots. With a characteristic board trace impedance defined (based on board material and trace dimensions), a matching series resistor into the trace from the output of the OPA634 and OPA635 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating OPA634, OPA635 SBOS097A impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the typical performance curve "Recommended RS vs Capacitive Load". This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high-speed part is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA634 and OPA635 onto the board. If socketing for the DIP package is desired, high frequency flush mount pins (e.g., McKenzie Technology #710C) can give good results. INPUT AND ESD PROTECTION The OPA634 and OPA635 are is built using a very highspeed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 8. +V CC External Pin Internal Circuitry -V CC FIGURE 8. Internal ESD Protection. These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (e.g., in systems with 15V supply parts driving into the OPA634 and OPA635), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible, since high values degrade both noise performance and frequency response. 17 IMPORTANT NOTICE Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgment, including those pertaining to warranty, patent infringement, and limitation of liability. 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