LTC3129
1
3129fc
For more information www.linear.com/LTC3129
Typical applicaTion
FeaTures DescripTion
15V, 200mA Synchronous
Buck-Boost DC/DC Converter
with 1.3µA Quiescent Current
The LT C
®
3129 is a high efficiency, 200mA buck-boost
DC/DC converter with a wide VIN and VOUT range. It includes
an accurate RUN pin threshold to allow predictable regula-
tor turn-on and a maximum power point control (MPPC)
capability that ensures maximum power extraction from
non-ideal power sources such as photovoltaic panels.
The LTC3129 employs an ultralow noise, 1.2MHz PWM
switching architecture that minimizes solution footprint by
allowing the use of tiny, low profile inductors and ceramic
capacitors. Built-in loop compensation and soft-start
simplify the design. For high efficiency operation at light
loads, automatic Burst Mode operation can be selected,
reducing the quiescent current to just 1.3µA.
Additional features include a power good output, less than
10nA of shutdown current and thermal shutdown.
The LTC3129 is available in thermally enhanced 3mm ×
3mm QFN and 16-lead MSOP packages. For fixed output
voltage options, see the functionally equivalent LTC3129-1,
which eliminates the need for an external feedback divider.
L, LT , LT C , LT M , Linear Technology, the Linear logo and Burst Mode are registered trademarks
and PowerPath is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
applicaTions
n Regulates VOUT Above, Below or Equal to VIN
n Wide VIN Range: 2.42V to 15V, 1.92V to 15V After
Start-Up (Bootstrapped)
n Wide VOUT Range: 1.4V to 15.75V
n 200mA Output Current in Buck Mode
n Single Inductor
n 1.3µA Quiescent Current
n Programmable Maximum Power Point Control
n 1.2MHz Ultralow Noise PWM
n Current Mode Control
n Pin Selectable Burst Mode
®
Operation
n Up to 95% Efficiency
n Accurate RUN Pin Threshold
n Power Good Indicator
n 10nA Shutdown Current
n Thermally Enhanced 3mm × 3mm QFN and
16-Lead MSOP Packages
n Industrial Wireless Sensor Nodes
n Post-Regulator for Harvested Energy
n Solar Panel Post-Regulator/Charger
n Intrinsically Safe Power Supplies
n Wireless Microphones
n Avionics-Grade Wireless Headsets
BST1
VOUT V
OUT
SW1 SW2
LTC3129
22nF
BST2
PGOOD
GND
FB
VCC
VIN
2.42V
TO 15V
V
IN
VCC
RUN
MPPC
PWM
22nF
10µH
10µF
10µF 10pF
2.2µF
3129 TA01a
PGND
3.32M
1.02M
5V AT 200mA, VIN > 5V
5V AT 100mA, VIN < 5V
Efficiency and Power Loss vs Load
100
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
POWER LOSS (mW)
70
80
90
50
30
20
60
40
10
0
1000
100
0.1
10
1
0.01
3129 TA01b
0.1 100 1000101
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 15V
EFFICIENCY
POWER LOSS
VOUT = 5V
LTC3129
2
3129fc
For more information www.linear.com/LTC3129
absoluTe MaxiMuM raTings
VIN, VOUT Voltages .................................... –0.3V to 18V
SW1 DC Voltage ............................ –0.3V to (VIN + 0.3V)
SW2 DC Voltage..........................–0.3V to (VOUT + 0.3V)
SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V
BST1 Voltage ....................(SW1 – 0.3V) to (SW1 + 6V)
BST2 Voltage ....................(SW2 – 0.3V) to (SW2 + 6V)
RUN, PGOOD Voltages ............................... –0.3V to 18V
VCC, FB, PWM, MPPC Voltages .................... –0.3V to 6V
PGOOD Sink Current .............................................15mA
Operating Junction Temperature Range
(Notes 2, 5) ............................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
MSE Lead Temperature (Soldering, 10 sec) .......... 300°C
(Notes 1, 8)
16 15 14 13
5 6 7 8
TOP VIEW
UD PACKAGE
16-LEAD (3mm × 3mm) PLASTIC QFN
9
10
11
12
4
3
2
1BST1
VIN
VCC
RUN
VOUT
PGOOD
PWM
NC
SW1
PGND
SW2
BST2
MPPC
GND
FB
NC
17
PGND
TJMAX = 125°C, θJC = 7.5°C/W, θJA = 68°C/W (NOTE 6)
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
1
2
3
4
5
6
7
8
VCC
RUN
MPPC
GND
FB
NC
NC
PWM
16
15
14
13
12
11
10
9
VIN
BST1
SW1
PGND
SW2
BST2
VOUT
PGOOD
TOP VIEW
MSE PACKAGE
16-LEAD PLASTIC MSOP
17
PGND
TJMAX = 125°C, θJC = 10°C/W, θJA = 40°C/W (NOTE 6)
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
pin conFiguraTion
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3129EUD#PBF LTC3129EUD#TRPBF LGDR 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3129IUD#PBF LTC3129IUD#TRPBF LGDR 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3129EMSE#PBF LTC3129EMSE#TRPBF 3129 16-Lead Plastic MSOP –40°C to 125°C
LTC3129IMSE#PBF LTC3129IMSE#TRPBF 3129 16-Lead Plastic MSOP –40°C to 125°C
Consult LT C Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
LTC3129
3
3129fc
For more information www.linear.com/LTC3129
elecTrical characTerisTics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
PARAMETER CONDITIONS MIN TYP MAX UNITS
VIN Start-Up Voltage l2.25 2.42 V
Input Voltage Range VCC > 2.42V (Back-Driven) l1.92 15 V
VIN UVLO Threshold (Rising) VCC > 2.42V (Back-Driven) l1.8 1.9 2.0 V
VIN UVLO Hysteresis l80 100 130 mV
Output Voltage Adjust Range l1.4 15.75 V
Feedback Voltage l1.151 1.175 1.199 V
Feedback Input Current FB = 1.25V 0.1 10 nA
Quiescent Current (VIN) – Shutdown RUN = 0V, Including Switch Leakage 10 100 nA
Quiescent Current (VIN) UVLO Either VIN or VCC Below Their UVLO Threshold, or
RUN Below the Threshold to Enable Switching
1.9 3 µA
Quiescent Current – Burst Mode Operation Measured on VIN, FB > 1.25V
PWM = 0V, RUN = VIN
1.3 2.0 µA
N-Channel Switch Leakage on VIN and VOUT SW1 = 0V, VIN = 15V
SW2 = 0V, VOUT = 15V
RUN = 0V
10 50 nA
N-Channel Switch On-Resistance VCC = 4V 0.75 Ω
Inductor Average Current Limit VOUT > UV Threshold (Note 4)
VOUT < UV Threshold (Note 4)
l
l
220
80
275
130
350
200
mA
mA
Inductor Peak Current Limit (Note 4) l400 500 680 mA
Maximum Boost Duty Cycle FB = 1.10V. Percentage of Period SW2 is Low in
Boost Mode (Note 7)
l85 89 95 %
Minimum Duty Cycle FB = 1.25V. Percentage of Period SW1 is High in
Buck Mode (Note 7)
l0 %
Switching Frequency PWM = VCC l1.0 1.2 1.4 MHz
SW1 and SW2 Minimum Low Time (Note 3) 90 ns
MPPC Voltage l1.12 1.175 1.22 V
MPPC Input Current MPPC = 5V 1 10 nA
RUN Threshold to Enable VCC l0.5 0.9 1.15 V
RUN Threshold to Enable Switching (Rising) VCC > 2.4V l1.16 1.22 1.28 V
RUN (Switching) Threshold Hysteresis 50 80 120 mV
RUN Input Current RUN = 15V 1 10 nA
PWM Input High l1.6 V
PWM Input Low l0.5 V
PWM Input Current PWM = 5V 0.1 1 µA
Soft-Start Time 3 ms
VCC Voltage VIN > 4.85V l3.4 4.1 4.7 V
VCC Dropout Voltage (VIN – VCC) VIN = 3.0V, Switching
VIN = 2.0V (VCC in UVLO)
35
0
60
2
mV
mV
VCC UVLO Threshold (Rising) l2.1 2.25 2.42 V
VCC UVLO Hysteresis 60 mV
VCC Current Limit VCC = 0V l4 20 60 mA
VCC Back-Drive Voltage (Maximum) l5.5 V
VCC Input Current (Back-Driven) VCC = 5.5V (Switching) 2 4 mA
VCC Leakage to VIN if VCC > VIN VCC = 5.5V, VIN = 1.8V, Measured on VIN –27 µA
VOUT UV Threshold (Rising) l0.95 1.15 1.35 V
LTC3129
4
3129fc
For more information www.linear.com/LTC3129
elecTrical characTerisTics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3129 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3129E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3129I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The junction temperature (TJ) is calculated from the
ambient temperature (TA) and power dissipation (PD) according to the
formula: TJ = TA + (PD • θJA°C/W), where θJA is the package thermal
impedance. Note that the maximum ambient temperature consistent
with these specifications, is determined by specific operating conditions
in conjunction with board layout, the rated thermal package thermal
resistance and other environmental factors.
Note 3: Specification is guaranteed by design and not 100% tested in
production.
PARAMETER CONDITIONS MIN TYP MAX UNITS
VOUT UV Hysteresis 150 mV
VOUT Current – Shutdown RUN = 0V, VOUT = 15V Including Switch Leakage 10 100 nA
VOUT Current – Sleep PWM = 0V, FB = 1.25V 10 nA
VOUT Current – Active PWM = VCC, VOUT = 15V (Note 4), FB = 1.25V 5 9 µA
PGOOD Threshold, Falling Referenced to Programmed VOUT Voltage –5.5 –7.5 –10 %
PGOOD Hysteresis Referenced to Programmed VOUT Voltage 2.5 %
PGOOD Voltage Low ISINK = 1mA 250 300 mV
PGOOD Leakage PGOOD = 15V 1 50 nA
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.
Note 4: Current measurements are made when the output is not switching.
Note 5: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
Note 6: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a much higher thermal resistance.
Note 7: Switch timing measurements are made in an open-loop test
configuration. Timing in the application may vary somewhat from these
values due to differences in the switch pin voltage during non-overlap
durations when switch pin voltage is influenced by the magnitude and
duration of the inductor current.
Note 8: Voltage transients on the switch pin(s) beyond the DC limits
specified in the Absolute Maximum Ratings are non-disruptive to normal
operation when using good layout practices as described elsewhere in the
data sheet and application notes and as seen on the product demo board.
Efficiency, VOUT = 2.5V Efficiency, VOUT = 3.3V
Typical perForMance characTerisTics
TA = 25°C, unless otherwise noted.
Power Loss, VOUT = 2.5V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3129 G01
1000
1 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
BURST
PWM
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3129 G03
1000
1 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
BURST
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
3129 G02
1000
1 100
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
BURST
PWM
LTC3129
5
3129fc
For more information www.linear.com/LTC3129
Typical perForMance characTerisTics
Efficiency, VOUT = 12V
Maximum Output Current
vs VIN and VOUT
Efficiency, VOUT = 5V
TA = 25°C, unless otherwise noted.
Power Loss, VOUT = 3.3V Power Loss, VOUT = 5V
Power Loss, VOUT = 12V Efficiency, VOUT = 15V
Power Loss, VOUT = 15V
VIN (V)
2
IOUT (mA)
250
200
150
100
50
0133 4
3129 G11
1510 1411 1285 96 7
VOUT = 2.5V
VOUT = 3.3V
VOUT = 4.1V
VOUT = 5V
VOUT = 6.9V
VOUT = 8.2V
VOUT = 12V
VOUT = 15V
No Load Input Current
vs VIN and VOUT (PWM = 0V)
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3129 G05
1000
1 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3129 G09
1000
1 100
PWM
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
3129 G04
1000
1 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
3129 G06
1000
1 100
PWM
BURST VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
40
80
60
30
20
10
0100.1
3129 G07
1000
1 100
BURST
PWM
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
3129 G08
1000
1 100
PWM
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
OUTPUT CURRENT (mA)
0.01
POWER LOSS (mW)
1000
100
10
1
0.1
0.01 100.1
3129 G10
1000
1 100
BURST
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
PWM
VIN (V)
2.5
IIN (µA)
30
25
20
15
10
5
012.5
3129 G12
14.510.54.5 8.56.5
VOUT = 2.5V
VOUT = 5V
VOUT = 10V
VOUT = 15V
FB DIVIDER CURRENT = 2µA
LTC3129
6
3129fc
For more information www.linear.com/LTC3129
Typical perForMance characTerisTics
Accurate RUN Threshold
vs Temperature (Normalized to 25°C)
Maximum Output Current
vs Temperature (Normalized to 25°C)
VCC Dropout Voltage vs Temperature
(PWM Mode, Switching)
VCC Dropout Voltage vs VIN
(PWM Mode, Switching)
Fixed Frequency PWM
Waveforms
TA = 25°C, unless otherwise noted.
Average Input Current Limit
vs MPPC Voltage
Burst Mode Threshold
vs VIN and VOUT Switch RDS(ON) vs Temperature
TEMPERATURE (°C)
–45
RDS(ON) (Ω)
1.3
1.2
1.1
1.0
0.8
0.7
0.6
0.5
0.9
0.4 –20
3129 G14
13055 10580305
VCC = 2.5V
VCC = 3V
VCC = 4V
VCC = 5V
FB Voltage vs Temperature
(Normalized to 25°C)
VIN (V)
2
LOAD (mA)
80
70
60
40
30
20
10
50
04
3129 G13
16
10 141286
VOUT = 2.5V
VOUT = 3.3V
VOUT = 4.1V
VOUT = 5V
VOUT = 6.9V
VOUT = 8.2V
VOUT = 12V
VOUT = 15V
TEMPERATURE (°C)
–45
CHANGE IN FB VOLTAGE (%)
1.00
0.50
0.75
0
–0.50
–0.75
–0.25
0.25
–1.00 –20
3129 G15
130
55 10580305
TEMPERATURE (°C)
–45
CHANGE IN RUN THRESHOLD (%)
2
0
–1
1
–2 –20
3129 G16
130
55 10580305
MPPC PIN VOLTAGE (V)
1.13
PERCENTAGE OF FULL INPUT CURRENT (%)
100
90
70
60
50
40
30
20
10
80
01.135
3129 G17
1.17
1.1651.161.1551.145 1.151.14
TEMPERATURE (°C)
–45
10
0
–5
5
–15
–10
–20
55 10580305
TEMPERATURE (°C)
–45
DROPOUT (mV)
60
50
30
20
40
0
10
–20
3129 G19
130
55 10580305
VIN (V)
2
DROPOUT (mV)
60
50
30
20
40
0
10
2.25
3129 G20
4
3 3.5 3.753.252.752.5
L = 10µH
VIN = 7V
VOUT = 5V
I
OUT
= 200mA
SW2
5V/DIV
SW1
5V/DIV
3129 G21
500ns/DIV
IL
200mA/DIV
LTC3129
7
3129fc
For more information www.linear.com/LTC3129
Fixed Frequency Ripple on VOUT
Typical perForMance characTerisTics
Step Load Transient Response in
Fixed Frequency
Step Load Transient Response in
Burst Mode Operation
PGOOD Response to a Drop
On VOUT
Burst Mode Waveforms Burst Mode Ripple on VOUT
Start-Up Waveforms
TA = 25°C, unless otherwise noted.
MPPC Response to a Step Load
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 200mA
C
OUT
= 10µF
3129 G22
200ns/DIV
IL
200mA/DIV
VOUT
20mV/DIV
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 5mA
COUT = 22µF (WITH THE RECOMMENDED
FEEDFORWARD CAPACITOR)
3129 G24
100µs/DIV
IL
100mA/DIV
VOUT
100mV/DIV
VIN = 7V
VOUT = 5V
IOUT = 50mA
C
OUT
= 22µF
3129 G25
1ms/DIV
IVIN
200mA/DIV
VOUT
5V/DIV
VCC
5V/DIV
RUN
5V/DIV
L = 10µH
VIN = 7V
VOUT = 5V
COUT = 10µF
I
OUT
= 50mA to 150mA STEP
3129 G26
500µs/DIV
IVOUT
100mA/DIV
VOUT
100mV/DIV
L = 10µH
VIN = 7V
VOUT = 5V
COUT = 22µF (WITH THE RECOMMENDED
FEEDFORWARD CAPACITOR)
I
OUT
= 5mA to 125mA STEP
3129 G27
500µs/DIV
IVOUT
100mA/DIV
VOUT
100mV/DIV
V
OUT
= 5V
3129 G28
1ms/DIV
PGOOD
2V/DIV
VOUT
2V/DIV
VIN = 5VOC
VMPPC SET TO 3.5V
CIN = 22µF, RIN = 10Ω,
VOUT = 5V, COUT = 22µF
I
OUT
= 25mA to 125mA STEP
3129 G29
2ms/DIV
IVOUT
100mA/DIV
VOUT
2V/DIV
VIN
2V/DIV
L = 10µH
VIN = 7V
VOUT = 5V
IOUT = 5mA
COUT = 22µF
3129 G23
50µs/DIV
IL
200mA/DIV
SW2
5V/DIV
SW1
5V/DIV
LTC3129
8
3129fc
For more information www.linear.com/LTC3129
pin FuncTions
BST1 (Pin 1/Pin 15): Bootstrapped Floating Supply for
High Side NMOS Gate Drive. Connect to SW1 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used.
VIN (Pin 2/Pin 16): Input Voltage for the Converter. Connect
a minimum of 4.7µF ceramic decoupling capacitor from
this pin to the ground plane, as close to the pin as possible.
VCC (Pin 3/Pin 1): Output voltage of the internal voltage
regulator. This is the supply pin for the internal circuitry.
Bypass this output with a minimum of 2.2µF ceramic
capacitor close to the pin. This pin may be back-driven by
an external supply, up to a maximum of 5.5V.
RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull
this pin above 1.1V to enable the VCC regulator and above
1.28V to enable the converter. Connecting this pin to a
resistor divider from VIN to ground allows programming a
VIN start threshold higher than the 1.8V (typical) VIN UVLO
threshold. In this case, the typical VIN turn-on threshold is
determined by VIN = 1.22V • [1+(R3/R4)] (see Figure 2).
MPPC (Pin 5/Pin 3): Maximum Power Point Control Pro-
gramming Pin. Connect this pin to a resistor divider from
VIN to ground to enable the MPPC functionality. If the VOUT
load is greater than what the power source can provide,
the MPPC will reduce the inductor current to regulate VIN
to a voltage determined by: VIN = 1.175V • [1+(R5/R6)]
(see Figure 3). By setting the VIN regulation voltage appro-
priately, maximum power transfer from the limited source
is assured. Note this pin is very noise sensitive, therefore
minimize trace length and stray capacitance. Please refer
to the Applications Information section for more detail
on programming the MPPC for different sources. If this
function is not needed, tie the pin to VCC.
GND (Pin 6/Pin 4): Signal Ground. Provide a short direct
PCB path between GND and the ground plane where the
exposed pad is soldered.
FB (Pin 7/Pin 5): Feedback Input to the Error Amplifier.
Connect to a resistor divider from VOUT to ground. The
output voltage can be adjusted from 1.4V to 15.75V by:
VOUT = 1.175V • [1+(R1/R2)]. Note this pin is very noise
sensitive, therefore minimize trace length and stray ca-
pacitance.
NC (Pins 8, 9/Pins 6, 7): Unused. These pins should be
grounded.
PWM (Pin 10/Pin 8): Mode Select Pin.
PWM = Low (ground): Enables automatic Burst Mode
operation.
PWM = High (tie to VCC): Fixed frequency PMW opera-
tion.
This pin should not be allowed to float. It has an internal
5M pull-down resistor.
PGOOD (Pin 11/Pin 9): Open drain output that pulls to
ground when FB drops too far below its regulated voltage.
Connect a pull-up resistor from this pin to a positive sup-
ply. This pin can sink up to the absolute maximum rating
of 15mA when low. Refer to the Operation section of the
data sheet for more detail.
VOUT (Pin 12/Pin 10): Output voltage of the converter.
Connect a minimum value of 4.7µF ceramic capacitor from
this pin to the ground plane, as close to the pin as possible.
BST2 (Pin 13/Pin 11): Bootstrapped floating supply for
high side NMOS gate drive. Connect to SW2 through a
22nF capacitor, as close to the part as possible. The value
is not critical. Any value from 4.7nF to 47nF may be used.
SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
PGND (Pin 15, Exposed Pad Pin 17/Pin 13, Exposed
Pad Pin 17): Power Ground. Provide a short direct PCB
path between PGND and the ground plane. The exposed
pad must also be soldered to the PCB ground plane. It
serves as a power ground connection, and as a means of
conducting heat away from the die.
SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.
(QFN/MSOP)
LTC3129
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block DiagraM
3129 BD
LDO
VREF
START
VREF
VCC
VCC
VCC_GD
START
START
4.1V
1.175V
VREF
+
–
SD
UVLO
+
–
–
+
+
–
–
+THERMAL
SHUTDOWN
+
–
–
+
PWM
600mV
–7.5%
OSC
GND
SLEEP
100mV
RESET
ENABLE
ILIM
IZERO
ISENSE
20mA
ISENSE
VREF_GD
500mA
PGND
CLAMP
+
–
1.175V
–
+
–
+
–
+
–
+
DRIVER
DRIVER
DRIVER
DRIVER
ISENSE
ISENSE
DRV_C
1.1V
UV
NC
VOUT
VCC
V
OUT
NC
FB
DRV_D
DRV_B
DRV_A
ISENSE
LOGIC
PGOOD
SOFT-START
+
–
MPPC
RUN
VCC
VIN
V
IN
BST1 SW1 SW2
D
C
A
B
BST2
PWM
SLEEP
VIN
0.9V
1.22V
1.175V
1.175V
VC
5M
LTC3129
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For more information www.linear.com/LTC3129
operaTion
INTRODUCTION
The LTC3129 is a 1.3µA quiescent current, monolithic, cur-
rent mode, buck-boost DC/DC converter that can operate
over a wide input voltage range of 1.92V to 15V and provide
up to 200mA to the load. Internal, low RDS(ON) N-channel
power switches reduce solution complexity and maximize
efficiency. A proprietary switch control algorithm allows the
buck-boost converter to maintain output voltage regulation
with input voltages that are above, below or equal to the
output voltage. Transitions between the step-up or step-
down operating modes are seamless and free of transients
and sub-harmonic switching, making this product ideal
for noise sensitive applications. The LTC3129 operates
at a fixed nominal switching frequency of 1.2MHz, which
provides an ideal trade-off between small solution size and
high efficiency. Current mode control provides inherent
input line voltage rejection, simplified compensation and
rapid response to load transients.
Burst Mode capability is also included in the LTC3129 and
is user-selected via the PWM input pin. In Burst Mode
operation, the LTC3129 provides exceptional efficiency at
light output loading conditions by operating the converter
only when necessary to maintain voltage regulation. The
Burst Mode quiescent current is a miserly 1.3µA. At higher
loads, the LTC3129 automatically switches to fixed fre-
quency PWM mode when Burst Mode operation is selected.
(Please refer to the Typical Performance Characteristics
curves for the mode transition point at different input and
output voltages.) If the application requires extremely low
noise, continuous PWM operation can also be selected
via the PWM pin.
A MPPC (maximum power point control) function is also
provided that allows the input voltage to the converter to
be servo'd to a programmable point for maximum power
when operating from various non-ideal power sources
such as photovoltaic cells. The LTC3129 also features
an accurate RUN comparator threshold with hysteresis,
allowing the buck-boost DC/DC converter to turn on and
off at user-selected VIN voltage thresholds. With a wide
voltage range, 1.3µA Burst Mode current and program-
mable RUN and MPPC pins, the LTC3129 is well suited
for many diverse applications.
PWM MODE OPERATION
If the PWM pin is high or if the load current on the converter
is high enough to command PWM mode operation with
PWM low, the LTC3129 operates in a fixed 1.2MHz PWM
mode using an internally compensated average current
mode control loop. PWM mode minimizes output voltage
ripple and yields a low noise switching frequency spec-
trum. A proprietary switching algorithm provides seamless
transitions between operating modes and eliminates dis-
continuities in the average inductor current, inductor ripple
current and loop transfer function throughout all modes of
operation. These advantages result in increased efficiency,
improved loop stability and lower output voltage ripple in
comparison to the traditional buck-boost converter.
Figure 1 shows the topology of the LTC3129 power stage
which is comprised of four N-channel DMOS switches and
their associated gate drivers. In PWM mode operation
both switch pins transition on every cycle independent of
the input and output voltages. In response to the internal
control loop command, an internal pulse width modulator
generates the appropriate switch duty cycle to maintain
regulation of the output voltage.
Figure 1. Power Stage Schematic
A
VCC
BST1
CBST1 CBST2
L
BST2VIN VOUT
SW1 SW2
VCC
VCC VCC
LTC3129
PGND PGND
3129 F01
B
D
C
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operaTion
When stepping down from a high input voltage to a lower
output voltage, the converter operates in buck mode and
switch D remains on for the entire switching cycle except
for the minimum switch low duration (typically 90ns). Dur-
ing the switch low duration, switch C is turned on which
forces SW2 low and charges the flying capacitor, CBST2.
This ensures that the switch D gate driver power supply
rail on BST2 is maintained. The duty cycle of switches A
and B are adjusted to maintain output voltage regulation
in buck mode.
If the input voltage is lower than the output voltage, the
converter operates in boost mode. Switch A remains on
for the entire switching cycle except for the minimum
switch low duration (typically 90ns). During the switch
low duration, switch B is turned on which forces SW1
low and charges the flying capacitor, CBST1. This ensures
that the switch A gate driver power supply rail on BST1 is
maintained. The duty cycle of switches C and D are adjusted
to maintain output voltage regulation in boost mode.
Oscillator
The LTC3129 operates from an internal oscillator with a
nominal fixed frequency of 1.2MHz. This allows the DC/DC
converter efficiency to be maximized while still using small
external components.
Current Mode Control
The LTC3129 utilizes average current mode control for the
pulse width modulator. Current mode control, both average
and the better known peak method, enjoy some benefits
compared to other control methods including: simplified
loop compensation, rapid response to load transients and
inherent line voltage rejection.
Referring to the Block Diagram, a high gain, internally
compensated transconductance amplifier monitors Vout
through a voltage divider connected to the FB pin. The
error amplifier output is used by the current mode control
loop to command the appropriate inductor current level.
The inverting input of the internally compensated average
current amplifier is connected to the inductor current
sense circuit. The average current amplifier's output is
compared to the oscillator ramps, and the comparator
outputs are used to control the duty cycle of the switch
pins on a cycle-by-cycle basis.
The voltage error amplifier monitors the output voltage,
VOUT through a voltage divider and makes adjustments to
the current command as necessary to maintain regulation.
The voltage error amplifier therefore controls the outer
voltage regulation loop. The average current amplifier
makes adjustments to the inductor current as directed by
the voltage error amplifier output via VC and is commonly
referred to as the inner current loop amplifier.
The average current mode control technique is similar to
peak current mode control except that the average current
amplifier, by virtue of its configuration as an integrator,
controls average current instead of the peak current. This
difference eliminates the peak to average current error
inherent to peak current mode control, while maintaining
most of the advantages inherent to peak current mode
control.
Average current mode control requires appropriate com-
pensation for the inner current loop, unlike peak current
mode control. The compensation network must have high
DC gain to minimize errors between the actual and com-
manded average current level, high bandwidth to quickly
change the commanded current level following transient
load steps and a controlled mid-band gain to provide a
form of slope compensation unique to average current
mode control. The compensation components required
to ensure proper operation have been carefully selected
and are integrated within the LTC3129.
Inductor Current Sense and Maximum Output Current
As part of the current control loop required for current
mode control, the LTC3129 includes a pair of current
sensing circuits that measure the buck-boost converter
inductor current.
The voltage error amplifier output, VC, is internally clamped
to a nominal level of 0.6V. Since the average inductor
current is proportional to VC, the 0.6V clamp level sets
the maximum average inductor current that can be pro-
grammed by the inner current loop. Taking into account
the current sense amplifier's gain, the maximum average
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inductor current is approximately 275mA (typical). In
Buck mode, the output current is approximately equal to
the inductor current IL.
IOUT(BUCK) ≈ IL • 0.89
The 90ns SW1/SW2 forced low time on each switching
cycle briefly disconnects the inductor from VOUT and VIN
resulting in about 11% less output current in either buck
or boost mode for a given inductor current. In boost mode,
the output current is related to average inductor current
and duty cycle by:
IOUT(BOOST) ≈ IL • (1 – D) • Efficiency,
where D is the converter duty cycle.
Since the output current in boost mode is reduced by the
duty cycle (D), the output current rating in buck mode is
always greater than in boost mode. Also, because boost
mode operation requires a higher inductor current for a
given output current compared to buck mode, the efficiency
in boost mode will be lower due to higher IL² • RDS(ON)
losses in the power switches. This will further reduce the
output current capability in boost mode. In either operating
mode, however, the inductor peak-to-peak ripple current
does not play a major role in determining the output cur-
rent capability, unlike peak current mode control.
With peak current mode control, the maximum output
current capability is reduced by the magnitude of inductor
ripple current because the peak inductor current level is the
control variable, but the average inductor current is what
determines the output current. The LTC3129 measures
and controls average inductor current, and therefore, the
inductor ripple current magnitude has little effect on the
maximum current capability in contrast to an equivalent
peak current mode converter. Under most conditions in
buck mode, the LTC3129 is capable of providing a mini-
mum of 200mA to the load. In boost mode, as described
previously, the output current capability is related to the
boost ratio or duty cycle (D). For example, for a 3.6V VIN
to 5V output application, the LTC3129 can provide up
to 150mA to the load. Refer to the Typical Performance
characteristics section for more detail on output current
capability.
operaTion
Overload Current Limit and IZERO Comparator
The internal current sense waveform is also used by the
peak overload current (IPEAK) and zero current (IZERO) com-
parators. The IPEAK current comparator monitors Isense
and turns off switch A if the inductor current level exceeds
its maximum internal threshold, which is approximately
500mA. An inductor current level of this magnitude will
occur during a fault, such as an output short-circuit, or
during large load or input voltage transients.
The LTC3129 features near discontinuous inductor current
operation at light output loads by virtue of the IZERO com-
parator circuit. By limiting the reverse current magnitude
in PWM mode, a balance between low noise operation and
improved efficiency at light loads is achieved. The IZERO
comparator threshold is set near the zero current level in
PWM mode, and as a result, the reverse current magnitude
will be a function of inductance value and output voltage
due to the comparator's propagation delay. In general,
higher output voltages and lower inductor values will
result in increased reverse current magnitude.
In automatic Burst Mode operation (PWM pin low), the
IZERO comparator threshold is increased so that reverse
inductor current does not normally occur. This maximizes
efficiency at very light loads.
Burst Mode OPERATION
When the PWM pin is held low, the LTC3129 is config-
ured for automatic Burst Mode operation. As a result,
the buck-boost DC/DC converter will operate with normal
continuous PWM switching above a predetermined mini-
mum output load and will automatically transition to power
saving Burst Mode operation below this output load level.
Note that if the PWM pin is low, reverse inductor current is
not allowed at any load. Refer to the Typical Performance
Characteristics section to determine the Burst Mode
transition threshold for various combinations of VIN and
VOUT. If PWM is low, at light output loads, the LTC3129
will go into a standby or sleep state when the output volt-
age achieves its nominal regulation level. The sleep state
halts PWM switching and powers down all non-essential
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operaTion
functions of the IC, significantly reducing the quiescent
current of the LTC3129 to just 1.3µA typical. This greatly
improves overall power conversion efficiency when the
output load is light. Since the converter is not operating
in sleep, the output voltage will slowly decay at a rate
determined by the output load resistance and the output
capacitor value. When the output voltage has decayed by
a small amount, typically 1%, the LTC3129 will wake and
resume normal PWM switching operation until the volt-
age on VOUT is restored to the previous level. If the load
is very light, the LTC3129 may only need to switch for a
few cycles to restore VOUT and may sleep for extended
periods of time, significantly improving efficiency. If the
load is suddenly increased above the burst transition
threshold, the part will automatically resume continuous
PWM operation until the load is once again reduced.
A feedforward capacitor on the feedback divider can be
used to reduce Burst Mode VOUT ripple. This is discussed
in more detail in the Applications Information section of
this data sheet.
Note that Burst Mode operation is inhibited until soft-start
is done, the MPPC pin is greater than 1.175V and VOUT
has reached regulation.
Soft-Start
The LTC3129 soft-start circuit minimizes input current
transients and output voltage overshoot on initial power up.
The required timing components for soft-start are internal
to the LTC3129 and produce a nominal soft-start dura-
tion of approximately 3ms. The internal soft-start circuit
slowly ramps the error amplifier output, VC. In doing so,
the current command of the IC is also slowly increased,
starting from zero. It is unaffected by output loading or
output capacitor value. Soft-start is reset by the UVLO on
both VIN and VCC, the RUN pin and thermal shutdown.
VCC Regulator
An internal low dropout regulator (LDO) generates a nomi-
nal 4.1V VCC rail from VIN. The VCC rail powers the internal
control circuitry and the gate drivers of the LTC3129. The
VCC regulator is disabled in shutdown to reduce quiescent
current and is enabled by raising the RUN pin above its
logic threshold. The VCC regulator includes current-limit
protection to safeguard against accidental short-circuiting
of the VCC rail.
Undervoltage Lockout (UVLO)
There are two undervoltage lockout (UVLO) circuits within
the LTC3129 that inhibit switching; one that monitors VIN
and another that monitors VCC. Either UVLO will disable
operation of the internal power switches and keep other
IC functions in a reset state if either VIN or VCC are below
their respective UVLO thresholds.
The VIN UVLO comparator has a falling voltage threshold
of 1.8V (typical). If VIN falls below this level, IC operation
is disabled until VIN rises above 1.9V (typical), as long as
the VCC voltage is above its UVLO threshold.
The VCC UVLO has a falling voltage threshold of 2.19V
(typical). If the VCC voltage falls below this threshold, IC
operation is disabled until VCC rises above 2.25V (typical)
as long as VIN is above its nominal UVLO threshold level.
Depending on the particular application, either of these
UVLO thresholds could be the limiting factor affecting the
minimum input voltage required for operation. Because the
VCC regulator uses VIN for its power input, the minimum
input voltage required for operation is determined by the
VCC minimum voltage, as input voltage (VIN) will always
be higher than VCC in the normal (non-bootstrapped)
configuration. Therefore, the minimum VIN for the part
to startup is 2.25V (typical).
In applications where VCC is bootstrapped (powered
through a Schottky diode by either VOUT or an auxiliary
power rail), the minimum input voltage for operation will
be limited only by the VIN UVLO threshold (1.8V typical).
Please note that if the bootstrap voltage is derived from
the LTC3129 VOUT and not an independent power rail, then
the minimum input voltage required for initial startup is
still 2.25V (typical).
Note that if either VIN or VCC are below their UVLO thresh-
olds, or if RUN is below its accurate threshold of 1.22V
(typical), then the LTC3129 will remain in a soft shutdown
state, where the VIN quiescent current will be only 1.9µA
typical.
LTC3129
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VOUT Undervoltage
There is also an undervoltage comparator that monitors
the output voltage. Until VOUT reaches 1.15V (typical), the
average current limit is reduced by a factor of two. This
reduces power dissipation in the device in the event of a
shorted output. In addition, N-channel switch D, which
feeds VOUT, will be disabled until VOUT exceeds 1.15V.
RUN Pin Comparator
In addition to serving as a logic level input to enable cer-
tain functions of the IC, the RUN pin includes an accurate
internal comparator that allows it to be used to set custom
rising and falling ON/OFF thresholds with the addition of
an optional external resistor divider. When RUN is driven
above its logic threshold (0.9V typical), the VCC regulator
is enabled, which provides power to the internal control
circuitry of the IC. If the voltage on RUN is increased further
so that it exceeds the RUN comparator's accurate analog
threshold (1.22V typical), all functions of the buck-boost
converter will be enabled and a start-up sequence will ensue,
assuming the VIN and VCC UVLO thresholds are satisfied.
If RUN is brought below the accurate comparator threshold,
the buck-boost
converter will inhibit switching, but the VCC
regulator and control circuitry will remain powered unless
RUN is brought below its logic threshold. Therefore, in
order to completely shut down the IC and reduce the Vin
current to 10nA (typical), it is necessary to ensure that
RUN is brought below its worst case low logic threshold of
0.5V. RUN is a high voltage input and can be tied directly
to VIN to continuously enable the IC when the input supply
is present. Also note that RUN can be driven above VIN
or VOUT as long as it stays within the operating range of
the IC (up to 15V).
With the addition of an optional resistor divider as shown
in Figure 2, the RUN pin can be used to establish a user-
programmable turn-on and turn-off threshold. This feature
can be utilized to minimize battery drain below a certain
input voltage, or to operate the converter in a hiccup mode
from very low current sources.
Figure 2. Accurate RUN Pin Comparator
LTC3129
ENABLE SWITCHING
ENABLE LDO AND
CONTROL CIRCUITS
LOGIC THRESHOLD
ACCURATE THRESHOLD
3129 F02
+
–
–
+
0.9V
RUN
1.22V
VIN
R3
R4
Note that once RUN is above 0.9V typical, the quiescent
input current on VIN (or VCC if back-driven) will increase to
about 1.9µA typical until the VIN and VCC UVLO thresholds
are satisfied.
The converter is enabled when the voltage on RUN exceeds
1.22V (nominal). Therefore, the turn-on voltage threshold
on VIN is given by:
VIN(TURN-ON) = 1.22V • (1 + R3/R4)
The RUN comparator includes a built-in hysteresis of
approximately 80mV, so that the turn off threshold will
be 1.14V.
There may be cases due to PCB layout, very large value
resistors for R3 and R4, or proximity to noisy components
where noise pickup may cause the turn-on or turn-off of the
IC to be intermittent. In these cases, a small filter capacitor
can be added across R4 to ensure proper operation.
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operaTion
PGOOD Comparator
The LTC3129 provides an open-drain PGOOD output that
pulls low if VOUT falls more than 7.5% (typical) below its
programmed value. When VOUT rises to within 5% (typical)
of its programmed value, the internal PGOOD pull-down
will turn off and PGOOD will go high if an external pull-up
resistor has been provided. An internal filter prevents
nuisance trips of PGOOD due to short transients on VOUT.
Note that PGOOD can be pulled up to any voltage, as long
as the absolute maximum rating of 18V is not exceeded,
and as long as the maximum sink current rating is not
exceeded when PGOOD is low. Note that PGOOD will
also be driven low if VCC is below its UVLO threshold or
if the part is in shutdown (RUN below its logic threshold)
while VCC is being held up (or back-driven). PGOOD is
not affected by VIN UVLO or the accurate RUN threshold.
In cases where VCC is not being back-driven in shutdown,
PGOOD will not be held low indefinitely. The internal PGOOD
pull-down will be disabled as the VCC voltage decays below
approximately 1V.
Maximum Power-Point Control (MPPC)
The MPPC input of the LTC3129 can be used with an
optional external voltage divider to dynamically adjust
the commanded inductor current in order to maintain
a minimum input voltage when using high resistance
sources, such as photovoltaic panels, so as to maximize
input power transfer and prevent VIN from dropping too
low under load. Referring to Figure 3, the MPPC pin is
internally connected to the non-inverting input of a gm
amplifier, whose inverting input is connected to the 1.175V
reference. If the voltage at MPPC, using the external volt-
age divider, falls below the reference voltage, the output of
the amplifier pulls the internal VC node low. This reduces
the commanded average inductor current so as to reduce
the input current and regulate VIN to the programmed
minimum voltage, as given by:
VIN(MPPC) = 1.175V • (1 + R5/R6)
The MPPC feature provides capabilities to the LTC3129 that
can ease the design of intrinsically safe power supplies.
Note that external compensation should not be required
for MPPC loop stability if input filter capacitor, CIN, is at
least 22µF.
The divider resistor values can be in the MΩ range to
minimize the input current in very low power applications.
However, stray capacitance and noise pickup on the MPPC
pin must also be minimized.
Figure 3. MPPC Amplifier with External Resistor Divider
LTC3129
1.175V
VC
CURRENT
COMMAND
VOLTAGE
ERROR AMP
3129 F03
MPPC
FB
R5
R6
RS
VSOURCE
*CIN
* CIN SHOULD BE AT
LEAST 22µF FOR
MPPC APPLICATIONS
VIN
+
–
+
–
+
–
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operaTion
Figure 4. Typical 2-Layer PC Board Layout (MSE Package)
The MPPC pin controls the converter in a linear fashion
when using sources that can provide a minimum of 5mA
to 10mA of continuous input current. For operation from
weaker input sources, refer to the Applications Informa-
tion section to see how the programmable RUN pin can
be used to control the converter in a hysteretic manner to
provide an effective MPPC function for sources that can
provide as little as 5µA or less. If the MPPC function is not
required, the MPPC pin should be tied to VCC.
Thermal Considerations
The power switches of the LTC3129 are designed to oper-
ate continuously with currents up to the internal current
limit thresholds. However, when operating at high current
levels, there may be significant heat generated within the
IC. In addition, the VCC regulator can also generate wasted
heat when VIN is very high, adding to the total power
dissipation of the IC. As described elsewhere in this data
sheet, bootstrapping of the VCC for 5V output applications
can essentially eliminate the VCC power dissipation term
and significantly improve efficiency. As a result, careful
consideration must be given to the thermal environment
of the IC in order to provide a means to remove heat from
the IC and ensure that the LTC3129 is able to provide its
full rated output current. Specifically, the exposed die
attach pad of both the QFN and MSE packages must be
soldered to a copper layer on the PCB to maximize the
conduction of heat out of the IC package. This can be ac-
complished by utilizing multiple vias from the die attach
pad connection underneath the IC package to other PCB
layer(s) containing a large copper plane. A typical board
layout incorporating these concepts is shown in Figure 4.
If the IC die temperature exceeds approximately 180°C, over
temperature shutdown will be invoked and all switching
will be inhibited. The part will remain disabled until the die
temperature cools by approximately 10°C. The soft-start
circuit is re-initialized in overtemperature shutdown to
provide a smooth recovery when the IC die temperature
cools enough to resume operation.
GND VIN
VCC
GND VOUT
COUT
CIN
L
3129 F04
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applicaTions inForMaTion
A standard application circuit for the LTC3129 is shown on
the front page of this data sheet. The appropriate selection
of external components is dependent upon the required
performance of the IC in each particular application given
considerations and trade-offs such as PCB area, input
and output voltage range, output voltage ripple, transient
response, required efficiency, thermal considerations and
cost. This section of the data sheet provides some basic
guidelines and considerations to aid in the selection of
external components and the design of the applications
circuit, as well as more application circuit examples.
Programming VOUT
The output voltage of the LTC3129 is set by connecting
the FB pin to an external resistor divider from VOUT to
ground, as shown in Figure 5, according to the equation:
VOUT = 1.175V • (1+ R1/R2)
sible. VCC is the regulator output and is also the internal
supply pin for the LTC3129 control circuitry as well as the
gate drivers and boost rail charging diodes. The VCC pin is
not intended to supply current to other external circuitry.
Inductor Selection
The choice of inductor used in LTC3129 application circuits
influences the maximum deliverable output current, the
converter bandwidth, the magnitude of the inductor current
ripple and the overall converter efficiency. The inductor
must have a low DC series resistance, when compared to
the internal switch resistance, or output current capabil-
ity and efficiency will be compromised. Larger inductor
values reduce inductor current ripple but may not increase
output current capability as is the case with peak current
mode control as described in the Maximum Output Cur-
rent section. Larger value inductors also tend to have a
higher DC series resistance for a given case size, which
will have a negative impact on efficiency. Larger values
of inductance will also lower the right half plane (RHP)
zero frequency when operating in boost mode, which can
compromise loop stability. Nearly all LTC3129 application
circuits deliver the best performance with an inductor value
between 3.3µH and 10µH. Buck mode-only applications
can use the larger inductor values as they are unaffected
by the RHP zero, while mostly boost applications generally
require inductance on the low end of this range depending
on how large the step-up ratio is.
Regardless of inductor value, the saturation current rating
should be selected such that it is greater than the worst-case
average inductor current
plus half of the ripple current. The
peak-to-peak inductor current ripple for each operational
mode can be calculated from the following formula, where
f is the switching frequency (1.2MHz), L is the inductance
in µH and tLOW is the switch pin minimum low time in
µs. The switch pin minimum low time is typically 0.09µs.
ΔIL(P−P)(BUCK) =VOUT
L
V
IN – VOUT
V
IN
⎛
⎝
⎜⎞
⎠
⎟1
f– tLOW
⎛
⎝
⎜⎞
⎠
⎟A
ΔIL(P−P)(BOOST) =V
IN
L
VOUT – V
IN
VOUT
⎛
⎝
⎜⎞
⎠
⎟1
f– tLOW
⎛
⎝
⎜⎞
⎠
⎟A
Figure 5. VOUT Feedback Divider
A small feedforward capacitor can be added in parallel with
R1 (in Figure 5) to reduce Burst Mode ripple and improve
transient response. Details on selecting a feedforward
capacitor are provided later in this data sheet.
VCC Capacitor Selection
The VCC output of the LTC3129 is generated from VIN by a
low dropout linear regulator. The VCC regulator has been
designed for stable operation with a wide range of output
capacitors. For most applications, a low ESR capacitor of
at least 2.2µF should be used. The capacitor should be
located as close to the VCC pin as possible and connected
to the VCC pin and ground through the shortest traces pos-
3129 F05
LTC3129
VOUT VOUT
R1
COUT
FB
CFF
R2
LTC3129
18
3129fc
For more information www.linear.com/LTC3129
It should be noted that the worst-case peak-to-peak in-
ductor ripple current occurs when the duty cycle in buck
mode is minimum (highest VIN) and in boost mode when
the duty cycle is 50% (VOUT = 2 • VIN). As an example, if
VIN (minimum) = 2.5V and VIN (maximum) = 15V, VOUT
= 5V and L = 10µH, the peak-to-peak inductor ripples at
the voltage extremes (15V VIN for buck and 2.5V VIN for
boost) are:
Buck = 248mA peak-to-peak
Boost = 93mA peak-to-peak
One half of this inductor ripple current must be added to
the highest expected average inductor current in order to
select the proper saturation current rating for the inductor.
To avoid the possibility of inductor saturation during load
transients, an inductor with a saturation current rating of
at least 600mA is recommended for all applications.
In addition to its influence on power conversion efficiency,
the inductor DC resistance can also impact the maximum
output current capability of the buck-boost converter
particularly at low input voltages. In buck mode, the
output current of the buck-boost converter is primarily
limited by the inductor current reaching the average cur-
rent limit threshold. However, in boost mode, especially
at large step-up ratios, the output current capability can
also be limited by the total resistive losses in the power
stage. These losses include, switch resistances, inductor
DC resistance and PCB trace resistance. Avoid inductors
with a high DC resistance (DCR) as they can degrade the
maximum output current capability from what is shown
in the Typical Performance Characteristics section and
from the Typical Application circuits.
As a guideline, the inductor DCR should be significantly
less than the typical power switch resistance of 750mΩ
each. The only exceptions are applications that have a
maximum output current requirement much less than what
the LTC3129 is capable of delivering. Generally speaking,
inductors with a DCR in the range of 0.15Ω to 0.3Ω are
recommended. Lower values of DCR will improve the ef-
ficiency at the expense of size, while higher DCR values
will reduce efficiency (typically by a few percent) while
allowing the use of a physically smaller inductor.
applicaTions inForMaTion
Different inductor core materials and styles have an impact
on the size and price of an inductor at any given current
rating. Shielded construction is generally preferred as it
minimizes the chances of interference with other circuitry.
The choice of inductor style depends upon the price, sizing,
and EMI requirements of a particular application. Table 1
provides a wide sampling of inductors that are well suited
to many LTC3129 applications.
Table 1. Recommended Inductors
VENDOR PART
Coilcraft
www.coilcraft.com
EPL2014, EPL3012, EPL3015, LPS3015,
LPS3314, XFL3012
Coiltronics
www.cooperindustries.com
SDH3812, SD3814, SD3114, SD3118
Murata
www.murata.com
LQH3NP, LQH32P, LQH44P
Sumida
www.sumida.com
CDRH2D16, CDRH2D18, CDRH3D14,
CDRH3D16
Taiyo-Yuden
www.t-yuden.com
NR3012T, NR3015T, NRS4012T,
BRC2518
TDK
www.tdk.com
VLS3012, VLS3015, VLF302510MT,
VLF302512MT
Toko
www.tokoam.com
DB3015C, DB3018C, DB3020C, DP418C,
DP420C, DEM2815C, DFE322512C,
DFE252012C
Würth
www.we-online.com
WE-TPC 2813, WE-TPC 3816,
WE-TPC 2828
Recommended inductor values for different operating
voltage ranges are given in Table 2. These values were
chosen to minimize inductor size while maintaining an
acceptable amount of inductor ripple current for a given
VIN and VOUT range.
Table 2. Recommended Inductor Values
VIN AND VOUT RANGE RECOMMENDED INDUCTOR VALUES
VIN and VOUT Both < 4.5V 3.3µH to 4.7µH
VIN and VOUT Both < 8V 4.7µH to 6.8µH
VIN and VOUT Both < 11V 6.8µH to 8.2µH
VIN and VOUT Up to 15.75V 8.2µH to 10µH
Feedforward Capacitor
The use of a voltage feedforward capacitor, as shown in
Figure 5, offers a number of performance advantages. A
feedforward capacitor will reduce output voltage ripple in
LTC3129
19
3129fc
For more information www.linear.com/LTC3129
applicaTions inForMaTion
Examining the previous equations reveals that the output
voltage ripple increases with load current and is gener-
ally higher in boost mode than in buck mode. Note that
these equations only take into account the voltage ripple
that occurs from the inductor current to the output being
discontinuous. They provide a good approximation to the
ripple at any significant load current but underestimate the
output voltage ripple at very light loads where the output
voltage ripple is dominated by the inductor current ripple.
In addition to the output voltage ripple generated across
the output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportional to the series resistance of the output capacitor
and is given by the following expressions where RESR is
the series resistance of the output capacitor and all other
terms as previously defined.
ΔV
P−P(BUCK) =
I
LOAD
R
ESR
1– tLOWf≅ILOADRESR V
ΔV
P−P(BOOST) =ILOADRESRVOUT
VIN 1– tLOWf
( )
≅ILOADRESR
VOUT
V
IN
⎛
⎝
⎜⎞
⎠
⎟V
In most LTC3129 applications, an output capacitor between
10µF and 22µF will work well. To minimize output ripple
in Burst Mode operation, or transients incurred by large
step loads, values of 22µF or larger are recommended.
Input Capacitor Selection
The VIN pin carries the full inductor current and provides
power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 4.7µF
should be located as close to the VIN pin as possible. The
traces connecting this capacitor to VIN and the ground
plane should be made as short as possible.
Burst Mode operation and improve transient response. In
addition, due to the wide VIN and VOUT operating range
of the LTC3129 and its fixed internal loop compensation,
some applications may require the use of a feedforward
capacitor to assure light-load stability (less than ~15mA)
when operating in PWM mode (PWM pin pulled high).
Therefore, to reduce Burst Mode ripple and improve
phase margin at light load when PWM mode operation is
selected, a feedforward capacitor is recommended for all
applications. The recommended feedforward capacitor
value can be calculated by:
CFF = 66/R1
Where R1 is the top feedback divider resistor value in MΩ
and CFF is the recommended feedforward capacitor value
in picofarads (use the nearest standard value). Refer to
the application circuits for examples.
Output Capacitor Selection
A low effective series resistance (ESR) output capacitor
of 4.7µF minimum should be connected at the output of
the buck-boost converter in order to minimize output volt-
age ripple. Multilayer ceramic capacitors are an excellent
option as they have low ESR and are available in small
footprints. The capacitor value should be chosen large
enough to reduce the output voltage ripple to acceptable
levels. Neglecting the capacitor's ESR and ESL (effec-
tive series inductance), the peak-to-peak output voltage
ripple in PWM mode can be calculated by the following
formula, where f is the frequency in MHz (1.2MHz), COUT
is the capacitance in µF, tLOW is the switch pin minimum
low time in µs (0.09µs typical) and ILOAD is the output
current in amperes.
ΔV
P−P(BUCK) =
I
LOAD
t
LOW
COUT
V
ΔV
P−P(BOOST) =ILOAD
fCOUT
VOUT – VIN +tLOWfVIN
VOUT
⎛
⎝
⎜⎞
⎠
⎟V
LTC3129
20
3129fc
For more information www.linear.com/LTC3129
When powered through long leads or from a power
source with significant resistance, a larger value bulk input
capacitor may be required and is generally recommended.
In such applications, a 47µF to 100µF low-ESR electrolytic
capacitor in parallel with a 1µF ceramic capacitor generally
yields a high performance, low cost solution.
Note that applications using the MPPC feature should
use a minimum CIN of 22µF. Larger values can be used
without limitation.
Recommended Input and Output Capacitor Types
The capacitors used to filter the input and output of the
LTC3129 must have low ESR and must be rated to handle
the AC currents generated by the switching converter.
This is important to maintain proper functioning of the
IC and to reduce output voltage ripple. There are many
capacitor types that are well suited to these applications
including multilayer ceramic, low ESR tantalum, OS-CON
and POSCAP technologies. In addition, there are certain
types of electrolytic capacitors such as solid aluminum
organic polymer capacitors that are designed for low
ESR and high AC currents and these are also well suited
to some LTC3129 applications. The choice of capacitor
technology is primarily dictated by a trade-off between
size, leakage current and cost. In backup power applica-
tions, the input or output capacitor might be a super or
ultra capacitor with a capacitance value measuring in the
Farad range. The selection criteria in these applications
are generally similar except that voltage ripple is generally
not a concern. Some capacitors exhibit a high DC leak-
age current which may preclude their consideration for
applications that require a very low quiescent current in
Burst Mode operation. Note that ultra capacitors may have
a rather high ESR, therefore a 4.7µF (minimum) ceramic
capacitor is recommended in parallel, close to the IC pins.
Ceramic capacitors are often utilized in switching con-
verter applications due to their small size, low ESR and
low leakage currents. However, many ceramic capacitors
intended for power applications experience a significant
loss in capacitance from their rated value as the DC bias
voltage on the capacitor increases. It is not uncommon for
a small surface mount capacitor to lose more than 50%
of its rated capacitance when operated at even half of its
maximum rated voltage. This effect is generally reduced
as the case size is increased for the same nominal value
capacitor. As a result, it is often necessary to use a larger
value capacitance or a higher voltage rated capacitor than
would ordinarily be required to actually realize the intended
capacitance at the operating voltage of the application. X5R
and X7R dielectric types are recommended as they exhibit
the best performance over the wide operating range and
temperature of the LTC3129. To verify that the intended
capacitance is achieved in the application circuit, be sure
to consult the capacitor vendor's curve of capacitance
versus DC bias voltage.
Using the Programmable RUN Function to Operate
from Extremely Weak Input Sources
Another application of the programmable RUN pin is that
it can be used to operate the converter in a hiccup mode
from extremely low current sources. This allows operation
from sources that can only generate microamps of output
current, and would be far too weak to sustain normal steady-
state operation, even with the use of the MPPC pin. Because
the LTC3129 draws only 1.9µA typical from VIN until it is
enabled, the RUN pin can be programmed to keep the IC
disabled until VIN reaches the programmed voltage level.
In this manner, the input source can trickle-charge an input
storage capacitor, even if it can only supply microamps of
current, until VIN reaches the turn-on threshold set by the
RUN pin divider. The converter will then be enabled using
the stored charge in the input capacitor, until Vin drops
below the turn-off threshold, at which point the converter
will turn off and the process will repeat.
This approach allows the converter to run from weak
sources such as thin-film solar cells using indoor lighting.
Although the converter will be operating in bursts, it is
enough to charge an output capacitor to power low duty
cycle loads, such as wireless sensor applications, or to
trickle charge a battery. In addition, note that the input
voltage will be cycling (with a small ripple as set by the
RUN hysteresis) about a fixed voltage, as determined by
the divider. This allows the high impedance source to
operate at the programmed optimal voltage for maximum
power transfer.
applicaTions inForMaTion
LTC3129
21
3129fc
For more information www.linear.com/LTC3129
applicaTions inForMaTion
When using high value divider resistors (in the MΩ
range) to minimize current draw on VIN, a small noise
filter capacitor may be necessary across the lower divider
resistor to prevent noise from erroneously tripping the
RUN comparator. The capacitor value should be minimized
so as not to introduce a time delay long enough for the
input voltage to drop significantly below the desired VIN
threshold before the converter is turned off. Note that
larger VIN decoupling capacitor values will minimize this
effect by providing more holdup time on VIN.
Programming the MPPC Voltage
As discussed in the previous section, the LTC3129 in-
cludes an MPPC function to optimize performance when
operating from voltage sources with relatively high source
resistance. Using an external voltage divider from VIN, the
MPPC function takes control of the average inductor current
when necessary to maintain a minimum input voltage, as
programmed by the user. Referring to Figure 3:
VIN(MPPC) = 1.175V • (1 + R5/R6)
This is useful for such applications as photovoltaic pow-
ered converters, since the maximum power transfer point
occurs when the photovoltaic panel is operated at about
75% of its open-circuit voltage. For example, when operat-
ing from a photovoltaic panel with an open-circuit voltage
of 5V, the maximum power transfer point will be when
the panel is loaded such that its output voltage is about
3.75V. Choosing values of 2MΩ for R5 and 909kΩ for R6
will program the MPPC function to regulate the maximum
input current so as to maintain VIN at a minimum of 3.74V
(typical). Note that if the panel can provide more power
than the LTC3129 can draw, the input voltage will rise
above the programmed MPPC point. This is fine as long
as the input voltage doesn't exceed 15V.
For weak input sources with very high resistance (hun-
dreds of Ohms or more), the LTC3129 may still draw
more current than the source can provide, causing VIN to
drop below the UVLO threshold. For these applications, it
is recommended that the programmable RUN feature be
used, as described in the previous section.
MPPC Compensation and Gain
When using MPPC, there are a number of variables that
affect the gain and phase of the input voltage control
loop. Primarily these are the input capacitance, the MPPC
divider ratio and the VIN source resistance (or current). To
simplify the design of the application circuit, the MPPC
control loop in the LTC3129 is designed with a relatively
low gain, such that external MPPC loop compensation is
generally not required when using a VIN capacitor value
of at least 22µF. The gain from the MPPC pin to the in-
ternal VC control voltage is about 12, so a drop of 50mV
on the MPPC pin (below the 1.175V MPPC threshold),
corresponds to a 600mV drop on the internal VC voltage,
which reduces the average inductor current all the way
to zero. Therefore, the programmed input MPPC voltage
will be maintained within about 4% over the load range.
Note that if large value VIN capacitors are used (which may
have a relatively high ESR) a small ceramic capacitor of
at least 4.7µF should be placed in parallel across the VIN
input, near the VIN pin of the IC.
Bootstrapping the VCC Regulator
The high and low side gate drivers are powered through
the VCC rail, which is generated from the input voltage, VIN,
through an internal linear regulator. In some applications,
especially at high input voltages, the power dissipation
in the linear regulator can become a major contributor to
thermal heating of the IC and overall efficiency. The Typical
Performance Characteristics section provides data on the
VCC current and resulting power loss versus VIN and VOUT.
A significant performance advantage can be attained in high
VIN applications where converter output voltage (VOUT) is
programmed to 5V, if VOUT is used to power the VCC rail.
Powering VCC in this manner is referred to as bootstrapping.
This can be done by connecting a Schottky diode (such
as a BAT54) from VOUT to VCC as shown in Figure 6.
With the bootstrap diode installed, the gate driver currents
are supplied by the buck-boost converter at high efficiency
rather than through the internal linear regulator. The in-
ternal linear regulator contains reverse blocking circuitry
LTC3129
22
3129fc
For more information www.linear.com/LTC3129
that allows VCC to be driven above its nominal regulation
level with only a very slight amount of reverse current.
Please note that the bootstrapping supply (either VOUT or
a separate regulator) must be limited to less than 5.7V so
as not to exceed the maximum VCC voltage of 5.5V after
the diode drop.
By maintaining VCC above its UVLO threshold,
bootstrapping, even to a 3.3V output, also allows opera-
tion down to the VIN UVLO threshold of 1.8V (typical).
Sources of Small Photovoltaic Panels
A list of companies that manufacture small solar panels
(sometimes referred to as modules or solar cell arrays)
suitable for use with the LTC3129 is provided in Table 3.
Table 3. Small Photovoltaic Panel Manufacturers
Sanyo http://panasonic.net/energy/amorton/en/
PowerFilm http://www.powerfilmsolar.com/
Ixys
Corporation
http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx
G24
Innovations
http://www.g24i.com/
applicaTions inForMaTion
Figure 6. Example of VCC Bootstrap
3129 F06
LTC3129
VOUT VOUT
BAT54
COUT
VCC
2.2µF
LTC3129
23
3129fc
For more information www.linear.com/LTC3129
Typical applicaTions
Low Noise 3.6V Converter Using Bootstrap Diode to Extend Lower VIN Range
Hiccup Converter Powers Wireless Sensor from Indoor Lighting Transmit Rate vs Light Level
(Fluorescent)
LIGHT LEVEL (Lx)
0
TRANSMIT RATE (Hz)
4.5
4.0
2.0
3.5
3.0
2.5
1.5
1.0
0.5
0
3129 TA02b
2000
800 1200 1600400
BST1
VOUT V
OUT
3.6V
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VIN
VIN
1.8V TO 15V
RUN
MPPC
PWM
VCC
NC
NC
22nF
6.8µH
2M BAT54
976k
10µF
10µF
33pF
2.2µF
3129 TA03
PGND
VIN < 3.6V, IOUT = 100mA
VIN > 3.6V, IOUT = 200mA
BST1
VOUT VOUT
3.6V
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
PGOOD
GND
VCC
VIN
VIN UVLO = 3.5V
RUN
MPPC
PWM
VCC
NC
NC
22nF 4.7µH
1M
2M
976k
2.37M
4.42M
470µF
6.3V
+
10pF
22µF4.7µF
2.2µF
3129 TA02a
PGND
PV PANEL
SANYO AM-1815
4.9cm × 5.8cm
PULSED IOUT
25mA FOR 5ms
LTC3129
24
3129fc
For more information www.linear.com/LTC3129
Typical applicaTions
Solar Powered Converter with MPPC Charges Storage Capacitor
Average Output Current
vs Light Level (Daylight)
BST1
VOUT VOUT
4.8V
PGOOD
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VIN
VIN UVLO = 4.3V
RUN
MPPC
PWM
NC
NC
22nF 4.7µH
1M
VCC
392k
8.4cm × 3.7cm
47µF
CERAMIC
4.7µF 1F
3.09M
2.2µF
3129 TA04a
PGND
PowerFilm
SP4.2-37
SOLAR
MODULE
1M
+
COOPER BUSSMANN
PB-5R0V105-R
LIGHT LEVEL (Lx)
1000
OUTPUT CURRENT (mA)
100.0
10.0
1.0
0.1
3129 TA04b
100000010000010000
LTC3129
25
3129fc
For more information www.linear.com/LTC3129
Li-Ion Powered 3V Converter with 3.1V Input UVLO Reduces Low Battery IQ to 3µA
15V Converter Operates from Three to Eight AA or AAA Cells
Typical applicaTions
BST1
VOUT VOUT
3V
200mA
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VCC
VIN
Li-Ion
+
RUN
MPPC
PWM
NC
NC
22nF 4.7µH
2M
UVLO = 3.1V
1.27M
4.7µF
10pF
10µF
33pF 1.58M
2.2µF
3129 TA05
PGND
1.02M
VCC
BST1
VOUT
VOUT
15V
25mA MINIMUM
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VIN
VIN
2.42V TO 15V
RUN
22nF 10µH
10µF
25V 3.01M
2.2µF
3129 TA06
PGND
THREE TO EIGHT
AA OR AAA
BATTERIES
MPPC
PWM
NC
NC
10µF
255k
22pF
LTC3129
26
3129fc
For more information www.linear.com/LTC3129
Typical applicaTions
Energy Harvesting Converter Operates from a Variety of Weak Sources
Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications
Percentage of Added Battery Life vs Light Level and Load
(PowerFilm SP4.2-37, 30sq cm Panel)
LIGHT LEVEL (Lx)
100
ADDED BATTERY LIFE (%)
1000
100
10
1
3129 TA09b
10,000
1,000
AVERAGE LOAD = 165µW
AVERAGE LOAD = 330µW
AVERAGE LOAD = 660µW
AVERAGE LOAD = 1650µW
AVERAGE LOAD = 3300µW
VCC
BST1
VOUT VOUT
5V
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VIN
BAS 70-05
UVLO = 3.3V
RUN
22nF 4.7µH
10µF
3.32M
2.2µF
3129 TA07
PGND
MPPC
PWM
NC
NC
100µF
CERAMIC 1.02M
4.99M
3.01M
BAS 70-06
INPUT SOURCES:
• RF
• AC
• PIEZO
• COIL-MAGNET
10pF
22pF
BST1
VOUT
SW1 SW2
LTC3129
22nF
3.20V
BST2
PGOOD
FB
GND
VCC
VCC
VIN
RUN
MPPC
PWM
NC
NC
22nF 3.3µH
TOKO DEM2812C
10pF
2.2µF
VOUT
BAT54
470µF
6.3V
74LVC2G04
3129 TA09
PGND
FDC6312P
DUAL PMOS
PV PANEL
SANYO AM-1815
OR
PowerFilm SP4.2-37
4.7µF
VIN UVLO = 3.7V
4.99M 4.22M
2.43M
D1 D2
S2S1
G2
CR2032
3V COIN CELL
VOUT
3V TO 3.2V
G1
2.43M
22µF
R4
2.43M
15pF
2.2µF
+
LTC3129
27
3129fc
For more information www.linear.com/LTC3129
package DescripTion
Please refer to http://www.linear.com/product/LTC3129#packaging for the most recent package drawings.
3.00 ±0.10
(4 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
1.65 ±0.05
(4 SIDES)
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
PIN 1
TOP MARK
(NOTE 6)
0.40 ±0.10
BOTTOM VIEW—EXPOSED PAD
1.65 ±0.10
(4-SIDES)
0.75 ±0.05 R = 0.115
TYP
0.25 ±0.05
1
PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER
15 16
2
0.50 BSC
0.200 REF
2.10 ±0.05
3.50 ±0.05
0.70 ±0.05
0.00 – 0.05
(UD16 VAR A) QFN 1207 REV A
0.25 ±0.05
0.50 BSC
PACKAGE OUTLINE
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1700 Rev A)
Exposed Pad Variation AA
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1700 Rev A)
Exposed Pad Variation AA
LTC3129
28
3129fc
For more information www.linear.com/LTC3129
package DescripTion
Please refer to http://www.linear.com/product/LTC3129#packaging for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
MSOP (MSE16) 0213 REV F
0.53 ±0.152
(.021 ±.006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
16
161514131211 10
12345678
9
9
18
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.10
(.201)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ±0.127
(.035 ±.005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ±0.038
(.0120 ±.0015)
TYP
0.50
(.0197)
BSC
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102
(.065 ±.004)
0.1016 ±0.0508
(.004 ±.002)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0.280 ±0.076
(.011 ±.003)
REF
4.90 ±0.152
(.193 ±.006)
DETAIL “B”
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
LTC3129
29
3129fc
For more information www.linear.com/LTC3129
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision hisTory
REV DATE DESCRIPTION PAGE NUMBER
A 5/14 Clarified VCC Leakage to VIN if VCC > VIN: from –7µA to –27µA 4
B 10/14 Clarified typ VOUT Current in Sleep Mode
Clarified PGOOD Pin Description
Clarified Operation Paragraph
4
8
15
C 10/15 Changed MAX VCC Current Limit
Modified MPPC section
Modified Table 3
3
15
22
LTC3129
30
3129fc
For more information www.linear.com/LTC3129
LINEAR TECHNOLOGY CORPORATION 2013
LT 1015 REV C • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LTC3129
relaTeD parTs
Typical applicaTion
Dual VIN Application, Using the LTC4412 PowerPath™ Controller
PART NUMBER DESCRIPTION COMMENTS
LTC3103 15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current
VIN(MIN) = 2.2V, VIN(MAX) = 15V, VOUT(MIN) = 0.8V, IQ = 1.8µA,
ISD <1µA, 3mm × 3mm DFN-10, MSOP-10 Packages
LTC3104 15V, 300mA Synchronous Step-Down DC/DC Converter with
Ultralow Quiescent Current and 10mA LDO
VIN(MIN) = 2.2V, VIN(MAX) = 15V, VOUT(MIN) = 0.8V, IQ = 2.8µA,
ISD <1µA, 4mm × 3mm DFN-14, MSOP-16 Packages
LTC3105 400mA Step-up Converter with MPPC and 250mV Start-Up VIN(MIN) = 0.2V, VIN(MAX) = 5V, VOUT(MIN) = 0 5.25VMAX, IQ = 22µA,
ISD <1µA, 3mm × 3mm DFN-10/MSOP-12 Packages
LTC3112 15V, 2.5A, 750kHz Monolithic Synch Buck/Boost VIN(MIN) = 2.7V, VIN(MAX) = 15V, VOUT(MIN) = 2.7V to 14V, IQ = 50µA,
ISD <1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages
LTC3115-1 40V, 2A, 2MHz Monolithic Synch Buck/Boost VIN(MIN) = 2.7V, VIN(MAX) = 40V, VOUT(MIN) = 2.7V to 40V, IQ = 50µA,
ISD <1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages
LTC3531 5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MIN) = 2V to 5V, IQ = 16µA,
ISD <1µA, 3mm × 3mm DFN-8 and ThinSOT Packages
LTC3388-1/
LTC3388-3
20V, 50mA High Efficiency Nano Power Step-Down Regulator VIN(MIN) = 2.7V, VIN(MAX) =20V, VOUT(MIN) = Fixed 1.1V to 5.5V,
IQ = 720nA, ISD = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages
LTC3108/
LTC3108-1
Ultralow Voltage Step-Up Converter and Power Manager VIN(MIN) = 0.02V, VIN(MAX) = 1V, VOUT(MIN) = Fixed 2.35V to 5V,
IQ = 6µA, ISD <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages
LTC3109 Auto-Polarity, Ultralow Voltage Step-Up Converter and Power
Manager
VIN(MIN) = 0.03V, VIN(MAX) = 1V, VOUT(MIN) = Fixed 2.35V to 5V,
IQ = 7µA, ISD <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages
LTC3588-1 Piezo Electric Energy Harvesting Power Supply VIN(MIN) = 2.7V, VIN(MAX) = 20V, VOUT(MIN) = Fixed 1.8V to 3.6V,
IQ = 950nA, ISD 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages
LTC4070 Li-Ion/Polymer Low Current Shunt Battery Charger System VIN(MIN) = 450nA to 50mA, VFLOAT + 4.0V, 4.1V, 4.2V, IQ = 300nA,
2mm × 3mm DFN-8, MSOP-8 Packages
BST1
VOUT V
OUT
12V
SW1 SW2
LTC3129
22nF
BST2
PGOOD
FB
GND
VCC
VCC
VIN
RUN
MPPC
PWM
NC
NC
22nF
MBR0520
FDN338
BSS314
10µH
10µF
12V WALL ADAPTER INPUT
10µF
25V
3.01M
2.2µF
3129 TA08
PGND
VIN = 12V, IOUT = 200mA
VIN = 3.6V, IOUT = 50mA
324k
LTC4412
GND
VIN
CTL
Li-Ion
+
GATE
STAT
SENSE
10pF